[0001] This application claims the benefit to European patent application no. EP19164247.9 filed March 21, 2019, which is hereby expressly incorporated by reference as part of the present disclosure.

[0002] The present invention concerns an apparatus for coupling a hollow waveguide to planar transmission media, and a radar system comprising such an apparatus.

[0003] Waveguides and microstrip circuits are commonly employed media for propagation of microwaves. Waveguides typically are hollow conductive conduits which have a rectangular or circular cross section designed to propagate microwaves with a minimum of loss. Waveguides might include longitudinally- oriented ridged structures within the waveguide conduit to change the

propagation characteristics of the waveguides and to adapt them for particular applications. Single-ridge and double-ridge waveguides are widely used.

[0004] A microstrip circuit comprises a dielectric material separating a ground plane metallization from a signal carrying microstrip line metallization. Microstrip circuits are planar circuits which are preferred in some applications because they offer compactness and their cost is negligible when compared to the more expensive and complex waveguide conduits.

[0005] Substrate-integrated waveguides (called SIW) have a planar structure, too. Substrate- integrated waveguides are typically obtained by two electrically conducting, essentially congruent, stratified layers separated by a thin dielectric material. In order to operate as a SIW, the two conducting layers are connected on opposing edges parallel to the wave propagation direction. Quite often, these connections are provided by arrayed metal filled vias. At least one of the electrically conductive layers may extend beyond the metal filled via arrays and might be referred to as ground plane metallization. The other electrically conductive layer may be limited to a patch-like area, just leaving sufficient printed circuit board (PCB) processing margin beyond the outermost metal filled via. The vias provide for a lateral confinement and guidance of an

electromagnetic wave, which travels in the via-confined dielectric material cross section, if the lateral free space between the via arrays is larger than half signal wavelength, and the vias are placed close

[0006] Quite often it is necessary to combine different types of media and to provide for a contiguous propagation of microwaves along such daisy-chained segments. The respective systems are called hybrid systems. It is hence important to be able to provide for a suitable transition e.g. from a microstrip circuit to a waveguide conduit and vice versa.

[0007] A double-ridge waveguide to microstrip coupling is disclosed in

US4,973,925. An end portion of a flat microstrip circuit is situated inside an end section of the hollow conduit of a 3-dimensional double-ridge waveguide. The coupling is enabled by a reduction of the gap width between the two ridges of the double-ridge waveguide. The gap width is gradually reduced towards the end section of the hollow conduit so that the substrate of the microstrip circuit fits right in-between the two ridges of the waveguide, such that the microstrip line of the microstrip circuit is in good electrical and physical contact with one of the ridges. The respective double-ridge waveguide to microstrip coupling is suitable for a longitudinal coupling only.

[0008] Another example of a longitudinal double-ridge waveguide to

microstrip coupling is disclosed in US5,095,292. The coupling is provided by a microstrip-fed dipole being centered on the open end of a flange of the double ridge waveguide. A gap is situated between the dipole metallization and the open end of the waveguide in order to obtain a low VSWR (Voltage Standing Wave Ratio) coupling. Due to the close proximity between the dipole-shaped

metallization and the waveguide flange, the dipole's radiation efficiency into free air is low, and the gap between dipole conductors and waveguide flange represent open-ended slotline stubs in electrical series connection between an intermediate slotline section (the conductor gap which exists between the dipole arms) and the waveguide ridges. These slot line stubs need to be dimensioned for essentially quarter wavelength at the operating band center frequency, in order to offer a low-impedance, physically non-contacting connection. While the omitting of electrical shielding means seems to be convenient and has been claimed to provide a tolerance-insensitive solution, the insertion loss is

dominated by radiation leakage and the mutual isolation between adjacent signal channels is insufficient for modern radar sensor applications.

[0009] The above-mentioned examples concern in-line transitions, where the propagation direction inside the microstrip circuit and the propagation direction inside the waveguide have essentially the same orientation.

[00010] An example of a perpendicular hollow waveguide to coplanar

transmission line coupling is disclosed in US6,002,305. Three coplanar

conductors, namely a center conductor and two side conductors form a

microstrip-like Quasi-TEM (transverse electromagnetic wave) propagation medium. The transition is provided in that a short end portion of the center conductor sticks into the hollow waveguide conduit. Continuously tapered, curved slots between coplanar center and side conductors provide for a smooth, prolonged transition to the short center conductor end. The latter is often referred to as "E-probe", since it couples with the hollow waveguide's electrical field, which is enhanced by a back-short, placed a quarter propagating

wavelength behind the "E-probe" position. The disclosed arrangement is basically limited to rectangular, elliptical or circular hollow waveguides and cannot be easily applied to ridged waveguides, which might be the only viable low-loss medium due to limited lateral space in multi-channel modern radar sensors.

Between the longitudinal ridges of such reduced cross-section waveguides, there is no sufficient room for accommodating the "E-probe".

[00011] In modern RF-systems, the packing density increases as the demand for additional features or functionality is increasing. Modern radar systems, as for instance being used in vehicles, may for example comprise an array of antenna elements which have to be fed with appropriate RF signals so as to be able to shape the radar beam which is emitted by a vehicle's front radar, for example.

On the one hand, the space which is available for such a radar system typically is quite small and on the other hand the range requirement and the resolution demands are quite high. For increasing the angular resolution, and to allow for using low power CMOS (complementary metal-oxide-semiconductor) circuitry for transmit functions, there is a technology trend to increase the number of independent, yet coherent radar signal channels significantly (24 and more per monolithically integrated front end circuit). This trend increases the demand for compact (e.g. ridged) waveguide transmission media and suitable, equally compact waveguide to planar transmission line couplings.

[00012] Typically, the design of hybrid systems is difficult due to the very different nature of the microstrip circuits and waveguide conduits. The routing of the RF signal paths is one of the key issues if a densely packed, robust design of an RF-system is required.

[00013] Because of the recent trend toward miniaturization and integration, microwave circuitry is required which allows flexible connection and routing schemes and which provides for additional degrees of freedom as far as the 3- dimensional orientation of electromagnetic wave tracks is concerned.

[00014] It is thus an object of the invention to provide an open and flexible design approach for the dense integration of multiple channels in RF-systems, such as RF antenna systems.

[00015] In accordance with the invention, hollow waveguide-based antenna systems are preferred because of their low-loss characteristics. Waveguide-based antenna systems are preferred which employ very precise injection molded, metallized components. In order to be able to use the respective waveguides, a suitable and efficient transition between a planar circuit, such as a SIW circuit, and a hollow waveguide is essential. This transition is herein referred to as 2D- circuit to 3D-circuit transition.

[00016] This object is being solved by an apparatus in accordance with claim 1 and a radar system in accordance with claim 15. Advantageous embodiments can be derived from the dependent claims and the below description.

[00017] In accordance with the present invention, an apparatus is provided which comprises

- a dielectric substrate having a top plane and an opposite bottom plane with an at least partial metallization,

- a ridged waveguide, - a metal patch, integrated in said dielectric substrate, which comprises at least one conductive via hole connecting said metal patch and said metallization,

- a slot, preferably a H-shaped slot, being provided in said metal patch and having an orientation with respect to a first axis of said top plane being defined by an angle in the range 0° < angle < 360°,

wherein said ridged waveguide

- is field-coupled to said slot via an air gap, which extends parallel to said top plane, so as to be able to couple an electromagnetic wave travelling along said ridged waveguide via said air gap into said slot and/or to couple an electromagnetic wave emitted by said slot said air gap into said ridged waveguide.

[00018] The apparatus of at least some embodiments is designed to provide for a 2D-circuit to 3D-circuit transition in the Radar band, in particular for use in millimeter-wave automotive Radar systems and devices.

[00019] The apparatus of at least some embodiments comprises a SIW element being coupled to a single-ridge waveguide or to a double-ridge waveguide, where the respective waveguide either is arranged in a second plane which is parallel to the main plane of the SIW element, or where the respective waveguide has a perpendicular arrangement with respect to the main plane of the SIW element.

[00020] The apparatus of at least some embodiments comprises a SIW element being coupled to a microstrip circuitry. The employment of a SIW element has the advantage that it can be well coupled with a planar microstrip line on the one hand and with a waveguide on the other hand, in a fully shielded manner.

[00021] At least some embodiments of the apparatus comprise a microstrip circuit which has an optimized microstrip line to metal patch transition. The initial width of the microstrip line is increased to provide for an impedance adjustment. The transition comprises a portion/section where the microstrip line is split into two short branches, each of which taps the edge field of the SIW-patch.

[00022] The embodiments proposed herein have a number of advantages as far as aspects of system integration are concerned. The respective elements have a small footprint and can thus be used in connection with antenna arrays which require down to a l/2 grid (half free space wavelength in words) size or spacing.

[00023] Since the speed of light slows down in a dielectric substrate relative to the speed in vacuum, the wavelength for an electromagnetic wave of a given frequency is smaller in the dielectric substrate material than it is in vacuum. The wavelength is reduced by a factor of 1/VS _R , with the relative electric permittivity of the substrate material S _R . Typical low-loss substrate materials used in the millimeter wave range have a permittivity between 2.5 and 3.5, resulting in wavelength reductions of 36 to 47%, as compared to the wavelength in free space. For a SIW, the cut-off frequency, i.e. the frequency, above which a fundamental waveguide mode can propagate, is reduced by the same factor as the free propagating wavelength in the dielectric material. A SIW with lateral open distance between metal filled vias of 1.5mm thus exhibits a fundamental TE (transverse electric) mode, which can propagate above a cut-off frequency of ca. 57.7GHz, as compared to a hollow waveguide of same rectangular cross section, which supports wave propagation only above 100GHz.

[00024] The embodiments proposed herein can be used in connection with Millimeter wave Radar systems.

[00025] The embodiments proposed herein can be used in connection with antenna arrays which have a grid size or spacing of 2mm or less, which refers to l/2 in free space, at 75GHz.

[00026] The embodiments proposed herein are well suited for use in beam steering antennas where a wide scanning angle range (e.g. ±90°) with a single, unambiguous beam can only be achieved with a l/2 grid size or spacing.

[00027] Further details and advantages of the invention will be described in the following by means of embodiments and with reference to the drawings.

Fig. 1A shows a perspective explosive view of a first embodiment;

Fig. IB shows a top view of the waveguide of the first embodiment;

Fig. 1C shows an x-z cross-section of a portion of the first embodiment and a magnified view of the vicinity of the slot;

Fig. ID shows a top view of the planar part of the first embodiment; Fig. IE shows a perspective view with an x-z cross-section of the waveguide of the first embodiment;

Fig. IF shows a perspective bottom view of the waveguide of the first

embodiment;

Fig. 2A shows a top view of the components of the first embodiment where the waveguide is provided in a 0°-orientation;

Fig. 2B shows a top view of the components of the first embodiment where the waveguide is provided in a 45°-orientation;

Fig. 2C shows a top view of the components of the first embodiment where the waveguide is provided in a 90°-orientation;

Fig. 3A shows a perspective explosive view of a second embodiment;

Fig. 3B shows a perspective bottom view of the waveguide of the second

embodiment;

Fig. 4A shows a perspective explosive view of a third embodiment;

Fig. 4B shows a top view of the third embodiment;

Fig. 5A shows a perspective explosive view of a fourth embodiment;

Fig. 5B shows a top view of the fourth embodiment;

Fig. 5C shows a perspective cross-section of the fourth embodiment;

Fig. 6A shows a perspective explosive view of a fifth embodiment;

Fig. 6B shows a perspective bottom view of portion of the fifth embodiment;

Fig. 7 shows a perspective bottom view of the waveguide of a sixth

embodiment;

Fig. 8A shows a perspective explosive view of the sixth embodiment (3-port embodiment);

Fig. 8B shows a cross-section of the sixth embodiment parallel to the y-z- plane;

Fig. 9 shows a perspective bottom view of the waveguide of a seventh

embodiment;

Fig. 10A shows a perspective explosive view of the seventh embodiment (2-port embodiment);

Fig. 10B shows a cross-section of the seventh embodiment parallel to the y-z- plane;

Fig. 11 shows a top view of a further embodiment. [00028] In connection with the present description, terms are used which also find use in relevant publications and patents. It is noted however, that the use of these terms shall merely serve a better comprehension. The inventive idea and the scope of the patent claims shall not be limited in their interpretation by the specific selection of the terms. The invention can be transferred without further ado to other systems of terminology and/or technical areas. In other technical areas, the terms are to be employed analogously.

[00029] For the purposes of the present description and claims, some of the essential elements or building blocks of an apparatus 100 are defined and described, before details of various embodiments are being addressed.

[00030] All embodiments of the apparatus 100 comprise a substrate integrated waveguide circuit (SIW) 110 and a ridged waveguide 200, as illustrated in Fig.

1A, for example. Such an embodiment might further comprise a planar microstrip circuit 150. The planar microstrip circuit 150 is optional.

[00031] An x-y-z coordinate system is used herein in order to be able to describe the components/elements and their 3-dimensional configuration. Any reference to the 3-dimensional relationship of the components/elements refers to the configuration after these components/elements have been assembled into an apparatus 100. If the word "above" is used, for example, this word refers to the representation in the corresponding figure only. The apparatus 100 and its components/elements can have a different orientation when being used.

[00032] As mentioned, all embodiments of the apparatus 100 comprise a ridged waveguide 200. The ridged waveguide as such carries the reference number 200. In order to be able to distinguish double ridged waveguides and single ridged waveguides, the reference numbers 200.1 and 200.2 are used, respectively.

[00033] At least some embodiments proposed herein either comprise a double ridge waveguide 200.1 with a rectangular cross-section or a double ridge waveguide 200.1 with a square cross-section (no distinction is made herein as far as their respective reference numbers are concerned). [00034] The dimensions of an exemplary waveguide (here a double ridge square waveguide 200.1) are depicted in Fig. IB. In case of a double ridge square waveguide 200.1, the following rules apply: WA = WB and WC = WD. In case of a double ridge rectangular waveguide 200.1, the following applies: WA ¹ WB and/or WC ¹ WD.

[00035] A double ridged waveguide 200.1 comprises two longitudinally extending ridges 201 (cf. Fig. 1A). The ridges 201 of these waveguides 200 extend parallel to the z-axis and are provided in order to increase the useful (single-mode) bandwidth of the waveguide 200.1, if compared with a rectangular plain rectangular waveguide. However, this increase of the bandwidth is at the expense of gradually increased attenuation (depending on the amount of cut-off frequency reduction by introducing the ridges of width WE and open gap WF2 into a rectangular waveguide of dimensions WA x WB. Also, the power-handling capability is lowered, which however is of no concern in Millimeter wave Radar systems. The most interesting property of a ridged waveguide 200 within the scope of the present document is its capability to provide a lower cut-off frequency than a plain rectangular waveguide of same inside dimensions. In order to avoid excessive loss, a waveguide is normally operated at least 23% above its cut-off frequency, which translates to the prerequisite of fc < 61.75GHz e.g., for the Automotive Millimeter wave frequency band designation (76GHz - 81GHz). An inside dimension WA = WB = 1.5mm ... 1.6mm is therefore possible, with moderate attenuation increase, rendering possible adjacent multichannel signal tracks with 2mm (ca. l/2 grid size or centerline spacing).

[00036] The (optional) microstrip circuit 150 of at least some of the

embodiments comprises a dielectric substrate 152, a ground plane

(metallization) 153, and a microstrip line 151. The microstrip line 151 extends longitudinally along the x-axis and, if used in connection with a substrate integrated waveguide circuit (SIW) 110, is designed to serve as interface or port of the metal patch 104 of the SIW circuit 110. The microstrip line 151 can either be used to feed the metal patch 104 with an RF-electromagnetic wave, or it is used to guide an RF-electromagnetic wave away from the metal patch 104. [00037] If a planar microstrip circuit 150 is employed together with a planar SIW circuit 110, then both circuits 150 and 110 define a common main plane (herein referred to as top plane TP).

[00038] The mode of propagation of such a microstrip line 151 is a quasi-TEM (transverse electromagnetic) transmission mode where the electric field lines in the substrate 152 underneath the microstrip line 151 have an orientation parallel to the z-axis. A quasi-TEM wave is propagating along the microstrip line 151 because part of the wave is travelling through the substrate 152 underneath the line 151 and part of the wave is travelling through the air above the line 151.

The term "quasi" indicates that this wave only resembles a TEM wave because it is propagating through two different media (substrate and air), which causes small longitudinal field components to occur. The direction of propagation is along the microstrip line 151 (either parallel to the positive x-axis or parallel to the negative x-axis).

[00039] Inside the SIW circuit 110 a TE transmission mode (its fundamental waveguide mode) is propagating. This propagation takes place inside the substrate 101 of the SIW circuit 110. Fig. 1C indicates the electric field (E-fields) orientations inside the substrate 101 on the left hand side (E2) and right hand side (El) of the slot 103. If the two E-field orientations have an opposite sign (referring to 180° phase difference, or push-pull excitation), the E-field E3 inside the gap 207 has an orientation parallel to the negative x-axis (cf. Fig. 1C). If El and E2 are of equal sign (i.e. would both be noted with an arrow in -z axis direction), E3 would vanish (in this case, the slot is not excited).

[00040] The embodiment of Fig. 1A - IF has a central plane of symmetry which has an orientation perpendicular to the plane of the drawing. The dash- dotted line in the middle of Fig. 1C represents the line where the central plane of symmetry intersects the x-z-plane. The respective plane of symmetry resembles an ideally electrically conductive boundary condition (the tangential E-field component vanishes here, i.e. all electrical field lines terminate normal to the plane). This means that the overall constellation of an SIW circuit 110 with a gap GP and a double ridged waveguide 200.1 above can be cut along the central plane of symmetry into two identical halves. If one rotates each of these two halves of the constellation by 90 degrees, an equivalent single ridged constellation is obtained where a single ridged waveguide 200.2 is situated so that it extends parallel to the top plane TP. Details of such a constellation with a single ridged waveguide 200.2 are described in connection with Fig. 7 and Figures 8A, 8B.

[00041] Fig. 7 was introduced in order to be able to show the structural similarities of the double ridged waveguide embodiments and the single ridged waveguide embodiments. Fig. 7 shows a perspective bottom view of a single ridged waveguide 200.2 which has a central gap 214 being formed in a

longitudinal ridge 201. This ridge 201 extends parallel to the y-axis. The central gap 214 is being formed so that it extends parallel to the x-z-plane. The embodiment of Fig. 7 has a central symmetry plane Sy, which in Fig. 7 is indicated by a dashed lines rectangle. If this embodiment is cut along the symmetry plane Sy into two identical halves, and if these two halves are folded so that the ridges 201 are facing each other, then a constellation is obtained which is similar to the double ridged waveguide embodiment of Fig. 3B, for example (the protrusions in Fig. 3B are not contained in this folded new constellation).

[00042] The single ridged waveguide 200.2 of Fig. 7 comprises two recessed segments 201.2 and two raised segments 201.1 of the ridge 201. The gap 214 is formed in the middle of the two raised segments 201.1. After the virtual cutting and folding of the two halves, the two raised segments 201.1 form the

transformer 206 (cf. Fig. IE), the function of which will be described later.

[00043] The constellation of the embodiment of Fig. 8A, 8B is based on the constellation of the embodiment of Fig. 7. It shall be noted, though, that the central gap 214 of Figs. 7, 8A and 8B offers a useful degree of design freedom, but its depth and / or width can be modified (accompanied by appropriate changes of other geometrical parameters of the embodiment) until the point, where it completely vanishes. This is possible without significant deterioration of the overall circuit performance.

[00044] Fig. 9 shows a perspective bottom view of another single ridged waveguide 200.2. This embodiment, however, does not feature a central gap 214 (i.e., its width or depth has been set to 0). A gap 214 could be introduced in this configuration as well, offering additional degrees of design freedom). The ridge 201 of Fig. 9 extends parallel to the y-axis and it comprises a recessed segment 201.2 and a raised segment 201.1. This embodiment of the apparatus 100 comprises one protrusion 215. This protrusion 215 is formed so that it protrudes from the horizontal facet of the recessed segment 201.2. The constellation of the embodiment of Fig. 9 basically corresponds to one half of the constellation of the embodiment of Fig. 7, where one of the ridged waveguide ports is unused for signal transmission, but reactively loaded (reflection coefficient of magnitude 1).

[00045] The constellation of the embodiment of Fig. 10A, 10B is based on the constellation of the embodiment of Fig. 9. Fig. 10B shows how the single ridged waveguide 200.2 is mounted on top of the SIW circuit 110. Only one via 108 is shown. The SIW circuit 110 might however comprise more than one via.

[00046] The optional protrusion 215 serves as capacitive element. This capacitive element serves the same function in the RF-regime as the two vias 109 which in the embodiment of Fig. ID form an inductive iris.

[00047] At least some of the embodiments comprise vias 109, serving as inductive iris and/or a protrusion 215, serving as capacitive element. Both achieve a wideband excellent matching of the two feed ports 101 and 201, and can be used alternatively or in combination.

[00048] The longitudinal section Fig. 10B shows the parallel air gap GP of the apparatus 100. In this constellation, where the waveguide 200.2 and the SIW circuit 110 are assembled, the ridge 201 is hanging down, that is, the lower facet 204 of the raised segment 201.1 of the ridge 201 is facing the top plane TP of the SIW circuit 110. The parallel air gap GP is defined by the top plane TP on the one side and by the lower facet 204 on the other side. The position of the slot 103 is visible, too. The uppermost layer of the SIW circuit 110 is situated parallel to the x-y-plane. The raised portion 201.1 of the waveguide 200.1 provides for a gap GP (cf. Fig. 10B) with a precisely defined gap width Gw.

[00049] Since a TEM transmission mode cannot propagate in a waveguide 200, an appropriate mode conversion and impedance transformation is required. The respective transition, as proposed herein, converts the SIW TE mode or the microstrip quasi-TEM transmission mode through the SIW TE mode into a TE mode inside the waveguide 200. This conversion or transition is herein referred to as field coupling, since there is no direct conductive connection between the SIW circuit 110 and the ridge(s) 201 of the waveguide 200.

[00050] The elements which influence the field coupling characteristics are designed with the objective to keep the field disturbance and stored blind energy at a minimum. In other words, no high-Q resonance effects are encountered throughout this transition. This means that this transition has small group delay and low dissipation loss. The equivalent circuit of these transitions generally resembles a wideband 2- or 3-pole bandpass.

[00051] The dielectric substrate of at least some of the embodiments might comprise two planar sections 101 and 152. The broken line Y1 in Fig. 1A shows a separation of these two sections 101 and 152.

[00052] It is also conceivable that the SIW substrate 101 of at least some of the embodiments comprises two different material layers whereas one of these two layers serves as substrate 152 of the microstrip circuit 150.

[00053] At least some of the embodiments comprise a common substrate 101 which carries the microstrip line 151 and the SIW circuit 110. In this case, the same dielectric material is used for both the microstrip circuit 150 and the SIW circuit 110.

[00054] The dielectric substrate 101, or the substrates 101, 152 define a top plane TP, as mentioned. This top plane TP is visible in Fig. 1C, 8B, and 10B, for example. The top plane TP comprises a metal structure or pattern 102 (cf. Fig. ID). Part of this metal structure 102 is designed as metal patch 104 and part of this metal structure 102 may be designed as microstrip line 151, if such a microstrip line 151 is provided. The top plane TP is herein also referred to as circuit side or main plane.

[00055] The dielectric substrate 101, or the substrates 101, 152 comprise an at least partially metallized bottom plane BP (cf. Fig. 1C) opposite to the top plane TP. The embodiments of Figures 1A, 1C and lDhave a bottom plane BP with a metallization 153. The metallization 153 may serve as ground plane if connected to the ground potential. [00056] At least some embodiments of the apparatus 100 comprise a bottom plane BP which is partially metallized. The bottom plane BP may, however, also be completely metallized.

[00057] At least some embodiments of the apparatus 100 comprise a microstrip circuit 150 which has an optimized microstrip line 151 to metal patch 104 transition 157. The initial width (parallel to the y-axis) of the microstrip line 151 is increased so as to provide for an impedance adjustment. The transition 157 comprises a portion/section which is wider than the microstrip line 151, if viewed from above (cf. Fig. 1A and ID), so that the microstrip line 151 is split into two short, high-impedance branches 158, each of which taps the edge field of the SIW-patch 104 at a point in suitable distance from the x/z plane.

[00058] In the present embodiment of the apparatus 100, the transition 157 comprises a transversal slit 156, as illustrated in Figures 1A and ID, so as to split the microstrip line 151 into two branches 158. This slit 156 is called transversal slit since it is perpendicular to the wave propagation direction in the microstrip line 151.

[00059] The transition 157 is designed for a good and very space efficient coupling of the microstrip line 151 with the patch 104.

[00060] Optionally, the slit 156 is covered using a solder resist so that no solder material will be able to bridge this slit 156 during a possible assembly step, employing a soldering process.

[00061] The SIW circuit 110 of at least some embodiments comprises the metal patch 104 being situated on the top plane TP. The metal patch 104 may have a square or rectangular footprint in the x-y-plane. The embodiment of Figures 1A, 1C and lDhas a metal patch 104 with a rectangular footprint. The corner shape of the metal patch 104 (beyond the outermost conductive vias) are free of choice, rounded (as shown) or chamfered.

[00062] The selective patterning of the microstrip line 151 and/or the metal patch 104 is well known in the art, and may be performed using

photolithographic techniques. [00063] The SIW circuit 110 of at least some embodiments further comprises a waveguide section the shape/size of which is defined by a number of conductive via holes 105. The conductive via holes 105 (cf. Fig. ID) connect the metal patch 104 and the at least partially metallized bottom plane BP. The vias 105 provide for a lateral and waveguide back-short. The difference of electrical potential between the patch 104 on the top plane TP and the metallized bottom plane BP is reduced to zero at or close to the vias 105.

[00064] At least some embodiments of the apparatus 100 comprise

circumferential lines of conductive via holes 105. Fig. ID shows a top view of the circuitry of the apparatus 100. In the present embodiment, there are inner circumferential guide lines 106 and outer circumferential guide lines 107 used for defining the via array tracks. The inner circumferential lines 106 and the outer circumferential lines 107 both have a U-shape, if viewed from above.

[00065] Due to the circumferential arrangement of vias 105, the

electromagnetic field of a wave travelling under the metal patch 104 is

essentially confined within the inner circumferential lines 106 (see Fig. ID).

[00066] At least some of the embodiments may comprise a double row (two parallel lines) of vias 105. In these embodiments, there are inner lines 106 of conductive via holes 105 and outer lines 107 of conductive via holes 105, and the via holes 105 of the inner lines 106 are preferably offset with respect to the via holes 105 of the outer lines 107. That is, the via holes 105 of the inner lines 106 mesh with the via holes 105 of the outer lines 107. The meshed

arrangement and about A _m /4 lateral distance between guide lines 106, 107 provides for an improved short circuit on the sides of the substrate integrated waveguide circuit 110 (i.e. reduced wave leakage into the substrates 101, 152 through the via-fences).

[00067] In at least some of the embodiments, the distance (pitch) PI (cf. Fig. ID) between two adjacent vias 105 corresponds to l/4.

[00068] The patch 104 in conjunction with the vias 105 and with an H-shaped slot 103 are designed to serve as a wave coupling element. The slot 103 basically causes the propagation of the wave to leave the common 2-dimensional top plane TP of the SIW circuit 110 and to enter the conduit of the waveguide 200 being situated above.

[00069] In at least some of the embodiments, the slot 103 is cut or etched into the metal patch 104 so that the substrate 101 is exposed within the boundary of the slot 103.

[00070] The term H-shaped is herein used in order to describe slots 103 which have a longitudinal, narrow middle section and two opposite end sections which are wider than the middle section. At least some of the embodiments proposed herein comprise an elongated slot 103 having the shape of a bar-bell (see for example Fig. ID) with a narrow segment in the middle and wider sections at the opposite ends. The dimensions and the shape of the H-shaped slot 103 can be chosen so as to manage higher local waveguide modes, which have influence on the center frequency adjustment of the useful operational band.

[00071] At least some of the embodiments are designed so that the H-shaped slot 103 is operated in a resonant mode if used in the Gigahertz frequency regime. The respective resonant frequency is not predefined by the slot shape alone, but involves the complex interaction of local higher order modes in the ridged waveguide 200 and the SIW volume surrounding the slot 103. A short circuit (e.g. a short established by one or two metallized vias 108) between the metal patch 104 and the ground plane metallization 153 right behind the slot 103 "forces" the electric field from the z-axis oriented direction within the input section of the patch 104 into the slot 103 bearing plane TP (x-y plane). Thus, the slot 103 can excite an electrical RF field inside a sector above the top plane TP, which is either established between the one ridge 201 of a single-ridged waveguide 200.2 (cf. Figures 8A, 8B, 10A, 10B) and the metallization of the top plane TP, or between the two ridges 201 of the double-ridged waveguide 200.1 (cf. E-field E3 in Fig. 1C).

[00072] At least some of the embodiments may comprise one or two additional vias 108 at or close to the rear side (if viewed from the microstrip line 151) of the slot 103. This via 108, or these vias 108 provide for the above mentioned short circuit between TP and BP or so-called "backshort" of the slot 103. [00073] All of the embodiments comprise an air gap GP between the slot 103 and the respective ridge 201 (cf. Fig. 8B, 10B.) or the two ridges 201 (cf. Fig. 1C).

[00074] Fig. 1C shows an x-z cross-section of a portion of the first embodiment and, on the right hand side, a magnified view of the direct vicinity above the slot 103. As one can see in Fig. 1C, there is an air gap GP right above the slot 103. This air gap GP is herein referred to as parallel air gap GP. The word parallel refers to the fact that this gap GP has an orientation parallel to the top plane TP, respectively parallel to the x-y plane.

[00075] Due to the fact that in this embodiment a double ridged waveguide 200.1 with two ridges 201 is employed, there is a gap 207 between these ridges 201 as well. This gap 207 extends parallel to the z-axis. The width WF1 of the lower part of the gap 207 is chosen so that the end facets 204 of the ridges 201 and the protrusions 205 do not touch or bridge the central slot portion of the gap 103 in the cross section of Fig. 1C. The width WF2 of the upper part 207 of the gap is larger, that is, WF2 > WF1.

[00076] If an appropriate width WF1 is chosen in relation to the gap 103 width, the waveguide 200.1 can be rotated about the z-axis in at least steps of 90°, without the end facets 204 and protrusions 205 to touch or bridge the gap 103.

It is one of the purposes of providing protrusions 205, to decouple design parameters WF1 and slot 103 width. Three examples of embodiments with rotated waveguides 200.1 are illustrated in Figures 2A - 2C.

[00077] The SIW circuit 110 of at least some embodiments comprises a metal patch 104 with additional vias 109 close to the transition 157, as illustrated in Fig. ID. Although these vias 109 can be interpreted as termination points (last in a row) of vias 105, they can fulfill a special role, if their positions relative to the microstrip transition 157 are optimized for overall matching and reduced radiation leakage. The vias 109 close to the transition 157 form an inductive element with respect to the tap positions of the two high impedance microstrip branches 158 on the SIW input boundary.

[00078] Other Quasi-TEM transmission lines like Coplanar Waveguide or Grounded Coplanar Waveguide can be employed and can replace the microstrip transmission line 151 in the following straight-forward manner: the signal carrying center conductor is attached at the joint of the two high impedance microstrip branches 158, resembling the microstrip 151. The outer ground- potential conductors of the (grounded) coplanar waveguide are each connected with the SIW patch 104 near to one via 109, in order to effect a low-reactance connection to the SIW ground plane BP. Grounded Coplanar transmission line can be seen as modification of microstrip line, by additionally providing ground potential carrying conductor strips alongside the microstrip, which is then called "center conductor". In order to avoid unintended leakage and coupling effects, these ground strips are connected to the BP conductor by a series of grounding vias, similar to vias 105 (e.g. depicted in Fig. ID) and with similar pitch PI . The function of the overall transition from Quasi-TEM port via SIW section into ridged waveguide 200.1 or 200.2 stays the same. Grounded Coplanar transmission line is sometimes preferred over microstrip (despite the extra effort and cost of plated-through grounding vias), because it has lower radiation tendency and lower mutual coupling between signal channels. Coplanar Waveguide (only requiring a single sided metal structure) might be preferred for its higher mechanical compliance when used as flexible interconnect element.

[00079] All embodiments further comprise the above-mentioned ridged waveguide 200. This ridged waveguide 200 is situated above the metal structure 102. That is, the ridged waveguide 200 is situated in the half-space above the top plane TP.

[00080] Depending on the respective embodiment, either a single ridged waveguide 200.2 or a double ridged waveguide 200.1 is employed. Figures 1A - IF, 2A - 2C, 3A, 3B, 4A, 4B, 5A - 5C, 6A, and 6B show embodiments comprising a double ridged waveguide 200.1 and Figures 7, 8A, 8B, 9, 10A, 10B, and 11 show embodiments comprising a single ridged waveguide 200.2.

[00081] A single ridged waveguide 200.2 may be employed to extract the electromagnetic wave from the inner SIW volume into the upper half-space above the top plane TP and redirect it to certain direction(s) parallel to the x-y plane.

[00082] A double ridged waveguide 200.1 may be employed in order to extract the electromagnetic wave from the inner SIW volume into the upper half-space, and to then conduct the electromagnetic wave parallel to the z-axis, with certain x-y plane aligned polarization, for example.

[00083] The double ridge waveguide 200.1 can be amended with an optional waveguide knee element in order to redirect the electromagnetic wave from the propagation direction parallel to the z-axis into a direction parallel to the x-axis or y-axis, if desired. Depending on the design of the waveguide knee element and the chosen orientation of the double ridge waveguide 200.1, the

electromagnetic wave may be caused to travel parallel to the x-axis or parallel to the y-axis, after it was directed away from the z-axis. Thus, an arbitrary spacing between planar circuit and ridged waveguide 200 propagation plane (parallel to x-y plane) can be introduced, which allows for higher circuit density on employed e.g. radar system PCBs, where only the very compact transition area, i.e. SIW patch 104 occupies PCB real estate. Any passive or active electronic circuit may occupy the same location in the x-y plane as ridged waveguide conduits, which may run parallel to x-y plane, but on different z level.

[00084] In at least some embodiments, the waveguide knee element is designed as E-plane bend.

[00085] Before addressing further details, some of the geometrical aspects of a double ridged waveguide 200.1 are described. As one can see in Fig. IB, for example, the double ridged waveguide 200.1 comprises four side walls a, b, c, d which together define a rectangular or square cross-section. The double ridged waveguide 200.1 further comprises a first ridge 201 protruding from the first side wall a into the hollow space 202 and a second ridge 201 protruding from a second side wall c, opposite to said first side wall a, into the hollow space 202 so that the hollow space 202 of the double ridged waveguide 200.1 has an FI- shaped cross-section.

[00086] All embodiments have in common that there is a narrow and well defined parallel gap GP between the top plane TP and the lower end facets 204 of two raised segments 201.1 of the single ridged waveguide 200.2 (cf. Fig. 8B), or the lower end facet 204 of the raised segment 201.1 of the single ridged waveguide 200.2 (cf. Fig. 10B), or the end facets 204 of the two ridges 201 of the double ridged waveguide 200.1 (cf. Fig. 1C). [00087] At least some of the embodiments comprise a double ridged waveguide 200.1 which has an overall length Lz (parallel to the z-axis). The two ridges 201 are a little shorter, that is the lower end facets 204 of the two ridges 201 do not touch the metal patch 104 after the apparatus 100 has been assembled. It is schematically indicated in Fig. 1A that the lower end facets 204 are recessed with respect to the lowest point of the waveguide 200.1. A transparent view is provided in Fig. 1A and broken lines are used to show these facets 204 at the lower end of the waveguide 200.1.

[00088] At least some of the embodiments comprise a double ridged

waveguide 200.1 which has precisely defined (contact) protrusions 205

positioned at the lower end facets 204. Fig. 1A shows an embodiment with one such protrusion 205 at each of the lower end facets 204. Broken lines are used to show the position and shape of these protrusions 205. The two protrusions 205 are clearly visible in Figures 1C, IE and IF.

[00089] In all embodiments four protrusions 205 can be used instead of two protrusions 205.

[00090] In all embodiments the protrusions 205 can have another shape than the conical shape which is shown in all Figures.

[00091] In Fig. 1A the contact areas, where the two protrusions 205 touch the metal patch 104 are shown by means of empty circles which are arranged in alignment with the x-axis. The two protrusions 205 are designed to conductively connect the end facets 204 of the ridges 201 of the waveguide 200 and the patch 104. Their mutual distance (parallel to the x-axis) is chosen so that there is enough room for the waveguide 200 to be rotated about the z-axis (if needed) without the protrusions 205 even partially covering the slot 103.

[00092] The mutual distance between protrusions 205, the gap width Gw (parallel to the z-axis) and gap width WF1 as well as low-impedance (waveguide transformer) section length LT are important to the overall operation of the transition. The mutual distance between protrusions 205 is usually perfectly controlled and reproduced by the piece-part production process (e.g. metalized plastics) for ridged waveguide part 200. The same holds for the height of the protrusions 205, which control the gap width Gw. However, the assembly process must provide for a positive, physical contact between protrusions 205 and the top plane TP of the SIW patch 104. The distance between contact points of protrusions 205 with surface 204 on one hand and the corners of the lower ridge faces on the other hand typically resembles a small series inductance, which has an influence on the required transformer section length LT, which is chosen appropriately in the design process. Care must be taken to reproduce the gap width WF1 with high accuracy, which might require special measures for controlling e.g. a galvanic plating step in ridged waveguide part 200 production process.

[00093] In case of a transition between an H-shaped slot 103 and a double ridged waveguide 200.1, the gap GP allows for the electrical field in the center region of H-shaped slot 103 to evolve. The protrusions 205 define the exact height GW, when put in physical contact with metal patch 104, and touch it e.g. within circular regions, which are depicted as 2 empty circles in Fig. 1A (the first of which is located between positions of SIW backshort vias 108, the second in the same distance to the center beyond the slot 103). The physical contacts are electrical contacts as well and define short lengths of local parallel plate lines (with conductors lower end facet 204 and metal patch surface 104, respectively), each terminated in a short circuit. The mutual distance between protrusions 205 (in x direction, refer to Fig. 1C) is typically much less than l/4. Therefore, the two shorted gap sections left and right of slot 103 are representing small inductances, which are series-connected between slot 103 and double ridge waveguide 200.1. Although this arrangement seems to be an unnecessary complication in first sight (as compared to a direct contact between lower end facet 204 and metal patch 104 surface), it yields a number of clear advantages:

- the contact points are precisely defined and confined by protrusions 205 positions, rather than by surface roughness-dependent, less reproducible contact locations when employing the full double ridge waveguide material cross section, as shown in Fig. IB.

- potential risk of partial covering and shorting of slot 103 by ridges 201 is vastly reduced, with the result of reduced position tolerance sensitivity between ridged waveguide 200.1 and slot 103. As an example, a double ridge waveguide 200.1 with inner dimensions WA = WB = 1.6mm, WE=0.5mm and WFl=0.4mm can be displaced with respect to the slot 103 center by ±50pm in both x- and y-directions (combined) and may be rotated around z-axis by ±10°, while keeping the 25dB reflection loss bandwidth beyond 71GHz to 85GHz. This is 2.8 times the desired full Automotive Radar bandwidth of 76GHz - 81GHz.

- The low impedance ridged waveguide transformer section length LT can be reduced with introduction of the gap GP of height Gw (see Fig. 1C). This eases piece part production and galvanic plating.

[00094] In case of a transition between an H-shaped slot 103 and a single ridged waveguide 200.2, the constellation in the vicinity of the slot 103 defines the coupling and impedance matching between SIW 110 and ridged waveguide 200.2. As can be seen in Fig. 8B, the ridged waveguide 200.2 is now constituted by ridges 201 as part of waveguide body 200, on one hand, and the surface of the electrically conducting top plane TP of the SIW 110, which is integrated inside substrate 101, on the other hand. Electromagnetic waves, traveling inside the SIW 110 towards the slot 103, e.g. in -x direction (into the paper plane) are coupled to the upper halfspace and excite waves inside that travel in opposite (+y and -y) directions. Thus, a power division function results (a microwave 3- port is obtained). The central region can be construed in much the same way as the respective central region of the double ridge waveguide transition shown in Fig. 1C. Now, the central double-ridge gap portion is terminated in a short circuit nearby (the central gap 214 is vertically kept < < l/4), while the single ridge gap portions constituted by end facets 204 and surface of the SIW top plane TP are continued and support travelling waves. Equivalently, now the central gap 214 appears series-connected with the two ridged waveguide branches (with respect to potential difference or voltage across the slot 103). It represents a small inductance, which can be used as design degree of freedom and tuning means for the impedance matching effectuated mainly by the small-gap portion of the outgoing ridges. It has to be noted, that no protrusions 205 or other physical / electrical contact points between ridges 201 and top plate TP are necessary here. These are resembled by the short-circuit condition at the ground of central gap 214 (in other words, TP of Fig. 1C has been exchanged with perfect electrical conducting symmetry condition in the x-z plane (dashed line surrounded area Sy of Fig. 7). [00095] At least some of the embodiments comprise a double ridged waveguide 200.1 which has a l/4 transformer 206, as illustrated in Fig. IE. This l/4 transformer 206 basically is a section of the waveguide 200.1 where the gap 207 between the ridges 201 is more narrow (that is, WF1 < WF2). Note, that l/4 in this context refers to the principal impedance transforming function, not to the exact physical length of this section (the step discontinuity at the transition from WF1 to WF2 and the gaps GP usually necessitate a reduction of the narrow gap section length).

[00096] At least some of the embodiments comprise a double ridged

waveguide 200.1 which has a contact ring and/or contact protrusions 208 at the lower end facet 209 of the walls a, b, c, d. Details are visible in the Figures IE, IF.

[00097] In Fig. 1A one can see contact protrusions 208.1 (at all four corners of the waveguide 200) which have a radial orientation with respect to the z-axis. In Fig. IE one can see (at one of the corners of the waveguide 200) a contact protrusion 208.1 which has a radial orientation with respect to the z-axis and short sections of a circumferential contact protrusion 208.2. Fig. IF shows all four radially oriented contact protrusions 208.1 and the circumferential contact protrusion 208.2.

[00098] The contact ring and/or contact protrusions 208 are employed in order to be able to reliably connect the grounded part of the metal patch 104 with the walls a, b, c, d of the waveguide 200. That is, the contact ring and/or contact protrusions 208 establish a conductive or low-reactance RF connection. This can be either achieved by using a conductive glue or solder material when

assembling the parts of the apparatus 100, or by using non-conductive glue with a very small residual thickness in the contact zone. In Fig. 1A one can see the footprint corresponding to the four radially oriented contact protrusions 208.1 and the circumferential contact protrusion 208.2 depicted by a frame-like area. The circumferential contact closes the coupling volume around slot 103 in an EM- tight fashion. Thus, excess loss by radiation can only emanate from the microstrip-line to SIW transition 157 and is very low.

[00099] At least some of the embodiments proposed herein comprise a waveguide 200 where at least part of the lower end facet 209 is recessed with respect to the radially oriented contact protrusions 208.1 and circumferential contact protrusion 208.2, as illustrated in Fig. IF. When applying liquid

conductive or non-conductive glues or when reflowing a solder material, this material will predominantly fill the area between the metal patch 104 and the recessed end facet 209. The joining process design may optionally employ capillary forces, which exist in a shallow gap between circumferential contact protrusion and metal patch 104, e.g. for automatically filling any voids with the contacting agent.

[000100] Instead of using contact protrusions 208.1 and/or a circumferential contact protrusion 208.2, in all embodiments the lower end facet 209 might be butt-coupled to the patch 104.

[000101] The height (parallel to the z-axis) of the radially oriented contact protrusions 208.1 and circumferential contact protrusion 208.2 exactly defines the position of the lower most portions of the two or four protrusions 205 relative to the level of the metal patch 104. Optionally, a small protrusion of elements 205 beyond the z-level of contact protrusions 208.1, 208.2 can be devised to obsolete any contacting agent between these protrusions 205 and the metal patch 104. When cross-linking or hardening the glue or when solder material solidifies, a minute shrinkage occurs, which can be employed for generating sufficient and permanent mechanical contact pressure on elements 205. In this way, contact agent spreading, e.g. into the open slot 103 area or a partial filling of gap GP can be avoided and hence associated deterioration of transition performance by assembly process spread can be mitigated.

[000102] The contrary method is possible as well : protrusions 205 may be made slightly shallower than the circumferential contact protrusion 208.2 (and appropriately shaped), with the proviso that a small, exactly defined amount of a contacting agent (preferably solder paste or conductive glue) shall be applied between the protrusions 205 and the metal patch 104. In this case, the solder or conductive glue fillets conform to the existing gap and establish an electrical, low-resistance contact. Care must be taken to not alter the resulting coupling slot 103 shape by migrating contact agent.

[000103] Since the position of the lower most portions of the two or four protrusions 205 relative to the metal patch 104 if exactly and reproducibly defined by this joining technology, the parallel gap GP between the end facet(s) 204 of the ridge(s) 201 and the metal patch 104 is exactly defined, too. The width Gw of the parallel gap GP (measured parallel to the z-axis) is important because it has a direct impact on the broadband matching of the transition and the apparatus 100.

[000104] As can be seen in the various Figures, with the exception of Figs. 4A and 4B, the H-shaped slot 103 is rotated by 45° with respect to the x-axis. The 45° orientation of the slot 103 is advantageous since it facilitates a number of different constellations, as will be addressed hereinafter. On the one hand, the specific orientation of the slot 103 provides for a slot resonance frequency which is sufficiently low. On the other hand, the specific orientation of the slot 103 allows the waveguide 200 to be rotated about the z-axis in 45° angular increments, for example.

[000105] Instead of a +45° orientation of the slot 103, a -45° orientation of the slot 103 can be used in connection with all embodiments. A slot 103 with a +45° orientation and a slot 103 with a -45° orientation produce the same coupling result. However, certain configurations, namely those with cross-polarized ridge orientation (ridges that are oriented normal to the plane of in- and outgoing wave propagations, as shown e.g. in Fig. 2C), the phase is changed by 180°, when the slot 103 is rotated by 90° around the z-axis.

[000106] Figures 2A - 2C show an H-shaped slot 103 with a +45° orientation. Fig. 2A shows a situation where the waveguide 200 is going to be mounted in a 0° orientation. Fig. 2B shows a situation where the waveguide 200 is going to be mounted in a 45° orientation and Fig. 2C shows a situation where the waveguide 200 is going to be mounted in a 90° orientation. It is to be noted that the planar elements (SIW 110 plus the optional microstrip line 151) of the apparatus 100 neither have to be rotated nor is it necessary to otherwise alter this/these planar element(s).

[000107] It is important that the width WF and length WE of the longitudinal gap 207 between the two ridges 201 is chosen so that the narrow middle segment of the H-shaped slot 103 sits between these ridges 201. Fig. 5B shows a respective example where the narrow middle segment of the H-shaped slot 103 is visible. The elongated slot 103 interacts with the waveguide 200.1 so that RF energy is transferred between fundamental modes of waveguide 200.1 and SIW cavity.

The excitation of higher order modes is relatively low (both ridged waveguide 200 and SIW 110 are used relatively close to their respective cut-off frequencies and the slot 103 is intentionally kept slim), therefore the different waveguide orientations of Figs. 2A and 2C, for example, differ only marginally in their VSWR characteristics.

[000108] A number of different material combinations are suitable for

implementing the constituents of the apparatus 100. First, examples for substrates 101 and 152 are given. For Automotive Radar, namely 77/79GHz Safety and Reliability Applications, lowest loss in the Millimeter wave range and high processing and environmental performance reliability are demanded. Most ADAS (Advanced Driver-Assistance System) front ends are fabricated as hybrid multilayer PCBs, the outer layers or at least one top layer of which are made from low-loss (small dielectric loss) RF material. These materials are mostly equipped with special-grade, low-profile (reduced roughness) copper cladding, which minimizes conductor (resistive) losses as well. Depending on the cost / performance trade-off, modern low-loss thermoset or Hydrocarbon materials like ROGERS® RO4830™, TACONIC® TSM-DS3, ISOLA® Astra MT or Panasonic Corporation R-5515 material might suffice. Even lower loss materials based on PTFE, like ROGERS® R03003G2™, TACONIC® NF30 or NELCO N9000 can be used. Another type of suitable material is LCP (Liquid Crystal Polymer), a class of partially crystalline aromatic polyesters. This is especially interesting for flexible interconnect solutions, which may be based on Quasi-TEM transmission line as microstrip, coplanar line or grounded coplanar line, which extends from said SIW transition section 157 towards a millimeter wave system board. In this case, TP and BP are the only metal layers on the 2 sides of a single layer dielectric sheet, layer TP attached to the ridged waveguide embodiments 200 described in the present document. Suitable LCP materials are offered e.g. by ROGERS®,

ULTRALAM 3000 series, or Panasonic FELIOS flexible circuit board (R-F705T). Second, examples for ridged waveguide embodiments (200, 200.1, 200.2) are given with the condition of providing cost-effective, high precision, electrically conducting 3D bodies in a high volume, high throughput production process. A viable solution path for these objectives is metallized plastics technology. Piece- parts of high precision and reproducibility can be fabricated in high volume out of single- or multicavity hard steel molding tools. Dependent on the chosen assembly- and joining processes, sufficiently high temperature capability is required. The ease and cost of a metallization process as well as the reliability of metal layer adhesion also depends on the plastic material choice. Generally, high-performance semi crystalline thermoplastics like PPS (Polyphenylene

Sulfide, e.g. Forton PPS grades of Celanese), PEI (Polyetherimide, e.g. ULTEM grades of Sabic), LCP (Liquid Crystal Polymers, e.g. VECTRA grades of Celanese) are possible options. For low mold shrinkage and piece parts warpage high glass- fiber (30 - 50%) and optionally mineral filler content is preferred for all types. Metallisation paths include PVD (Physical Vapor Deposition) as first step or direct metal plating in aqueous media, dependent on the plastics chemistry. Second, copper and optional silver plating achieves the required high RF conductivity and thus sufficiently low loss. Further materials from the PA (Polyamide)

thermoplastics family exist, which are readily plate able but have certain restrictions with respect to peak temperatures applied in assembly- and joining processes for hybrid embodiments 100.

[000109] Figures 3A and 3B show another embodiment where a double ridged waveguide 200.1 is coupled to the patch 104 of the metal structure 102 on top of a substrate 101. Reference is made to the description of Figures 1A - IF as far as the basic elements of this embodiment are concerned. The present

embodiment comprises one protrusion 205 at the lower-most facet 204 of each of the two ridges 201. The gap GP is defined by the fact that the lower-most facets 204 are recessed relative to the level of the end facet 209. The respective recess 213 is visible in Fig. 3B. In Fig. 3B one can also see that the transformer 206 has a gap width (called WF1) which is smaller than the gap width (called WF2) above the transformer 206. For details refer to Fig. 1C. In contrast to the embodiment of Fig. 1A, not only the gap width is decreased in the transformer 206 section, but additionally its width is increased : WEI > WE. The widening of the ridges 201 is to some extent equivalent to the narrowing of the gap in respect of achieving the required lower characteristic impedance in the

transformer section 206. Thus, excessively narrow gap width WF1 can be avoided, which might be especially useful when using rapid prototyping

technologies for development samples. [000110] This embodiment, too, has two vias 108 which are situated right behind the slot 103. The arrangement and purpose is the same as in connection with the first embodiment.

[000111] It is not absolutely necessary to provide vias 105 for the definition of the SIW circuit 110. Instead of using vias 105, it is possible to employ a substrate which is structured or tailored so that the wave is appropriately confined in the SIW circuit 110. Such a structured substrate might comprise metallized side walls in order to provide for a short circuit along the boundary of the substrate. Such contiguous side walls can for instance be created by milling slots into the uppermost layer of a multilayer printed circuit board and

subsequent galvanic process to connect BP and TP SIW planes, instead of employing arrays of drilled via holes.

[000112] In Fig. 3A a substrate 101 is shown which has a wider first portion and a narrower second portion. The second portion sits underneath the metal patch 104. The wider first portion may carry the microstrip line 151 (not shown in Fig. 3A). The substrate 101 may be structured or tailored so that in fact it has a wider first portion and a narrower second portion, provided that the narrower second portion has metallized side walls, as mentioned above. Likewise, the substrate 101 may be structured or tailored by means of appropriately positioned vias 105 (like in Fig. 1A).

[000113] Figures 4A and 4B show another embodiment where a double ridged waveguide 200.1 is coupled to the patch 104 of the metal structure 102 on top of a substrate 101. Reference is made to the description of Figures 1A - IF, and 3A, 3B as far as the basic elements of this embodiment are concerned. The present embodiment comprises an H-shaped slot 103 which has a 90°-orientation (the narrow middle section of the slot 103 is perpendicular with respect to the x- axis), whereas in Fig. 3A the H-shaped slot 103 has a 45°-orientation. Please note that the H-shaped slot 103 of Fig. 3A has a less pronounced H-shape than the H-shaped slot 103 of Fig. 4A, 4B.

[000114] Fig. 4A, 4B further show details of the transformer 206 (represented by dashed lines in Fig. 4A). The respective transformer 206 has a gap width WF1 which is smaller than the standard width WF2 of the gap 207 of the double ridged waveguide 200.1. The gap width WF1 of the transformer 206 is chosen so that the narrow middle section of the slot 103 fits between the walls of the ridges 201 which define the transformer 206. The wideband operation of the wave transition requires adjustment of the slot 103 geometry for achieving a

resonance approximately in the centre of the useful frequency band, although in 90° orientation the longitudinal slot extension is more limited by the SIW side walls. This is possible by extending the length of its side-arms, making it more similar to the letter "H". Also in this configuration, the combination of gap GP between lower ridge end facets 204 and metal patch 104 and protrusions 205 allow proper operation of the embodiment despite in the projection onto the x-y plane, waveguide ridges and H-shaped slot 103 overlap.

[000115] A certain disadvantage of the embodiment shown in Fig. 4A and Fig.

4B is, that there are fewer choices of ridged waveguide orientations. Referring to Figs. 2A to 2C, an orientation of the ridges 201 parallel to the x-z plane (Fig. 2A) is possible, the -45° orientation shown in Fig. 2B as well (minor re-tuning of geometrical parameters might be necessary). A new possibility is to orient the ridged waveguide 200 alternatively in +45° orientation, without a change or re tuning w.r.t. -45° oriented situation. The cross-polarized orientation of Fig. 2C is not possible with the embodiment of Figs. 4A and 4B, nor does an option of phase reversal exist.

[000116] In Fig. 4B a dashed line Y2 shows the boundary of the structured or tailored substrate 101. If vias 105 are provided, then these vias 105 are arranged along the dashed line Y2.

[000117] Figures 5A, 5B, and 5C show another embodiment where a double ridged waveguide array 200.3 is coupled to a metal structure array 102.1 on top of a common back side metallization 120 (the back side metallization 120 is not shown in Fig. 5C). The array 102.1 comprises five planar structures 130, as illustrated in Fig. 5A. The double ridged wave guide array 200.3 comprises five double ridged waveguides integrated into one bulk element. The two waveguides on the left and right of the array 200.3 have a 90°-orientation (like in Fig. 2C). The three waveguides in the middle have a 0°-orientation (like in Fig. 2A). The H-shaped slots 103 of the five planar structures 130 have the following

orientation with respect to the x-axis (from left to right) : 45°; 45°; 45°; 45°;

-45°. The arrows A1 - A5 in Fig. 5A symbolize the possible propagation directions of the respective electromagnetic waves in an attached waveguide-based signal distribution layer. The latter can be composed of tracks of single- or double-ridge waveguides, while keeping the advantage of small waveguide array pitch (e.g. l/2) parallel to the x-y plane. Note, that the respective ridges oriented towards the assumed propagation directions A1 to A4 are connected to the slot 103 edges which lie on the backshort side of the feeding SIW sections, while the outward- oriented ridge associated with direction A5 is again connected to the slot side connected with the backshort, by virtue of a 90° pivoted slot 103 orientation. Thus, all propagation directions A1 to A5 provide identical electrical phases and signal delays.

[000118] The cross-section of Fig. 5C shows details of the double ridged waveguide array 200.3.

[000119] Each of the double ridged waveguides of the array 200.3 comprises a l/4 transformer 206, as illustrated in Fig. 5C. This l/4 transformer 206 basically is a section of the respective waveguides where the gap 207 between the ridges 201 is narrower, as discussed before. The embodiment of Figs. 5A to 5C is a combination of elements according to Figs. 4A and 4B. The additional widening of the ridges in the transformer 206 section can be clearly seen in Figs. 5B and 5C.

[000120] It goes without saying, that any number of transitions can be combined in linearly arrayed bulk embodiments, that wave propagation

directions A2, A3 and A4 can be individually reversed to exit the transition region e.g. in -i-x-axis direction, and A1 and A5 can be exchanged individually or both with directions parallel to A2, A3 or A4. It is also possible to arrange for 2 x n transitions, with the propagation directions all pointing away from the bulk transition arrangement.

[000121] Figures 6A, 6B show another embodiment where six individual double ridged waveguides 200.1 are inserted into bays/openings/receptacles 216 of a connector or socket 210 (hereinafter called frame). This frame 210 sits on top of a planar structure 130 which carries/comprises six metal structures 102. The frame 210 is precisely aligned with respect to the slots 103 of the metal structures 102. In this embodiment, all slots 103 have a 45°-orientation with respect to the x-axis. [000122] The individual double ridged waveguides 200.1 have a complementary structure so as to mechanically define their position relative to the position of the slots 103 when the individual waveguides 200.1 are inserted into the respective bays/openings/receptacles 216 of the frame 210.

[000123] Fig. 6B shows a detailed view of a short section of the frame 210 together with one double ridged waveguide 200.1 which was inserted into the respective bay 216 of the frame 210.

[000124] It can be derived from Fig. 6B that the double ridged waveguide 200.1 comprises two recessed end facets 204 and that each end facet 204 carries one contact protrusion 205, as discussed before. This embodiment comprises a l/4 transformer 206, too.

[000125] In at least some embodiments, the frame 210 comprises guiding structures to mechanically guide the individual waveguides 200.1 during their insertion, and keep them in place during operation. Fig. 6B further depicts a preferred position of short alignment ridges on the outer corners of the waveguide 200.1. They preferably have a tight fit inside accordingly arranged grooves of frame 210, preferably on opposite sides, while having a loose fit radially. A small air gap is provided between all other outside surfaces of waveguide 200.1 and their opposed counterparts on the frame 210 (inside surfaces). In this way, compression forces required for safe fixation balance locally in the outer ridged waveguide corners, and any deformation /

compression of the central ridge gap is avoided.

[000126] In at least some embodiments, the individual waveguides 200.1 are permanently fixed (e.g. using glue) inside the frame 210 during the assembly of the apparatus 100.

[000127] In at least some embodiments, the individual waveguides 200.1 are temporarily fixed inside the frame 210 during the assembly of the apparatus 100 (e.g. by using a tight fit mechanical insertion with plastic / elastic material deformation). If needed, e.g. for testing purposes, the individual waveguides 200.1 can be removed from the frame 210. [000128] In case of embodiments which comprise a frame 210 for one or more than one individual waveguide 200.1, a mechanical contact pressure is established between the protrusions 205 and the metal patch 104. Like with other embodiments, no glue or contact agent is required.

[000129] At least some of the embodiments comprise a double ridged

waveguide 200.1 which has a contact ring and/or contact protrusions 208 which are dimensionally stable. It is, however, also possible to provide a contact ring and/or contact protrusions 208 which have a certain elasticity. This is possible because the critical gap width GW is an assembly procedure definitely defined by the dimensions of the protrusions 205.

[000130] As mentioned before, some embodiments comprise a single ridged waveguide 200.2 instead of a double ridged waveguide 200.1. Respective embodiments are shown in Figures 7, 8A, 8B, 9, 10A, 10B, 11.

[000131] In case of a single ridged waveguide 200.2, a longitudinal conduit is provided which has a rectangular shape. It has one or two open ends 211 parallel to the x-z-plane and an open bottom parallel to the x-y-plane. The other four out of six planes of the rectangular shape are defined by side walls and by a top layer of the planar circuits 110 (and 150). In the center of the conduit there is a single ridge 201 which longitudinally extends parallel to the y-axis.

[000132] The single ridge 201 is designed so that it has at least one raised portion 201.1 and at least one recessed portion 201.2.

[000133] Fig. 8B shows the parallel air gap GP of the apparatus 100. In this constellation, where the waveguide 200.2 and the SIW circuit 110 are

assembled, the ridge 201 is hanging down, that is, the lower facets 204 of the raised segments 201.1 of the ridge 201 are facing the top plane TP of the SIW circuit 110. The parallel air gap GP is defined by the top plane TP on the one side and by the lower facets 204 on the other side. The positions of the gap 214 and the slot 103 are visible, too. The uppermost layer of the SIW circuit 110 is situated parallel to the x-y-plane. The raised portions 201.1 of the waveguide 200.2 provide for the gap GP (cf. Fig. 8B) with a precisely defined gap width Gw [000134] Fig. 10B shows the parallel air gap GP of another apparatus 100. In this constellation, where the waveguide 200.2 and the SIW circuit 110 are assembled, the ridge 201 is hanging down, that is, the lower facet 204 of the raised segment 201.1 of the ridge 201 is facing the top plane TP of the SIW circuit 110.

[000135] As mentioned before, at least some of the embodiments offer the option to rotate the waveguide 200 about the z-axis when assembling the apparatus 100.

[000136] Fig. 11 shows a top view of an exemplary single ridged waveguide apparatus 100. The ridged waveguide 200.2 which has an overall length Ly (parallel to the y-axis) in this constellation.

[000137] The position of the H-shaped slot 103 relative to the position of the waveguide 200.2 is shown. Like in other embodiments, the H-shaped slot 103 has a 45°-orientation with respect to the x-axis (i.e. b = 45°). The angle f indicates the orientation of the longitudinal axis of the single ridged waveguide 200.2 with respect to the x-axis. In Fig. 11, the angle cp=90°.

[000138] There are a number of other constellations/orientations possible, as will be summarized in the following. A useful angle range can be defined, as follows: 0° < f < 360°.

[000139] Preferred embodiments are defined by the following standard constellations/orientations: f = 0°, f = 90°, f =180°, and f = 270°. For these standard constellations/orientations, one design of the SIW circuit 110 and slot 103 fits all these four angles, if b = +45° or -45°. The f positions 90° and 270° are special in a sense, that a 180° electrical transmission phase change is obtained by switching the slot angle b between +45° and -45° positions.

[000140] At least some embodiments are defined by the following intermediate constellations/orientations: f = 45° and cp = 225°. For these two intermediate constellations/orientations, only slight modifications of the SIW circuit 110 and slot 103 are required, if b = +45°. [000141] At least some embodiments are defined by the following intermediate constellations/orientations: f = 135° and f = 315°. For these two intermediate constellations/orientations, only slight modifications of the SIW circuit 110 and slot 103 are required, if b = -45°.

[000142] Reference numbers