BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention pertains to the field of antennas and more particularly to antenna configurations and signal processing of antenna signals for beam narrowing. 2 . Description of the Prior Art

Marine radar antennas are required by the International Ξlectrotechnical Committee (IEC) specifications 936 and 872 to rotate at 20 RPM. Revisions of IEC 936 contain requirements for antennas en High Speed Craft to rotate at 40 RPM while maintaining the azimuth resolution requirements imposed on the 20 RPM antennas. The increased vessel speed and antenna rotation rate requirement, significantly increases the wind loading on the motors that rotationally drive the antenna. Mechanical solutions to this problem have been to enclose the antenna in a radome, thereby shielding it from the wind so that the load on the drive motor is that provided by the antenna alone, placing wings on the antenna to provide better aerodynamic performance, or to provide a more powerful drive motor. All these solutions add significant cost to the radar system. It is well known that the radiation beamwidth of an antenna, which determines the angular resolution of the radar system, is a function cf its size. Consequently, reducing the sail area of the antenna, while maintaining conventional antenna design and signal processing techniques, increases the radiation beamwidth, -hereby adversely effecting the angular resolution of the radar system.

One technique in the prior art for decreasing the beam width of an antenna of a given size, thereby increasing the angular resolution, utilizes sum and difference signals such as those provided in a monopulse antenna system. Two antennas of equal size and configuration and having independent signal output ports, are placed side-by-side. The signals at the output ports at each half of the antenna are summed and differenced to provide sum and difference signals, the peak of the sum pattern being angular aligned with the null of the difference pattern, thus defining the boresight of the antenna. The sum is divided by the difference, which is a very low signal at boresight. Division

by this low signal provides a spike at boresight. As the antenna is rotated from the target position the sum signal decreases while the difference signal increases. This respective decrease and increase of the sum and difference signals causes the ratio of the sum to difference signal pattern to decrease rapidly. Thus, the radiation pattern of the antenna is narrowed and the target position is more accurately determined.

Though processing the signal returns by dividing the sum signal by the difference signal improves target position accuracy in a single target environment, in a multiple target environment, however, this type of processing may mask true targets and give rise to ghost targets. For example, when two targets having equal radar cross sections are with in the same range bin and are positioned at equal angles on opposite sides of the boresight line, the signal at the output of the sum channel will be that of a target at the antenna's boresight having a radar cross section that is twice that of each target weighted by the level of the antenna sum pattern at the angular position of the targets. Concomitantly, the signal at the output of the difference channel will be a deep null. This results in a ratio of the sum channel output to the difference channel output that is near infinite, a strong indication of a target where there actually is no target, i.e. a ghost target indication.

Another monopulse technique utilized in the prior art for beam narrowing derives a sum and difference signal, as described above, and, instead of dividing the sum signal by tne difference signal, subtracts the magnitude of difference signal from the sum signal. Since the difference signal is at a null on boresight it has little effect on the peak of the sum signal. At angles off boresight, however, the difference signal becomes significant and its subtraction from the sum signal appreciably reduces the composite signal. If the difference pattern has levels at the angular positions of the 3dB points of the sum pattern that are equal to these 3dB levels, the composite signal is zero at these angular positions. When these conditions are met the beam width of the composite antenna pattern is effectively one half that of the sum pattern beam width.

Monopulse subtraction processing provides a somewhat better multi-target performance than monopulse division processing. In

monopulse subtraction processing, two targets within the difference beam, at equal angles on either side of the boresight line, add to provide a signal that appears to be on the boresight line. When this signal is subtracted from the sum beam signal, which is the sum of the two target signals in the sum beam, a null, which is a function of the angular position of the two targets, appears on boresight at the composite signal output. Thus, the ghost target at the boresight midway between the two targets is eliminated. When the antenna is scanned from its target balanced position, the two targets become asymmetrically positioned relative to the antenna boresight line, one providing a greater return in the sum pattern than the other. When the peak of the sum beam is at the one target position a deep null appears in the difference pattern, thus permitting the resolution of the two targets.

The composite signal output of monopulse subtraction processing, however, exhibits relatively high sidelobe levels. This is due to the fact that the level of the difference pattern lobes in the sidelobe region of the sum beam is appreciably greater than the sidelobe level of the sum beam. Consequently, subtracting the much greater signal of the difference pattern from the sidelobe of the sum pattern creates an undesirable high sidelobe level for the composite signal pattern, which may adversely effect target location determination.

SUMMARY OF THE INVENTION

It is an object of this invention to provide an antenna and signal processor which has low sidelobes and improves the target resolution and position determination accuracy over that obtainable from a conventional antenna of equal size.

In accordance with the principles of the invention, an antenna of predetermined length is divided into three sections, two sections are combined as an interferometer and the third is a continuous antenna which has a main lobe that is broader than the lobes of the interferometer. The antenna has two modes of operation, transmit and receive. In the transmit mode, the three sections are combined to form a single antenna having a radiation pattern that is determined by the illumination function, which

is chosen to provide a desired sidelobe level for the transmit beam. In the receive mode, the continuous antenna section, which is illuminated by a portion of the illumination function, forms an antenna that is apart from the two sections that form the interferometer. The sections forming the interferometer are illuminated by the remainder of the illumination function. Output signals of the interferometer are added to and subtracted from the output signal of the center section. These sum and difference signals are coupled to a video filter wherein they are processed to provide a narrow beamwidth low sidelobe antenna pattern.

The invention will be more clearly understood from the following description of an embodiment thereof with reference to the accompanying drawings. It should be understood that the embodiment described is an example of the invention and other embodiments may be implemented within the purview of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

Figure 1 is a block diagram of a preferred embodiment of the invention.

Figure 2 a graph of an amplitude distribution which may be applied to the elements of the antenna of Figure 1.

Figure 3A is a radiation pattern for the antenna of Figure 1 in the transmit mode when the amplitude distribution shown in Figure 2 is applied.

Figure 3B is a radiation pattern of a central section of the antenna of Figure 1 when the central section of the amplitude distribution of Figure 2 is applied. Figure 3C is a radiation pattern of the interferometer formed by end sections of the antenna of Figure 1 when the end sections cf the amplitude distribution shown in Figure 2 is applied.

Figure 4 shows radiation patterns established by adding the interferometer pattern of Figure 30 to the central section pattern of Figure 3B and by subtracting the interferometer pattern from the central section pattern.

Figure 5 is a functional block diagram of the video filter of Figure 1.

Figures 6 - 9 are patterns at various stages in the video filter of Figure 1.

Figure 10 is a radiation pattern of the antenna system provided at the video filter output terminal. Figure 11 is a two way radiation pattern of the central and end sections of Figure 1 operating as a continuous antenna.

Figure 12 is a two way radiation pattern at the video filter output terminals for a two target operating environment.

Figure 13 is a two way radiation pattern of a conventional antenna for a two target operating environment.

DESCRIPTION OF THE PREFERRED EMBODIMENT

Refer now to Figure 1. A segmented antenna system may include an antenna 11 which is of length S. The antenna 11 may be divided into three sections, a center section 13 of length S/2 and left 15a and right 15b edge sections, each of length S/4. Left edge 15a and the right edge 15b are combined in a manner to form an interferometer having a base line that is equal to 3S/4. Interferometer 15a, 15b and the center section 13 may be coupled to a magic Tee 17, or other hybrid device, which provides the sum Σ and difference Δ of the interferometer and center section signals. The sum signal output port 17a of the agic Tee 17 may be coupled via a circulator 19 to a receiver 21. The difference signal output port 17b of the magic Tee 17 may be coupled directly to the receiver 21. Since the interferometer 13a, 13b has a base line that is longer than the length of the central section 13, it produces an antenna pattern which is narrower than the antenna pattern produced by the central section. Consequently, the interferometer 15a, 15b has a higher resolution capability than the central section. Due the higher resolution capability of the interferometer 15a, 15b, in the description to follow, its output signal will be designated H, high resolution, and the central section 13 output signal will be designated L, low resolution. Signals representative of the sum Σ and difference Δ signals are coupled from the receiver 21 to a power divider and hybrid circuit 22 wherein signals representative of the high H and the low L signals are derived from H = (Σ - Δ)/2 and L = (Σ + Δ)/2. These high H and low L representative

signals are coupieα to a video filter 24, to which signals representative of the sum Σ and difference Δ signals are also coupled. A transmitter 23, for providing radiation signals, may be respectively coupled to the interferometer 15a, 15b and tne central section 13 of the antenna 11 via the circulator 19 and the sum signal port of the magic Tee 17. Since the transmitter feeds both tne central section 13 and the interferometer 15a, 15b simultaneously it provides the radiation signal for the ent _i re antenna 11. F ⁱ gure 2 ⁱ s an illustration of an antenna illumination function 25 that may be applied over the entire aperture of the antenna 11. The illustrative function 25 is a standard Taylor taper naving defining parameters n = 2 and si _d elobe level ₍ si ₎ = 16dB. It snould be unαerstood that this distribution function ⁱ s not limiting and that other ll ummatior functions may be used. Since the distribution is over the entire aperture, _t ne left 15a and right 15b edge sections have distribution functions 25a and 25b, respectively, while the central portion 13 has the distrioutior function 25c. For S equal to 22.5λ ₍ λ _i s the operating wavelength of tne antenna), the aperture distribut _i on funct ⁱ on of Figure 2 gives rise to the antenna patterns shown _i n F ⁱ gures 3 - 3C. Figure 3A s the transmit radiation pattern T of the entire aperture of antenna 11; Figure 3B is the rece _i ve pattern L for the central section 13; ana Figure 3C is the receive pattern H for the interferometer 15a, 15b. The sum Σ = L + H and d ⁱ fference Δ = 1 -H patterns at the output of tne mag _i c Tee 17 are snown in Figure 4.

Tne four signals ⁽ L, b , Σ, and Δ) are couoled to the v _i oec f ⁱ lter 24 wnerein tne signals are coup ed tnrough logarithm _i c amplifiers to a processor such as that shown m Figure 5. i- constant A, is added to log(Δ) m summer 27a. This sum _i s subtracteα from log ⁽ ∑ ⁾ m summer 27b and tne difference _i s mult ⁱ plied oy g4 in amplifier 29a. Suostraction of logarithms ⁱ s the equivalent of dividing the antilog of tne minuend b _" tne antilog cf tne subtranenc. Tnough the constant A, is cnosen tc contrc_ tne sidelobe level, this division may gives rise _t c sp ⁱ kes in regions where tne Δ pattern approaches zero. Spικe levels may ce controlled cy coupling tne outc: terminal of tne

amplifier 29a to a iimiter 3ia having an output terminal 32a at which a signal representative of log[Σ/(a ₁ Δ)] ^g4 is provided, where a _x is the antilog of A-, . With A _x = 0 (a _x = 1) and g4 = 1, the pattern at the output terminal 32a of Iimiter 31a, as a function cf antenna azimuth angle from a fixed target, is given in Figure 6.

Output terminal 32a of the iimiter 31a is coupled to summer 27c, wherein the signal at the output terminal 32a is added to a signal coupled tc the summer 27c from the output terminal 32b of summer 27d. The signal coupled from the output terminal 32b of summer 27d is the sum of log(∑) multiplied by g2 in amplifier 29b plus iog(H) multiplied by g3 in amplifier 29c, the product oeing added to A ₂ in summer 27e prior to being coupled to summer 27d. The signals at the output terminal 32b of summer 27d and the output terminal 32c of summer 27c are respectively representative of log (a ₂ Σ ^g2 H ^g3 ) and log {a ₂ H ^< 3 ³ [Σ ^g2+ ^V ( _aι Δ) ^g4 ] } , where a. is the antilog cf A ₂ . Patterns at the terminals 32b and 32c, . when A ₂ =-18, g2=0.5, and g3=0.5, are respectively shown in Figures 7 and 8. Output terminal 32c of summer 27c is coupled to summer 27f wherein a constant A3 is added. The signal at the output terminal cf summer 27f is multiplied by g5 in amplifier 29d and coupled tc summer 27g wherein Log(L), multiplied by gl in amplifier 29e, is subtracted therefrom. The difference signal at the output terminal cf summer 2 ^" ! q is multiplied by g6 in amplifier 29f and coupled to iimiter 31b having an output terminal 32d at which a signal is provided which is representative of

log{ [ (a ₂ a ₃ ) ^g5q6 /a ₁ ^g4g5g6 ] [ _H ^g3( 3 ^5< ? ⁶ /L ^glg6 ] [∑ <g2 ⁺ g ⁴⁾ g5 ^g 6 _Δ g ⁴ g5g6 _{] }}

A pattern for this signal is given in Figure 9, when A _x =0, A ₂ =- 18, g2=g3=2.5, g4=g5=g6=l.

The output terminal 32d of Iimiter 31b is coupled to summer 27h to which the output terminal 32c of summer 27c and a constant

A ₄ are also coupled. The three signals coupled to summer 27h are added to provide a signal at the output terminal 32e which is representative of

log [ C ( H ^{n l} /L ⁿ² ) ( Σ ^{n 3} /Δ ^{n 4} ) ] .

Thus the sharpened antenna pattern is given by :

C ( H ^nl /L ⁿ² ) ( Σ ⁿ³ /Δ ⁿ⁴ )

where :

C = ( a ₂ a ₃ ) ^ ^{g 6} a ₂ a ₄ /a ₁ ^{g4 ( 1} ^ ^{6 )} nl = g3 ( l - g5g6 ) n2 = gig 6 n3 = (g2 + g4) (1 + g5g6) n4 = g4 (1 - g5g6)

The output pattern of the video filter, the desired antenna pattern, when A ₁ =A^=0, A2--18, A3=6, gl=g4=g5=g6=l, and g2=g3=0.5 is shown in Figure 10. The beamwidth of the antenna pattern is approximately 1.40.

All of the above patterns are two way patterns, i.e. they are the result of transmitting the pattern shown in Figure 3A and processing the received and H patterns of Figures 3B and 3C, respectively. As stated above, the antenna has an overall length of 22.5λ and a Taylor distribution with n = 2 and si = 16dB over the entire antenna. A conventional antenna having a length of 22.5λ and a Taylor distribution with n - 2 and si = 16dB will have an antenna pattern as shown in Figure 11. It is apparent from these Figures that a significant reduction in the beamwidth and sidelobe levels has been achieved with the segmented antenna and the video filter.

The above discussion has considered but one target. In a multiple target environment the beam narrowing realized with the segmented antenna II and the signal processing performed by the video filter 24 provides significant improvements in target resolution over that achievable with a conventional antenna of equal size without generating ghost targets. A two way pattern generated by the segmented antenna 11 and video filter 24 combination is shown in Figure 12. The targets are of equal radar cross-section, are at equal ranges, and are in Rayleigh distributee noise with an angular separation of 2.5o. The video filter processing parameters for the pattern of Figure 12 are:

--,=8; A ₂ =l; A ₃ =A ₄ =0; gi=g4=g5=g6=i; g2=g3=0.5. Two discernible pattern peaks 35a and 35b, respectively representing target detections, are clearly indicated in Figure 12. These pattern peaks are separated by 2.5° and positioned 1.250 on either side of a deep null 37, no ghost targets are discernible. A conventional antenna having a length equal to the length of the segmented antenna 11 ano a Taylor aperture taper with n=2 and sl=16dB would have a beam width of approximately 2.80. This conventional antenna would not be able to resolve two targets with 2.50 of angular separation.

A comparison cf the two target pattern generated by the segmented antenna 11 with the video filter 24 processor to a two target pattern, shown in Figure 13, generated by a conventional antenna having a length twice that of the segmented antenna, indicates the superior performance of the segmented antenna system. The target indication peaks 35a and 35b in Figure 12 are substantially at the same angular positions as the target indication peaks 39a and 39b in Figure 13. The sidelobes 41 in Figure 12 have substantially the same peak levels as the sidelobes 43 in Figure 12. The null depth 37 between targets in Figure 12, however, is substantially greater than the null depth 45 between targets for the conventional antenna pattern cf Figure 13.

While only certain embodiments of the invention have been described, it will be apparent to those skilled in the art that various changes and modifications may be made within the purview of the appended claims without departing from the true scope and spirit of the invention in its broader aspects.