The present invention refers to a circuit of a voltage controlled oscillator (VCO).

More specifically, the invention refers to a circuit of a voltage controlled oscillator including a varactor, as defined in the preamble of claim 1.

In modern digital wireless telecommunication systems, voltage controlled oscillators (VCO) represent a fundamental part for the synthesis of a carrier frequency, thanks to their use in PLL (Phase Locked Loop).

In this type of application, the design of the oscillators is made particularly difficult by the high frequency of the synthesised signal and by the constraints imposed by power consumption.

The traditional topology with LC resonators represents the solution used the most to satisfy the specifications of spectral purity and power consumption imposed by current communication standards.

However, there are different reasons that push for innovative topologies to be sought:

- the use of CMOS technologies with an ever smaller channel length requiring the use of ever lower supply voltage;

- the constraint of a low power consumption imposed by low data rate type applications for wireless sensor networks;

- the need of designing oscillator circuits with wide tuning range for broad band wireless systems.

Figure 1 shows a circuit 1 of a broad band oscillator according to a first prior art.

The circuit 1 comprises, in a per se known way, a bridge structure including two cross- coupled MOS transistors M ₃ and M4 of the N type and two cross-coupled MOS transistor M ₅ and M ₆ of the P type. The circuit 1 also comprises an LC resonator 2 placed in parallel between said pairs of transistors M ₃ and M ₄ of the N type and M ₅ and M ₆ of the P type _. The transistors M ₃ , M ₄ , M ₅ and M ₆ overall represent the active part of the circuit 1 that has the task of compensating for the losses of the LC resonator 2.

A per se known current mirror is indicated with 3, said mirror comprising a pair of transistors Mi and M ₂ to which a supply voltage V _DD is applied. The current mirror 3 generates a current I _B used to bias the circuit 1.

The same supply voltage V _D D is applied to the two transistors M ₅ and M ₆ of the P type through an associated voltage source. Alternatively, the supply voltage of the current mirror 3 is different from the supply voltage of the two transistors M ₅ and M ₆ of the P type.

The LC resonator 2 comprises an inductor L _D and two varactors Cy supplied by a control

Voltage Vtune.

The circuit 1 uses the two varactors Cy to control the oscillation frequency, which is equal to the resonant frequency of the LC resonator 2, in a continuous manner by modulating the control voltage V _tun e.

In order to increase the variation range of the oscillation frequency, and thus to obtain a broad band oscillator, the circuit 1 includes a first set 4a of capacitors C _S W _I ,...,C _S WN and a second set 4b of capacitors C _SWI ,- - - ₅ CSW _N , respectively identical to the capacitors Cswi ,... ,C _SWN of the first set 4a. Each capacitor CSWI,... ,CSWN, of the first set 4a and of the second set 4b, respectively, is connected to a respective switch Mswi ,...,M _S WN obtained with a MOS transistor of the N type controlled by an associated gate voltage Bi ... BN-

The capacitors C _S WI _? - - -,C _S W of the first set 4a and of the second set 4b make it possible to obtain a discrete variation of the oscillation frequency.

The main advantage of such a solution consists in being able to obtain broad band oscillators without needing to use a varactor with high values of the figure of merit K _v , where the figure of merit K _v is defined as follows:

AC ,.

* ^■ A " V ^Y tune

where AC is the incremental variation of the capacity of the varactor with a AVtune variation of the control voltage.

Indeed, high values of K _v jeopardise the performance of phase noise due to the known phenomenon of noise conversion AM-PM.

The main drawbacks of such a solution are the following:

- the phase noise is not optimised in the different oscillation frequency sub-bands identified by the capacitors CSWI-CSWN of the first set 4a and of the second set 4b of capacitors;

- there is an oversize in terms of current consumption by the circuit 1 (and thus power consumption); such an oversize is necessary in order to ensure a reliable oscillation start condition in the lower band limit, i.e. with a large capacity of the varactors Cv and an associated low figure of merit K _v .

Figure 2 shows a circuit 1 of a broad band oscillator according to a second prior art, in which similar elements have been given the same numbers as in figure 1.

In this circuit 1 a varactor structure Cv with high figure of merit K _v is used.

In order to eliminate the problem relative to the degradation of the phase noise due to the AM-PM effect, each varactor C _v is subdivided into N parts connected in parallel and each biased with an associated biasing voltage VBI , .. .,VBN.

The resistors RB and the capacitors Co represent the biasing network of the varactors Cv- The capacitors Co are used to decouple in DC the varactors Cv from the drain nodes of the transistors M ₃ , M ₄ , M ₅ and M ₆ . The resistors RB are biasing resistors used so as to prevent the varactors CQ from short circuiting. The circuit 1 makes it possible to linearize the figure of merit Kv and to minimise the maximum value of its derivative with respect to the control voltage V _tun e for the same overall variation of the capacity of the varactors Cv.

However, such a solution leads to an increase of the phase noise.

Figure 3 shows a circuit 1 of a broad band oscillator according to a third prior art, in which similar elements have been given the same numbers as in figures 1 and 2.

In this circuit 1 the two varactors Cv and four switched inductors LDI and LQ ₂ are used. The discrete variation of the oscillation frequency is therefore carried out by varying both the inductive and the capacitive components of the LC resonator 2.

In the circuit 1 of figure 3, the first set 4a and the second set 4b of capacitors comprises a single capacitor Cswi connected to a respective switch Mswi made by a MOS transistor of the N type controlled through a gate voltage Bj.

The circuit 1 of figure 3 makes it possible to overcome the limits of the previously described circuits as far as the optimization of the phase noise and power consumption is concerned.

As mentioned above, the LC resonator 2 comprises four inductors, a first inductor LDI and a second inductor L _D2 arranged on a first branch 5 of the LC resonator 2 and a third inductor L _D i and a fourth inductor LD ₂ arranged on a second branch 6 of the LC resonator 2, respectively. The first branch 5 and the second branch 6 of the LC resonator 2 are connected to each other through a switch Msw obtained with a MOS transistor of the N type which, however, jeopardises the quality factor of the first and of the third inductor LQI .

These are what cause there to be limits in terms of phase noise or excessive power consumption due to the reduction of the Q factor of the LC resonator 2.

Such limits can be overcome by using a switch Msw with low resistance and with great W/L ratio. This however, reduces the tuning range.

Figure 4 shows a circuit 1 of a broad band oscillator according to a fourth prior art, in which similar elements have been given the same numbers as in figures 1 , 2 and 3.

In this circuit 1 an LC resonator 2 is used based on switched inductors in a parallel configuration.

Such a circuit 1 comprises two differential inductors Li mutually coupled by means of a mutual inductance coefficient M and a switch M _sw inserted in series with the inductor L ₂ .

Such an approach has several advantages with respect to the previous solutions since the parallel configuration makes it possible not to degrade the quality factor of the equivalent parallel inductance due to the resistance of the switch Msw- Such a configuration makes it possible to optimise the phase noise at the different oscillation frequencies and to obtain reliable start conditions avoiding oversizing of the circuit in terms of consumption and power.

The connection of the switch Msw however, is not optimised since the biasing voltage of the drain (source) of the MOS transistor which implements the switch Msw is equal to VDD-VGSPMOS where VQSPMOS is the gate-source voltage of the transistor Msw-

Consequently, the gate-source voltage in switching on conditions is equal to VQSPMOS, and therefore the triode resistance, since the W/L ratio is fixed, is not the minimum which can potentially be reached.

Such a problem can be eliminated by using complementary switches but this leads to a substantial increase of the parasitic capacities thereof.

The purpose of the present invention is therefore of proposing a circuit of a voltage controlled oscillator that has a better performance and a lower power consumption with respect to the aforementioned solutions of the prior art. These and other purposes are achieved with a circuit of a voltage controlled oscillator the characteristics of which are defined in claim 1.

Particular embodiments are the subject of the dependent claims, the content of which should be considered as an integral and integrating part of the present description.

Further characteristics and advantages of the invention shall become clearer from the following detailed description, given purely by way of an example and not for limiting purposes, with reference to the attached drawings in which:

- figure 1 , already described, is a schematic diagram of a circuit of a voltage controlled oscillator according to a first prior art;

- figure 2, already described, is a schematic diagram of a circuit of a voltage controlled oscillator according to a second prior art;

- figure 3, already described, is a schematic diagram of a circuit of a voltage controlled oscillator according to a third prior art;

- figure 4, already described, is a schematic diagram of a circuit of a voltage controlled oscillator according to a fourth prior art;

- figure 5 is a schematic diagram of a circuit of a voltage controlled oscillator according to the present invention;

- figure 6a is a top schematic view of an exemplary embodiment of a layout of a resonator of the circuit according to the present invention; and

- figure 6b is a schematic diagram of the resonator of figure 6a.

Figure 5 illustrates a circuit of a voltage controlled oscillator according to the present invention in which similar elements have been given the same reference numerals as in figures 1 -4 previously described.

The circuit 1 comprises a bridge structure including two cross-coupled MOS transistors M ₃ and M ₄ of the N type and two cross-coupled MOS transistors M ₅ and M ₆ of the P type.

The circuit 1 also comprises an LC resonator 2 placed in parallel between said pairs of transistors M ₃ and M ₄ of the N type and M ₅ and M ₆ of the P type _. The transistors M ₃ , M4, M ₅ and M ₆ overall represent the active part of the circuit 1 that has the task of compensating for the losses of the LC resonator 2.

The two transistors M ₃ and M ₄ are connected to a current mirror 3 comprising a pair of transistors M| and M ₂ to which a supply voltage V _D D is applied. The current mirror 3 generates a current I _B used for biasing the circuit 1.

The same supply voltage VD _D is applied to the two transistors M ₅ and M ₆ of the P type through an associated voltage source. Alternatively, the supply voltage of the current mirror 3 is different from the supply voltage of the two transistors M ₅ and M ₆ of the P type.

The LC resonator circuit 2 includes two pairs of differential inductors Li and L ₂ mutually coupled by means of a mutual inductance coefficient M, each pair comprising a first inductor Li arranged on a respective branch 10a of an external loop, and a second inductor L ₂ arranged on a respective branch 12a of an internal loop.

The two first inductors Li have a common terminal.

The two second inductors L ₂ are connected to each other through a switch Msw obtained with a MOS transistor of the N type biased with a gate voltage B ₀ . The switch Msw connects or disconnects the second inductors L ₂ , in parallel to the first inductors

The branches 12a of the internal loop are connected to each other through a first and a second varactor Cv33 subjected to a control voltage V _tun e applied to a common node A.

The branches 10a of the external loop are connected to the branches 12a of the internal loop through respective de-coupling capacitors Co arranged to the decouple in DC the two second inductors L ₂ and the varactors Cy ₃₃ from the drain nodes of the transistors M ₃ , M ₄ , M ₅ and M ₆ .

The control voltage V _tun e is also connected to a third and a fourth varactor Cyi2- The branches 12a of the internal loop are connected to a ground GND through two biasing resistances R _B so as to bias the switch Msw, the first and the second varactor C _V 33 through the ground voltage.

The set of de-coupling capacitors C _D and of biasing resistors RB make it possible to bias the switch Msw so as to ensure a low loss.

The linearization of the varactors Cvi2 and Cv ₃₃ of the LC resonator 2 is carried out by obtaining said varactors C _V i2 and C _V 33 with MOS accumulation transistors with different thicknesses of oxide, said oxide thickness varying between the varactors C _V | ₂ and the varactor Cv33.

The topology of the linearized varactor therefore does not need alternative biasing points with respect to those already available in the circuit.

The operating principle of the circuit of the present invention, which is analogous to the operation of the circuits described above with reference to figures 1-4, shall now be described.

The bridge structure including the two cross-coupled MOS transistors M ₃ and M ₄ of the N type and the two cross-coupled MOS transistors M ₅ and M ₆ of the P type implements a negative resistance. Such a negative resistance has the task of compensating for the loss resistance of the LC resonator 2 so as to maintain an oscillation with constant width across said resonator 2. The oscillation frequency is determined by the following expression: where L _eq and C _eq represent the overall equivalent inductance and the overall equivalent capacity of the circuit LC resonator 2, respectively. In order to vary the oscillation frequency in a continuous manner, the C _eq is varied by modifying the value of the control voltage Vtune value applied to the common node A of the varactors Cv i2 and Cv33- The control voltage V _tune takes on continuous values comprised between the ground voltage and the supply voltage.

In order to vary the oscillation frequency in a discrete manner the two pairs of differential inductors Li and L ₂ which are mutually coupled by means of a mutual inductance coefficient M are used. The inductance L _eq of the LC resonator 2 is varied in a discrete manner by activating or deactivating, in parallel to the first inductor Li, the second inductor L ₂ by using the switch Msw- For this purpose the switch Msw is subjected to a gate voltage B ₀ that takes on two discrete values (ground voltage or supply voltage VDD)-

Figure 6a illustrates a top schematic view of a layout of the resonator 2 in which the first inductors L| and the second inductors L ₂ are obtained through two respective concentric loops of conductive material made on a silicon substrate. Figure 6b shows the diagram of the resonator 2 in which the various terminals a-f are the same as those indicated in figure 6a.

Such a layout has specifically been designed so as to maximise the mutual inductance between the first inductors Li and the second inductors L ₂ simultaneously reducing the area of silicon they occupy.

The advantages of the circuit of a voltage controlled oscillator according to the present invention are:

- optimisation of the phase noise over a very high range of frequencies thanks to the topology with switched inductors and the used varactor structure;

- improvement of the tuning range with respect to the prior art thanks to the lower parasitic capacities;

- improved trade-off between the phase noise and the tuning range thanks to the used varactor structure;

- minimisation of the silicon area of the switched inductor structure thanks to the use of an appropriate layout;

- reduction of the power consumption of the voltage controlled oscillator thanks to the switched inductor structure. Of course, without affecting the principle of the invention, the embodiments and the manufacture details can be widely varied with respect to what has been described and illustrated purely as a non limiting example, without for this reason departing from the scope of protection of the invention as defined in the attached claims.