[001] The present invention relates to high voltage direct current (HVDC) transmission systems for high power DC grids. Typically such systems use a line commutated converter (LCC) or a voltage source converter (VSC).

[002] Most existing DC transmission systems are based on current source thyristor converter technology because thyristor devices have low losses and are available in robust high current capacity single wafer capsules. However, thyristors inject significant low frequency harmonics, which must be eliminated by large passive filtering, cannot decouple the real and reactive power injected into the network, and require large passive components leading to large footprint systems.

[003] HVDC transmission systems based on a voltage source converter were developed to address these shortcomings. VSC technology is the most suitable topology for multi-terminal HVDC and DC grids because active power reversal is achieved without DC link voltage polarity change, and resiliency to AC side faults (no risk of commutation failure as with line-commutating HVDC systems). However, vulnerability to DC side faults and an absence of reliable high voltage DC circuit breakers restrict their application to point-to-point connection.

[004] There would be significant operational and cost benefits if the existing point- to-point lines could be interconnected or tapped on the DC side, hence creating multi- terminal DC networks. However, many technical challenges face the implementation of meshed DC grids. Major problems include the interface of current source and voltage source HVDC systems (especially in power reversal), lack of appropriate DC circuit breakers, power flow control difficulty in meshed grids and interconnecting DC transmission lines with different DC voltage levels.

[005] Common practice with known DC/DC converters for VSC is to trip the switches after a DC fault. However, this has two main drawbacks. Firstly, transient overcurrents resulting from tripping delays causes semiconductor overheating and additional cooling requirements are necessary before continued operation after recovery from a fault. Secondly, there is a need to re-synchronize the DC/DC converter with the AC grid after clearance of the fault.

[006] There is therefore a need for a universal DC/DC converter which avoids the aforementioned drawbacks.

[007] According to the present invention there is provided a DC/DC converter comprising :- a first bridge for connection via an inductor to a first DC power source;

a second bridge for connection via an inductor to a second DC power source; an AC transformer connecting the first and second bridges;

wherein the first bridge comprises :- a first branch having a pair of bi-directional switches, each including at least one IGBT semiconductor element, connected across the first power source and to a first terminal of the primary winding of the AC transformer;

a second branch having a pair of bi-directional switches, each including at least one IGBT semiconductor element, connected across the first power source and to a second terminal of the primary winding of the AC transformer; and

wherein the second bridge comprises :- a first branch having a pair of bi-directional switches, each including at least one IGBT semiconductor element, connected across the second power source and to a first terminal of the secondary winding of the AC transformer;

a second branch having a pair of bi-directional switches, each including at least one IGBT semiconductor element, connected across the second power source and to a second terminal of the secondary winding of the AC transformer; and

control means for providing square wave control signals to the semiconductor elements to enable a switching sequence of the semiconductor elements to provide differing current paths according to the type of first power source and second power source and direction of power transfer. [008] The DC/DC converter of the present invention encompasses the advantages of existing DC/DC converters and eliminates their drawbacks. In particular, the converter is suitable for high voltage stepping ratios. This is due to the possibility of zero local reactive power operation at the converter bridges at rated power due to AC capacitors designed to compensate the inductive reactive power absorbed by the AC inductors and/or transformer. Zero reactive power operation at rated active power also enables minimization of conduction currents, parasitic element losses and zero current switching. This maximizes efficiency and reduces IGBT current rating.

[009] Moreover the DC/DC converter of the present invention is suitable for the interfacing of conventional current source LCC HVDC with the growing VSC HVDC technologies facilitating active power reversal between both. This is crucial for countries like China where the majority of existing HVDC systems are yet of the LCC type. The converter can also enable DC power flow regulation in meshed DC grid transmission lines.

[0010] The converter of the present invention also inherently and naturally possesses open circuit characteristics in response to DC side faults. This means no control action is necessary by blocking/tripping active semiconductor switches to protect the converter from overcurrent. In particular, the converter current naturally drops to zero at the non- faulty side following a DC fault, whereas it can remain at 1 p.u at the faulty side of the converter with a peak transient fault current of less than 2 p.u which is tolerable by most IGBTs. This eliminates any need to trip active switches at both bridges and no grid resynchronization is needed.

[0011] The DC/DC converter of the present invention uses self-commutated IGBT technology which has several advantages in spite of its higher losses, particularly with regard to known converters using thyristors. Firstly, thyristors are much slower devices, hence this imposes a limit on maximum frequency which in turn increases the footprint of the passive elements. Secondly, harmonic generation is generally higher with thyristors which causes higher reactive power consumption and lower efficiency. [0012] Finally, the DC/DC converter of the present invention has galvanic isolation by magnetic- or mutual air-core coupling of the AC inductors.

[0013] In one embodiment, the control means varies the frequency of the control signals to provide control of power transfer. when interconnecting two VSC-HVCD systems.

[0014] In another embodiment, the control means used fixed frequency phase-shift square-wave control to provide control of power transfer when interconnecting LCC and a VSC-HVDC systems.

[0015] In a preferred embodiment, the AC transformer has an air-core mutually coupling the primary winding and secondary winding.

[0016] In another preferred embodiment, AC capacitors are connected in parallel with the primary winding and the secondary winding of the AC transformer.

[0017] In one embodiment, one or more said bi-directional switches comprises a pair of series connected IGBT semiconductor element, each having a diode in parallel.

[0018] Conveniently, every bi-directional switch comprises a pair of series connected IGBT semiconductor element, each having a diode in parallel. one.

[0019] In another embodiment, one or more said bi-directional switches comprises a diamond configuration of diodes with one IGBT semiconductor element connecting opposing apexes of the configuration.

[0020] Conveniently, every bi-directional switch comprises said diamond configuration of diodes with one IGBT semiconductor element connecting opposing apexes of the configuration

[0021] An example of the present invention will now be described with reference to the accompanying drawings, in which: -

Figure 1 illustrates a DC/DC converter of a first embodiment of the present invention; Figures 2A-2H illustrate current flows for different switching of the IGBT semiconductor elements in the bridges of the converter shown in figure 1 when used for connecting a voltage source converter and a voltage source converter;

Figures 3A-3H illustrate current flows for different switching of the IGBT semiconductor elements in the bridges of the converter shown in figure 1 when used for connecting a line commutated converter and a voltage source converter;

Figure 4 illustrates an equivalent circuit to figure 1 ;

Figure 5 shows simulation results for the operation of the DC/DC converter shown in figure 1.

Figure 6 shows simulation results for the case of a pole-to-pole DC fault at the bridge 1 side, which is connected to a LCC HVDC system

Figure 7 shows simulation results for the case of a pole-to-pole DC fault at the bridge 2 side, which is connected to a VSC HVDC system

Figure 8 illustrates a DC/DC converter of a second embodiment of the present invention.

[0022] Referring to figure 1 , a DC/DC convertor of a first embodiment of the present invention, the convertor comprises a first bridge identified as bridge 1 and a second bridge, identified as bridge 2. Bridge 1 is connected between a line 20 (DC side), which is connected via an inductor LI to a DC voltage source VI (first power source), and a line 21 which is also connected to the voltage source VI . Bridge 2 is connected between a line 30 (DC side), which is connected via an inductor L2 to a DC voltage source V2 (second power source), and a line 31 which is also connected to the voltage source V2.

[0023] Bridge 1 comprises two branches, each having a pair of bi-directional switches, each bi-directional switch in the form of two IGBT semiconductor elements.

[0024] The first branch has an upper section which comprises a bi-directional switch in the form of an IGBT semiconductor element S 1 1 with its collector connected to line 20. The emitter of element Sl l is connected to the emitter of another IGBT semiconductor element S14 which has its collector connected to a line 22 (AC side). A diode is connected in parallel with each of the IGBT semiconductor elements. Thus, a first bi-directional switch is produced in the form of a pair of series connected IGBT semiconductor elements, each having a diode in parallel.

[0025] The first branch has a lower section which comprises a bi-directional switch in the form of an IGBT semiconductor element SI 8 with its collector connected to the line 21. The emitter of element SI 8 is connected to the emitter of another an IGBT semiconductor element SI 5 which has its collector also connected to line 22. This comprises a second bi-directional switch. A diode is connected in parallel with each of the IGBT semiconductor elements. Thus, a second bi-directional switch is produced in the form of a pair of series connected IGBT semiconductor elements, each having a diode in parallel.

[0026] The second branch has an upper section which comprises a bi-directional switch in the form of an IGBT semiconductor element S13 with its collector also connected to line 20. The emitter of element S13 is connected to the emitter of another IGBT semiconductor element S12 which has its collector connected to a line 23 (AC side). A diode is connected in parallel with each of the IGBT semiconductor elements. Thus, a third bi-directional switch is produced in the form of a pair of series connected IGBT semiconductor elements, each having a diode in parallel.

[0027] The second branch has a lower section which comprises a bi-directional switch in the form of an IGBT semiconductor element S16 with its collector also connected to the line 21. The emitter of element S16 is connected to the emitter of another element S17 which has its collector also connected to line 23. A diode is connected in parallel with each of the IGBT semiconductor elements. Thus, a fourth bidirectional switch is produced in the form of a pair of series connected IGBT semiconductor elements, each having a diode in parallel.

[0028] Bridge 2 comprises two branches, each having a pair of bi-directional switches, each bi-directional switch in the form of two IGBT semiconductor elements.

[0029] The first branch has an upper section which comprises a bi-directional switch in the form of an IGBT semiconductor element S21 with its collector connected to line 30. The emitter of element S21 is connected to the emitter of another element S24 which has its collector connected to a line 32. A diode is connected in parallel with each of the IGBT semiconductor elements. Thus, a fifth bi-directional switch is produced in the form of a pair of series connected IGBT semiconductor elements, each having a diode in parallel.

[0030] The first branch has a lower section which comprises a bi-directional switch in the form of an IGBT semiconductor element S28 with its collector connected to the line 31. The emitter of element S28 is connected to the emitter of another element S25 which has its collector also connected to line 32. A diode is connected in parallel with each of the IGBT semiconductor elements. Thus, a sixth bi-directional switch is produced in the form of a pair of series connected IGBT semiconductor elements, each having a diode in parallel.

[0031] The second branch has an upper section which comprises a bi-directional switch in the form of an IGBT semiconductor element S23 with its collector also connected to line 30. The emitter of element S23 is connected to the emitter of another element S22 which has its collector connected to a line 33 A diode is connected in parallel with each of the IGBT semiconductor elements. Thus, a seventh bi-directional switch is produced in the form of a pair of series connected IGBT semiconductor elements, each having a diode in parallel.

[0032] The second branch has a lower section which comprises a bi-directional switch in the form of an IGBT semiconductor element S26 with its collector also connected to the line 31. The emitter of element S26 is connected to the emitter of another element S27 which has its collector also connected to line 33. A diode is connected in parallel with each of the IGBT semiconductor elements. Thus, an eighth bidirectional switch is produced in the form of a pair of series connected IGBT semiconductor elements, each having a diode in parallel.

[0033] The lines 22 and 23 are connected across the windings Lacl of one side of a transformer 101 with a capacitor CI in parallel. The lines 32 and 33 are connected across the windings Lac2 of the other side of the transformer 101 with a capacitor C2 in parallel.

[0034] The arrow I ₁ shows the current flow through inductor LI and the arrow I ₂ shows the current flow through inductor L2.

[0035] Due to the generalised topology, different current paths exist, which influences the AC voltage shapes on the transformer's sides. The direction of the current flow can be changed by switching either top or bottom IGBT in each back-to-back bidirectional switch pair.

[0036] When the DC/DC converter described with reference to figure 1 connects a voltage source converter (VSC) of voltage VI and a voltage source converter (VSC) of voltage V2, the DC currents in both sides of the converter can be positive or negative with respect to the direction of the power flow. In this case, there are 8 combinations depending on the turning ON and OFF of the power switches which influences the AC voltage shapes on the sides Lacl and Lac2 of the transformer. A control means 100 provides gate signals to the IGBT semiconductor elements S11-S18 and S21-S28. Tables 1 and 2 show the switching sequences of these IGBT semiconductor elements for bridge 1 and bridge 2 respectively according to those gate signals. VI, II and PI represent the DC side voltage, current and power respectively of bridge land V2, 12 and P2 represent the DC side voltage, current and power respectively of bridge 2

Table 2

[0037] Referring to figure 2A, this shows the situation for I ₁ >0, 1 ₂ <0, +ve half cycle at bridge 1, and +ve half cycle at bridge 2.

[0038] Referring to figure 2B, this shows the situation for I ₁ >0, 1 ₂ <0, +ve half cycle at bridge 1, and -ve half cycle at bridge 2.

[0039] Referring to figure 2C, this shows the situation for I ₁ >0, 1 ₂ <0, -ve half cycle at bridge 1, and +ve half cycle at bridge 2.

[0040] Referring to figure 2D, this shows the situation forI ₁ >0, 1 ₂ <0, -ve half cycle at bridge 1, and -ve half cycle at bridge 2.

[0041] Referring to figure 2E, this shows the situation for I ₁ <0, 1 ₂ >0, +ve half cycle at bridge 1, and +ve half cycle at bridge 2.

[0042] Referring to figure 2F, this shows the situation for I ₁ <0, 1 ₂ >0, +ve half cycle at bridge 1, and -ve half cycle at bridge 2.

[0043] Referring to figure 2G, this shows the situation for I ₁ <0, 1 ₂ >0, -ve half cycle at bridge 1, and +ve half cycle at bridge 2.

[0044] Referring to figure 2H, this shows the situation for I ₁ <0, 1 ₂ >0, -ve half cycle at bridge 1, and -ve half cycle at bridge 2.

[0045] In figures 2A-2H, the switches which do not have current flow are shown in phantom outline whilst the overall current flows are shown in grayscale.

[0046] When the DC/DC converter described with reference to figure 1 connects a line commutated converter (LCC) of voltage VI and a voltage source converter (VSC) of voltage V2, the DC current is fixed at the LCC and the DC voltage must be variable in magnitude and polarity to control power transfer magnitude and direction respectively. In this case, there are 8 combinations depending on the turning ON and OFF of the power switches which influences the AC voltage shapes on the sides Lacl and Lac2 of the transformer. Control means 100 provides gate signals to the IGBT semiconductor elements S11-S18 and S21-S28. Tables 3 and 4 show the switching sequences of these IGBT semiconductor elements for bridge 1 and bridge 2 according to those gate signals. VI, II and PI represent the DC side voltage, current and power respectively of bridge land V2, 12 and P2 represent the DC side voltage, current and power respectively of bridge 2

Table 3

Table 4

[0047] Referring to figure 3A, this shows the situation for I ₁ >0, 1 ₂ <0, +ve half cycle at bridge 1, and +ve half cycle at bridge 2.

[0048] Referring to figure 3B, this shows the situation for I ₁ >0 and I ₂ <0, +ve half cycle at bridge 1, and -ve half cycle at bridge 2.

[0049] Referring to figure 3C, this shows the situation for I ₁ >0, I ₂ <0, -ve half cycle at bridge 1, and +ve half cycle at bridge 2.

[0050] Referring to figure 3D, this shows the situation for I ₁ >0, I ₂ <0, -ve half cycle at bridge 1, and -ve half cycle at bridge 2.

[0051] Referring to figure 3E, this shows the situation for I ₁ >0, 1 ₂ >0, +ve half cycle at bridge 1, and +ve half cycle at bridge 2.

[0052] Referring to figure 3F, this shows the situation for I ₁ >0, 1 ₂ >0, +ve half cycle at bridge 1, and -ve half cycle at bridge 2.

[0053] Referring to figure 3G, this shows the situation for I ₁ >0, 1 ₂ >0, -ve half cycle at bridge 1, and +ve half cycle at bridge 2. [0054] Referring to figure 3G, this shows the situation for I ₁ >0, 1 ₂ >0, -ve half cycle at bridge 1, and -ve half cycle at bridge 2.

[0055] Figure 4 illustrates the equivalent circuit of the DC/DC converter shown in figure 1 from the AC side for the purposes of circuit analysis. Although both AC signals are square wave, only fundamental components will be used in the analysis as it is the main cause of the power flow.

Substitute equation (2) into equation (1)

Substitute equation (3) into equation (2)

Calculation of apparent power (S) for bridge 1

Power balance between AC and DC sides

Assuming lossless bridges, power on the DC sides should be equivalent to power on the AC side.

Assuming bridges generating full square waves from DC side currents:

Now substitute with

[0056] From the converter main power transfer equation, three control variables exist as follows:

1. Phase shift angle control between the two bridges (δ)

2. Converter switching frequency (co)

3. DC link voltages (V _dcl , V _dc2 )

[0057] The control strategy of the DC/DC converter depends on what the converter of the invention is connecting (voltage or current source).

[0058] In the case when the two sides of the DC/DC converter are connected to DC voltage source links (fixed V _dcl , V _dc2 ), the available control techniques are (δ) and (co). However, control of (δ), would result in unstable operation since the relation between P and δ is inverse with infinite power at δ=0. This means that the only feasible control technique when the DC/DC converter is connecting two voltage sources is varying the converter switching frequency (co).

[0059] In the case when either side of the converter is not connected to a voltage source (V _dc , or V _dc2 are not fixed), for example, one side of the DC/DC converter is connected to a current source TCC and the other side to a voltage source VSC, this means the transfer power can be controlled by varying the converter switching frequency (co) or the DC link voltage of the current source side.

[0060] Fig. 5 shows simulation results for the operation of the DC/DC converter embodying the present invention. In this simulation, bridge 1 is connected to an LCC and bridge 2 is connected to a VSC. The figure shows results for rated power operation, full power reversal and partial power operation; figure 5(a) bridge 1 active and reactive power, figure 5(b) bridge 2 active and reactive power, figure 5(c) bridge 1 DC voltage and current, and figure 5(d) bridge 2 DC voltage and current.

[0061] The simulated operation is detailed as follows:

From t=0 to t=0.5s, converter is operating with full rated power in forward direction (power transfer from bridge 1 to bridge 2).

From t=0.5 to t=1.0s, converter is operating with full rated power in reverse direction (power transfer from bridge 2 to bridge 1).

From t=1.0 to t=1.5s, converter is operating with half rated power in forward direction.

From t=1.5 to t=2.0s, converter is operating with half rated power in reverse direction.

[0062] Figures 5(a) and 5(b) show successful operation of the DC/DC converter with active power transfer (PI and P2) in both directions at rated and partial loading. Local reactive power at the bridges (Ql and Q2) is zero at full rated power with minor reactive power contribution at partial loading. Fig. 5(c) shows that for bridge 1, the current II is controlled to a fixed value of 1 p.u at all stages due to operation in conjunction with current source (LCC). Active power reversal and partial power operation is achieved by control of the DC voltage externally from the front-end converter (omitted here and represented by equivalent DC voltage VI). This can be seen from the control of VI between 1, -1, 0.5 and -0.5 p.u for the various directions and levels of power transfer. Fig. 5(d) illustrates bridge 2 DC voltage and current. Since bridge 2 is connected to VSC HVDC, voltage V2 is fixed at 1 p.u and current direction and magnitude determine level of power transfer.

[0063] As noted above, the DC/DC converter of the present invention inherently and naturally possesses open circuit characteristics in response to DC side faults.

[0064] Firstly, the DC faults scenario of bridge 1 will be considered with reference to figure 6.

[0065] Figure 6 shows simulation results for the case of a pole-to-pole DC fault at the bridge 1 side, which is connected to a LCC HVDC system. Full rated power is being sourced from bridge 2 when the permanent fault is applied at t=1.0s at bridge 1. Figure 6(a) shows bridge 1 active and reactive power, figure 6(b) shows bridge 2 active and reactive power, figure 6(c) shows bridge 1 DC voltage and current, and figure 6(d) shows bridge 2 DC voltage and current.

[0066] It can be seen that figures 6(a) and (b) show how active power transfer falls to zero during the fault and bridge 1 (where the fault is applied) is drawing reactive power only to feed the fault. Figure 6(c) shows VI drops to zero due to the fault at t=1.0s and the current II is yet controlled to lp.u since this side of the DC/DC converter is connected to a current source system. The peak transient overcurrent is less than 2 p.u. This lies within the safe operating region of most IGBTs, hence confirming the "no need to trip" characteristic of the faulty side IGBTs. Figure 6(d) shows how the current 12 at the non-faulty side of the DC/DC converter naturally (without any external control or semiconductor tripping) drops to zero when the fault is applied at bridge 1. This confirms the open circuit characteristics that this converter possesses which makes it favourable as a DC circuit breaker in the event of a fault.

[0067] Figure 7 shows simulation results for the case of a pole-to-pole DC fault at the bridge 2 side, which is connected to a VSC HVDC system. Full rated power is being sourced from bridge 1 when the permanent fault is applied at t=1.0s at bridge 2. Figure 7(a) shows bridge 1 active and reactive power, figure 7(b) shows bridge 2 active and reactive power, figure 7(c) shows bridge 1 DC voltage and current, and figure 7(d) shows bridge 2 DC voltage and current.

[0068] It can be seen that figures 7(a) and (b) show how active power transfer falls to zero during the fault and bridge 2 (where the fault is applied) is drawing reactive power only to feed the fault. Figure 7(c) shows VI constant at lp.u with the current II falling naturally (with no external control or semiconductor tripping) to almost zero confirming the circuit breaker behaviour inherently possessed by the DC/DC converter of the present invention at the non-faulty side in response to DC faults. Figure 7(d) shows how V2 drops to zero due to the fault applied at t=1.0s and the current 12 is maintained at rated 1 p.u value. Peak transient overcurrent at the fault-side lies within IGBTs safe operating region hence allowing continued operation with no tripping.

[0069] It can be seen from figures 6 and 7 that because of the rated current at the faulty side and the zero current at the non-faulty side, semiconductor tripping is unnecessary with the DC/DC converter of the present invention. This enables continued operation after fault recovery with no need to re-synchronize with the grid.

[0070] It will be apparent that the present invention is capable of modification, the details of which will be readily apparent to the person skilled in the art

[0071]. In this respect, figure 8 illustrates a DC/DC converter of a second embodiment of the present invention. Components common to the first embodiment bear the same reference numbers. It can be seen from figure 8 that each of the eight bidirectional switch in figure 1 has been replaced by bi-directional switches (S1-S8) each in the form of a diamond configuration of diodes with one IGBT semiconductor element connecting opposing apexes of the configuration. In particular, referring to switch SI, the diamond configuration comprises a pair of diodes DIA and DIB interconnected at point W for forward bias. The point W is connected to the line 20. Another pair of diodes DIC and DID are interconnected at point Y for forward bias The point Y is connected to line 22. The end of each pair of diodes are interconnected at points X and Z. An IGBT semiconductor is connected between points X and Z. Thus, a first bi-directional switch is formed. Each of the switches S1-S8 has the same configuration as described above. As will be apparent to a person skilled in the art, the switches S1-S8 can be controlled in a similar manner to the switches of figure 1. The results replicate the simulation results