More specifically the invention relates to a capacitance measuring circuit, comprising an oscillator circuit, where the sensor capacitance forms a link of two or more series connected impedances, which is connected to the capacitance measuring circuit only at the end terminals of the series connection, and where one terminal of the sensor capacitance is an end terminal of the series connection, which is connected to a constant voltage and the other end terminal of the series connection is connected to a sensing node, which oscillates with a square wave voltage, in order to maintain constant voltage levels during each of the half-periods, and which at the same time senses the charging current flowing into the series connected impedances, to enable the capacitance measuring circuit to be triggered at the moments when the charging current thus sensed has changed to predetermined levels.

Capacitances have for many years been measured by a range of well known standard RC oscillator circuits, where the frequency is determined by measuring the charging voltage directly on the sensor capacitance and comparing it with reference voltages.

This measurement is a high impedance measurement and the standard RC oscillator circuits will be sensitive to stray capacitances, as these will appear in parallel with the sensor capacitance.

Changes in these capacitances are therefore impossible to discern from each other. The stray capacitances in the standard RC oscillator circuits comes mainly from the wires connecting the sensor capacitance to the oscillator circuit, and from the input capacitance of the oscillator circuit itself and as they often are of the same magnitude as the sensor capacitance, and not constant, they present a serious problem in practice.

In US patent 4737706, a capacitance measuring circuit of the kind, where the sensor capacitance forms a link of two or more series connected impedances, is shown.

Here the current sensing and square wave generation is performed by an operational amplifier with a feed- back resistor and the output voltage of the operational amplifier, which is a function of the charging current in the series connected impedances, is compared in a voltage comparator with a positive and a negative reference voltage during respectively the positive and the negative cycles of the oscillator.

As explained this circuit has the advantage, that errors coming from the stray capacitance of the sensing node and the stray capacitance of the connection to the sensor capacitance are reduced.

This capacitance measuring circuit functions by measuring the charging current of the sensor capacitance, through the series connected impedances, connected to the low impedance sensing node of the measuring circuit. Hereby the high impedance of the sensor capacitance is insulated from the low impedance sensing node by the series connected impedances, which means that the influence of the stray capacitance at the sensing node and the influence of the stray capacitance of the connecting node between the sensor capacitance and the series connected impedance are separated.

By placing the series connected impedance directly at the sensor capacitor, the stray capacitance at the connecting node between the sensor capacitor and the series connected impedance may be very small, and not of any importance.

The low impedance of the sensing node will charge the stray capacitance of the sensing node fast, and long before the trigger level of the charging current is reached and therefore the influence of the stray capacitance of the sensing node is reduced.

In practice the capacitance measuring circuit according to US patent 4737706 has a number of limitations, primarily because the ability of the circuit to reduce the influence of stray capacitances is depending on a fast and very low impedance source to generate the square wave voltage, which is impressed on the series connected impedances, and the square wave voltage, as generated in US patent 4737706, by the output of an operational amplifier has relatively long rise times and a relatively high source impedance.

Therefore stray capacitances are only reduced to a certain degree.

Secondly the precision of the circuit relies mainly on the speed and the precision of the measurement of the charging current and its conversion into a voltage.

The input bias currents and offset voltages of the operational amplifier in the circuit according to US patent 4737706 are temperature dependent and will influence the measurement of the charging current into the series connected impedances, and temperature dependent variations in the open loop amplification factor of the operational amplifier will influence the conversion of the charging current into the output voltage presented to the voltage comparator, and at last the speed of the available operational amplifiers is limited because the output value is a voltage.

For these reasons the accuracy of the circuit according to US patent 4737706 is not ideal.

In patent EP 1 386 173, an accurate and fast square wave voltage is impressed on that end of the series connected impedances, which is connected to the sensing node of the capacitance measuring circuit by shifting the complete capacitance measuring circuit alternately between two voltages with constant voltage levels in the half periods. The square wave will be defined by these two voltage levels and if fast, low impedance switches and low impedance voltage sources are implemented and if the current comparator is fast and has a low input impedance, then this circuit generates a far more accurate square wave on the sensing node than the circuit according to US 4737706 and an important condition for reducing the influence of stray capacitances is fulfilled. The capacitance measuring circuit according to EP 1 386 173 has however the limitation that a rather high supply voltage is necessary because, in addition to the supply voltage for the current comparator etc, there has to be a certain voltage available over and also under this supply voltage in order to keep the supply current alive when the whole circuit is shifted up and down.

In practice the supply voltage for the circuit is around three times the supply voltage for the comparator and the other components and in times where power consumption is a major issue, this high supply voltage is a definite drawback.

Furthermore, the shifting up and down of the whole circuit generates a substantial noise voltage which interferes with the time measuring circuits which are connected to the capacitance measuring circuit in order to provide the capacitance values.

Furthermore both the capacitance measuring circuits according to US 4737706 and to EP 1 386 173, shares the drawback that the accuracy of the square wave impressed on the series connected impedances is dependent on the characteristics of an amplifying circuit, that on one hand has to measure a low voltage or a low current very precisely and on the other hand has to be very fast and provide a low input impedance in order to source a precise square wave.

The precision of the square wave is very important, because the charging of the capacitor is determined by the integral of each of the periods of the square wave, which means that long rise and fall times will reduce the integral of the periods and hereby lengthen the periods, compared to the periods of capacitance measuring circuits with a perfect square wave.

Because the rise and fall times are temperature dependent, the importance of a precise square wave is obvious.

In practice the square waves typically have a period of 4 microseconds and rise- and fall times of typically 80 ns with the square wave generated by the operational amplifier of

US patent 4737706 and typically 20 ns with the square wave generated by the current amplifier of EP 1 386 173.

These rise- and fall times, each corresponds to 2% respectively 0.5% of the period of 4 microseconds and that again means that the square waves generated by the two circuits are 4% respectively 1% from being ideal.

With the ever increasing demand on accuracy an important advantage could be gained by reducing these rise times.

It is the object of the invention to provide an improved capacitance measuring circuit of the type initially mentioned, but without the aforementioned drawbacks. According to the invention a highly accurate and very fast square wave voltage is impressed on the series connected impedances from a separate low impedance source, and the measurement of the charging current, into the series connected impedances is performed by a separate high accuracy current sensing device, connected in series between the low impedance square wave source and the series connected impedances.

In a first preferred embodiment of the invention, the square wave generation is performed by a separate buffer and the measurement of the charging current is performed by a separate current sensor mounted between the buffer and the series connected impedances and a single comparator in connection with a reference current for switching the circuit when the sum of the signal from the current sensor and the reference current crosses the reference level of the comparator.

If the reference level of the comparator is chosen near zero, the amplification factor of the comparator does not influence the switching level.

The series connected impedances consists preferably, but not necessarily of the capacitance to be measured with one end grounded and the other end connected to the current sensor through a resistor.

In this embodiment of the invention the advantage of separating the generation of an almost perfect square wave and the precision measurement of the charging current in two separate units is demonstrated.

In a second preferred embodiment of the invention, the square wave generation is performed by a separate buffer and the measurement of the charging current is performed by a separate current sensor mounted between the buffer and the series connected impedances and two separate comparators for switching the circuit when the signal from the current sensor crosses the reference levels of respectively the positive sensing and the negative sensing comparator.

In a third preferred embodiment of the invention, the square wave generation is performed by a separate buffer and the measurement of the charging current is performed by a separate current sensor mounted between the buffer and the series connected impedances and two separate comparators for switching the circuit when the signal from the current sensor crosses the reference levels, set by bias resistors, of respectively the positive sensing and the negative sensing comparator.

In this embodiment both comparators are continuously sensing the charging current and a logic unit switches the appropriate comparator output to control the buffer and one of the comparators operate at the high level of the square wave and the other at the low level of the square wave, which places less demands of the common mode rejection of the comparators. In a fourth preferred embodiment of the invention, the square wave generation is performed by a separate buffer and the measurement of the charging current is performed by a separate current sensor mounted between the buffer and the series connected impedances and a single comparator with a bias circuit, switches the buffer through a logic unit.

In a fifth preferred embodiment of the invention, the square wave generation is performed by a separate buffer and the measurement of the charging current is performed by a separate current sensor mounted between the buffer and the series connected impedances and a single comparator with a bias circuit, switches the buffer through a logic unit.

A bootstrap circuit for the power supply of the comparator, places less demands on the common mode range and the common mode rejection of the comparator.

In a sixth embodiment of the invention, the sensing node is connected to a plurality of series connected impedances, including a plurality of sensor capacitances, through a multiplexing circuit.

This embodiment has the advantage that several sensor capacitances are measured by the same circuit, which gives the possibility of for example using differential measuring methods.

The invention will now be described in further detail where,

Fig. 1 shows a preferred embodiment of the circuit according to the invention.

Fig. 2 shows the square wave voltage from the buffer.

Fig. 3 shows the charging current through the series connected impedance Rt, with the fast transitions and the exponentially changing charging current between the transitions.

Fig. 4 shows the voltage on the measured capacitor.

Fig. 5 shows the square wave from the buffer generating the reference current.

Fig 6 shows another preferred embodiment of the circuit according to the invention with two separate comparators with programmed switching levels.

Fig 7 shows another preferred embodiment of the circuit according to the invention with two separate comparators with switching levels defined by bias circuits

Fig 8 shows another preferred embodiment of the circuit according to the invention with one comparator with switching levels defined by bias circuits.

Fig 9 shows another preferred embodiment of the circuit according to the invention with one comparator with switching levels defined by bias circuits and a bootstrapped power supply for the comparator. The capacitance measuring circuit of figure 1, contains the buffer Bl, with the output voltage Ug, which, through the current sensor CS, is connected to the series connected impedances Rtl/Ctl or Rt2/Ct2 through the switch SWl.

The output signal Um, from the current sensor CS is measured by the comparator K. Reference currents are created by the buffer B2 and the resistor RS.

The buffer Bl has the supply voltages UdI and UsI, which typically are +- 2.5 VDC, and the buffer B2 has the supply voltages Ud2 and Us2 which both have to be higher than the supply voltages for B 1.

The output voltage of buffer B2 is shown in figure 5.

The comparator K has the supply voltages Udk and Usk which have to be high enough to ensure that the signal from CS does not exceed the common mode voltage of K.

When the output voltage of Bl, which is shown in figure 2, goes to the positive value UdI, a charging current Jrt, which is shown in figure 3, is injected through the series connection of the current sensor CS, the switch SWl and one of the series connected sets of impedances Rt/Ct.

At the same time the buffer B2 injects a positive reference current Jt+ through the resistor RS into the connection node between the current sensor CS and the switch SWl.

With ideal buffers, and with a very low series impedance for CS, Jt+ has the value (Ud2- Udl)/RS.

When the charging current has fallen to the level Jt+, the comparator K, through the logic unit L, switches the buffer Bl to the negative value UsI and oscillations with a period controlled by the values of Rt/Ct and the different voltages are created.

In figure 4, the voltage on the capacitor Ct, which is not measured, may be compared with the measured charging current Jrt in figure 3, for better understanding of the function of the capacitance measuring circuit according to the invention.

The logic unit L controls the interaction between the buffers, comparators and switches and may provide information to time measuring systems with start and stop pulses for a certain number of oscillations with each Rt/Ct combination and information on when Rtl/Ctl are active.

With separate buffers, which are optimized for only sourcing a nearly perfect square wave, rise and fall times of less than 5 ns and output impedances of less than 10 ohm may be obtained.

With comparators, which are based on very fast operational amplifiers, with 1000 V/microsecond slew rate and a very low noise voltage of 0.5 nV , and which only have to measure the charging currents, a very high precision current measurement may be obtained. The signal voltage of the current sensor CS, and hereby the noise, and the reduction of the influence of the stray capacitances may be traded by increasing or decreasing the series impedance of the current sensor CS, relative to the value of Rt.

The current sensor may take the form of a simple low value resistor, a current transformer or any other device which can provide a precision signal from the value of the charging current.

A very important characteristic of the capacitance measuring circuit according to the invention is that the value of the series impedance of the current sensor, has no influence on the accuracy of the charging current when this value is kept constant.

The capacitance measuring circuit of figure 6, contains the buffer B, with the output voltage Ug, which, through the current sensor CS, is connected to the series connected impedances Rtl/Ctl or Rt2/Ct2 through the switch SWl.

The positive and negative charging currents are measured separately by the two separate comparators K+ and K-, which are each optimized for operating around the levels for Ud and Us and which have the triggering levels REF+ and REF-, respectively.

The capacitance measuring circuit of figure 7, contains the buffer B, with the output voltage Ug, which, through the current sensor CS, is connected to the series connected impedances Rtl/Ctl or Rt2/Ct2 through the switch SWl.

The positive and negative charging currents are measured separately by the two separate comparators K+ and K-, which are each optimized for operating around the levels for Ud and Us respectively.

The comparators are biased by the resistor networks R1/R2 and R3/R4 and are optimized for triggering at around zero voltage difference between the two comparator inputs.

The capacitance measuring circuit of figure 8, contains the buffer B, with the output voltage Ug, which, through the current sensor CS, is connected to the series connected impedances Rtl/Ctl or Rt2/Ct2 through the switch SWl.

The positive and negative charging currents are measured by the comparator K, which is biased by the resistor network Rl and R2/R4 and which is optimized for triggering at around zero voltage difference between the two comparator inputs.

Because only one comparator is used with a trigger level around zero voltage the important advantage is obtained that a shift of trigger level of the comparator will not result in a change of period, but only result in a relative change of the two half-periods of the oscillation. The capacitance measuring circuit of figure 9, contains the buffer B, with the output voltage Ug, which, through the current sensor CS, is connected to the series connected impedances Rtl/Ctl or Rt2/Ct2 through the switch SWl.

The positive and negative charging currents are measured by the comparator K, which is biased by the resistor network Rl and R2/R4 and which is optimized for triggering at around zero voltage difference between the two comparator inputs.

The supply voltages, Vdk and Vsk, for the comparator K, is generated through the resistors R5/R6 and the capacitors C1/C2 by bootstrapping the voltages Vd and Vs from the output of B.

Hereby the specifications for the common mode range and the common mode rejection of the comparator K are far less important.

Because only one comparator is used with a trigger level around zero voltage the important advantage is obtained that a shift of trigger level of the comparator will not result in a change of period, but only result in a relative change of the two half-periods of the oscillation.

Where fast, low noise operational amplifiers are implemented as comparators, a feedback impedance may advantageously be connected from the output to the inverting input of the operational amplifier in order to provide a stable amplification factor.

Due to the fact that a preferred embodiment of the invention has been illustrated and described herein, it will be apparent to those skilled in the art that modifications and improvements may be made to forms herein specifically disclosed.

Accordingly, the present invention is not to be limited to the forms which are specifically disclosed, the supply voltages, for example, may assume higher or lower values.