Background of the Invention

A. Field of the Invention

The present invention is directed generally to the field of economizers for spring- biased DC actuators. In particular, the present invention provides a control circuit for power reduction using an electronic chopper incorporating an oscillator having a variable duty cycle responsive to the source voltage to maintain average coil holding current at a level to assure minimum required magnetomotive force (MMF) for the actuator coil. Additionally, the invention includes an electronic dropout switch providing high impedance for rapid current decay upon deactivation of the actuator.

B. Prior Art

Spring-biased DC actuators are employed for numerous applications including high power contactors requiring fail-safe capability. Such actuators typically require significant MMF to overcome armature and contactor inertia and spring force for closing the contactor. However, once closed, the contactor may be retained in a closed position at significantly lower MMF. Various prior art methods have been applied to provide initial high currents to the actuator coil to obtain contact closure with subsequent reduction of the current to maintain MMF at a level required to resist the spring force and maintain the contact in the closed position. To assure minimum MMF in conditions of high temperature, wherein coil resistance may be significantly altered, and power changes due to voltage sagging in the power supply, most designs require a significant safety factor to be applied to the current level maintained in the coil, precluding optimizing of performance. Prior art systems employing chopper circuits to reduce average current in the coil must provide sufficient MMF to accommodate worst-case conditions. Higher currents which need to be maintained exacerbate the situation by contributing to heating of the coil.

In addition, the use of most prior art choppers as an economizer circuit results in difficulties during de-actuation of the circuit. Addition of a chopper controller normally increases dropout time due to slow decay of coil current through the low-impedance return circuit employed in chopper economizers. Regeneration in the coil created by motion of the armature while current is still flowing in the coil impacts the velocity and kinetic energy of the armature, thereby extending the dropout time and weakening the action that breaks open sticky contacts. The present invention overcomes the difficulties in the prior art to allow

high efficiency in conservation of coil power, enhancement of mechanical performance, and potential reduction in size of actuators for given applications.

Summary of the Invention The present invention provides an improved economizer for DC actuator coil current control in spring-biased DC actuators. The actuator coil circuit receiving power from a voltage source is controlled by a chopper circuit which incorporates a switch in the coil circuit which is closed responsive to a control signal generated by a power-switching circuit. The power-switching circuit incorporates a threshold detector which provides a triggering signal to initiate operation of the actuator and the control circuit responsive to a predetermined source voltage, a gate signal generator responsive to the triggering signal and having a time constant sufficient to drive the coil through a pickup interval, and a variable duty cycle oscillator for generating the switch control signal. The oscillator passes the gate signal during the pickup interval and subsequently varies its duty cycle responsive to the voltage of the voltage source.

Modification of duty cycle for the oscillator is accomplished in a first embodiment under voltage control and in a second embodiment under current control, both operated in response to the power source for the coil. The change in the duty cycle increases or decreases the period in which the switch for control of the coil circuit remains on, providing energy pulses to sustain average coil current at the desired MMF. The ability to compensate for voltage sag and resultant current decrease in the coil reduces the safety factor necessary to assure minimum MMF to adequately resist spring force. Environmental temperature variations become the only consideration for safety margin in the voltage controlled embodiments while the current controlled embodiment compensates for temperature variation as well.

The present invention also incorporates a dropout switch for opening the sustaining current return circuit for the coil, which operates in concert with the power switching circuit to maintain coil current. The dropout switch is opened by removal of source voltage, producing a high impedance for rapid current drain from the coil, thereby eliminating the regeneration effect typically present in economizer circuits.

Brief Description of the Drawings

The present invention will be more clearly understood in reference to the following drawings and detailed description.

FIG. 1 is a block diagram of a first embodiment of the invention employing a voltage- controlled oscillator;

FIG. 2 is a circuit schematic for the embodiment of FIG. 1; FIG. 3 is a graph demonstrating armature and contact position in relation to coil current controlled by the embodiment of the invention disclosed in FIG. 2;

FIGS. 4a, 4b and 4c graphically depict variation of the duty cycle for the oscillator of the embodiment shown in FIG. 2 based on a reduction in source voltage;

FIG. 5 is a block diagram of a second embodiment of the invention employing a current-controlled oscillator;

FIG. 6 is an electrical circuit schematic of the embodiment shown in FIG. 5; FIG. 7 is a graph demonstrating position of the armature and contacts in relation to coil current for the embodiment of the invention shown in FIG. 6;

FIGS. 8a, 8b and 8c graphically depict the duty cycle for the embodiment of the invention shown in FIG. 6 based on a reduction in source voltage;

FIGS. 9a and 9b graphically represent armature position and the effects of regeneration based on residual current in the coil of an economizer system without the dropout circuit of the present invention;

FIGS. 10a and 10b graphically represent the armature position during deactivation of the actuator with the dropout circuit of the present invention;

FIG. 11 is an electrical circuit schematic for an embodiment of the invention substantially similar to FIG. 2 employing MOSFET switches and demonstrating component values for a typical application.

Detailed Description

Referring to the drawings, FIG. 1 shows a first embodiment of the invention in block diagram form. As exemplary of the application of the invention, a high-power contactor 100 constitutes the application in which the invention is used. High-power contactors require rapid armature acceleration for proper contact closure and rapid drop-out of the contactor to preclude arcing and to break open welded contacts. Significant armature power is required to minimize contact bounce to limit signal spikes and minimize weld strength caused by arcing during contact bounce during closure under electric load, and again preclude contactor damage by arcing. In FIG. 1, the contactor 100 is driven by coil 102 which is schematically represented by inductor L _c representing a 1600-turn coil having an inductance of approximately 0.2 H and a resistance R _c of 21 ohms. Power is supplied by a voltage source V _s which, in the applications shown, comprises a 28 VDC source, and opening and closing the contactor is accomplished by switch S ₀ . The present invention is embodied in the electronic controller 104 which comprises a chopper for reducing power from the voltage source to the coil. A switch S, opens the coil circuit, and sustaining current for the coil flows through a sustaining circuit incorporating diode D ₅ .

Switch S _j is controlled by an oscillator 106, a threshold voltage amplifier and a gate amplifier V _GATE . The threshold voltage amplifier or threshold voltage detector determines if the source V _s provides sufficient potential for pick-up to occur in the coil. Once the voltage threshold has been exceeded, the gate amplifier provides a gate signal which closes switch S< allowing the coil to receive full source voltage during the pick-up interval. After pick-up, the oscillator 106 begins chopping the source current by periodically opening switch Sj, allowing the holding current to decay to a steady state level based on the duty cycle of the oscillator. The oscillator is voltage-controlled to compensate for source voltage variations. Specifically, the duty cycle is increased when voltage sag occurs in the power supply, thus maintaining essentially constant average current in the coil. A drop-out switch S is provided in the current return to accommodate the second feature of the invention, rapid drop-out. Switch S ₂ is maintained in the closed position by the source voltage and, when source voltage is removed by opening of switch S ₀ , switch S ₂ opens which substantially increases the impedance in the coil circuit allowing rapid current drain.

FIG. 2 provides a detailed schematic for the embodiment of the invention shown in FIG. 1. FIG. 3 graphically demonstrates the coil current for the circuit of FIG. 2 and corresponding armature and contact position. When source voltage is applied through switch S ₀ , the threshold detector amplifier A _t establishes a reference voltage at the inverting input through resistor R., and zener diode Dj. Initially, source voltage is less than the breakdown voltage of D,, hence D< is a high impedance and the source voltage is initially impressed at the inverting input of A^ At the noninverting input of A,, the source voltage is divided by resistors R ₂ , R ₃ and R , impressing a lower voltage. Therefore, the inverting input prevails,

maintaining the output of A _! in an initial low condition. The low state of the output of amplifier A, prevents the gate generator, which comprises amplifier A ₂ , from producing an interval gate while the source voltage is too low to provide adequate pick-up MMF. When the source voltage exceeds the zener reference voltage for D ₁₍ a constant voltage V _j ^p is provided to the inverting input of the threshold detector amplifier A _j . Rising source voltage above V^ _EF increases the voltage at the noninverting input for Aj. Resistors R ₂ , R ₃ and R ₄ are calculated with reference to diode D _j such that the noninverting input for A _j exceeds the reference voltage when the source voltage reaches a desired pick-up voltage, nominally 23 volts DC for the 28 volts DC source shown in the embodiment in the drawings. When the noninverting input has sufficient voltage to switch amplifier A _j , the output of A _j goes high and is instantly pulled up to the source voltage through resistor R ₅ . In the embodiment shown in the drawings, substantial positive feedback through resistors R ₅ and R ₄ to the noninverting input of amplifier A _j introduces significant hysteresis in the threshold detector cycle. Consequently, the source voltage must be reduced to a voltage much lower than the pick-up voltage in order to reset the threshold detector.

When the threshold detector voltage is pulled up to source voltage, the output of amplifier A, provides a trigger for the monostable gate generator amplifier A ₂ . A ₂ single shots a rectangular output which drives through the inverting input of amplifier A ₃ , which comprises the oscillator as will be described in greater detail subsequently, closing switch S _j of FIG. 1 which is embodied in the circuit of FIG. 2 as a darlington transistor Q The time duration for the gate pulse is determined by charging of capacitor C _j through resistor R ₇ , resulting in a decaying voltage impressed at the noninverting input to gate generator A ₂ . The inverting input of A ₂ allows the darlington switch Q _j to be biased on through resistors R and R ₁₀ , initiating the pick-up cycle of the contactor. During the gate signal, the output of A ₂ is low providing a sink preventing charging of capacitor C ₃ to maintain the output of amplifier A ₃ in a high condition driving switch Q _i through resistors R ₁₇ and R ₁₈ . The gate generator interval ends when voltage developed across Cj exceeds the voltage developed at the inverting inputs of A ₂ via the source V _s .

The voltage controlled oscillator 106 of FIG. 1 comprises amplifier A ₃ and its associated circuitry in FIG. 2. The oscillator is free running or self-sustaining.

With the oscillator output high and the inverting input low, the noninverting input of A ₃ is switched to high as an initial condition at t„ as shown in FIG. 4b. With the output of A ₃ high, capacitor C ₃ is charged through resistors R _ϊ4 and R ₁₅ . The voltage at the noninverting input to A ₃ increases exponentially toward the voltage value of the output of A ₃ . The resulting cycle is shown in FIG. 4b, with the noninverting input signal V+ and the inverting input signal V- shown in relation to the reference voltage V _REF . Rise time for the inverting input is governed by the time constant T _RJSE which is equal to (R ₁₄ //R ₁₅ )C ₃ . The asymptote for the exponential increase of the inverting input is the output voltage for

amplifier A ₃ , Vosc.

At time t _v , the inverting input exceeds the noninverting input, driving the oscillator output low which completes an on-cycle for darlington switch Q ₁ With darlington Q _j switched off, sustaining current for the coil flows through the free-wheeling diode D ₅ . Substantial positive feedback for the oscillator amplifier A ₃ insures rapid transition in driving the noninverting input V- low. With the oscillator output signal V _osc low, capacitor C ₃ begins discharging into the low potential sink of the output of A ₃ . For the embodiment shown in the drawings, discharge of C ₃ is accomplished through resistor R ₁₅ only due to diode D ₇ blocking the path through resistor R ₁₄ . The discharge rate for capacitor Cg is, therefore, different than the charging rate with diode D ₇ forming a pass/blocking circuit regulating the duty cycle of the oscillator. At time t-w, the inverting input signal V-, decaying exponentially toward the ground potential asymptote with a time-constant τ _FALL governed by (R ₁₅ )C ₃ , is discharged below the noninverting input signal V+, the oscillator is again driven into a high state and the cycle is repeated. The oscillating frequency is entirely determined by selection of resistors R _H , R ₁₅ and capacitor C ₃ .

As previously discussed, the duty cycle of the oscillator in the present circuit is voltage controlled. Referring to FIG. 4a, if the source voltage V _s drops from 28 volts to 16 volts, the behavior of the oscillator can be shown in FIG. 4b. Reduction in the source voltage results in a reduction in the oscillator output voltage V _osc , affecting the upper asymptote for the charging cycle of the inverting input signal V-. T _RJSE has a significantly lower slope, increasing the time that the oscillator remains on, hence maintaining darlington Q _j on, providing current from the source to the coil. Since the ground potential provides the lower asymptote for the discharging cycle of the inverting input signal V- capacitor C^ discharges through resistor R ₁₅ starting from a lower potential, and since the low state of the noninverting input signal V+ is not shifted down as wholly determined by the constant voltage V _RE , the inverting input signal V- reaches the noninverting signal V+ in a shorter time. The cooperation of these two factors increases the duty cycle of the amplifier. This results in a fairly constant average voltage to the coil over a range of 36 volts to 14 volts for a nominal 28 volt condition with the embodiment shown in the drawings. The relative oscillator output signal V _osc is shown in FIG. 4c for both the nominal condition of source voltage at 28 volts and the sag condition of 16 volts as shown in FIGS. 4a and 4b described previously.

The drop-out feature of the present invention is provided by darlington transistor Q ₂ which acts as switch S ₂ of FIG. 1. When source voltage is applied to the circuit, switch k is fully on, allowing operation of the circuit in a powered mode through switch Qi and in a current return mode through free-wheeling diode D ₅ . Voltage from the source divided through resistors R ₂₀ and R ₂₁ provides the current bias to turn on switch Q ₂ . Capacitor C ₄ provides capacity for sustaining Q ₂ on which Q ₁ is in the off interval of the oscillator cycle.

The desired fast drop-out to rapidly drain current from the coil when the actuator is turned off is accomplished by the drop-out switch Q ₂ . Removal of source power causes Q ₂ to open, breaking the low impedance current-return circuit. In the embodiment shown in the drawings, reverse-biased zener diode D ₄ is then in the coil discharge loop in series with free- wheeling diode D ₅ . The dynamic resistance of diode D ₄ immediately and substantially raises the discharge time constant of the coil through D ₄ . This results in rapid coil-current decay. The operation of the present circuit without and with the fast drop-out switch is shown in FIGS. 9a and 9b and 10a and 10b, respectively. Without the rapid drop-out capability, the current in the coil decays slowly through free-wheeling diode D ₅ , as shown by curve 900. The armature position is shown in curve 910, with the contact state shown in curve 920. As the armature begins to move, the remaining current in the coil causes regeneration in the coil due to the continued magnetization of the armature. This in turn results in a reduction of velocity for the armature, as is clearly seen in FIG. 9b wherein the slope of curve 910 changes as the regeneration bump occurs. This slowing of velocity, which occurs substantially simultaneously with opening of the contacts, impedes rapid displacement of the contacts from the closed position, thereby increasing the opportunity for arcing as the contacts open and increasing the probability that welded contacts will not be opened by the less energetic armature action.

With the drop-out switch in place, removing source power results in an immediate decay of the coil current, as shown in FIG. 10a, with coil current represented by signal 1000 which immediately drops to zero. Armature position, shown by curve 1010, has significantly less velocity change since no regeneration occurs due to residual coil current. As shown in an expanded scale in FIG. 10b, the armature attains a full open position significantly sooner than shown in FIGS. 9a and 9b. The overall operation of the present invention is shown in FIG. 3 in a time profile comparing coil current, armature position and contact position. The coil current is shown as a function of time by trace 300, while the armature position is shown by trace 310 and contact position is shown by trace 320. The initial power pull-in interval shown in FIG. 3 includes the pick-up time t _PICKUP during which the gate signal amplifier provides the gate signal initiating power to the coil for closing of the armature. The time for initiation of operation of the oscillator to begin economizing by chopping the coil current is shown as t _delta . During the pick-up interval, the MMF for the coil builds to a total 1920 AT with a coil current of 1.2 amps and a total power supplied of 34 watts. When t _delta is reached and the economizer circuit is engaged, the chopped source voltage is readily apparent in the saw- tooth shape of the coil-current signal. In a few coil time constants, the holding current decays to a steady state holding level, resulting in an MMF of 610 AT with a coil power dissipation of only 3.4 watts. The drop-out portion of the cycle after removal of the source voltage from the coil is shown corresponding to FIGS. 9a and 9b for the case without the

drop-out switch of the present invention, identified as corresponding curves 900 and 910.

A second embodiment for the invention is shown in FIG. 5 in block diagram form. The basic conception of the invention is the same, however, the voltage-controlled oscillator is replaced with a current-controlled oscillator 106'. A detailed schematic for the current-controlled oscillator controller is shown in FIG. 6. Operation of the threshold detector and gate detector are substantially identical to that previously described for the voltage-controlled chopper, however, the circuit arrangements have been inverted to accommodate a PNP darlington Q _r for switch S _j . Overall concept of operation for the current-controlled circuit is similar to that disclosed for the voltage-controlled circuit, however, the oscillator amplifier A ₃ reacts to small voltage fluctuations in a sense resistor R _s connected to the noninverting input of the amplifier. The coil current rises and falls with power supplied and cut off from the voltage source through the darlington switch Q _j which is in turn controlled by the oscillator. Hence, since resistor R _s is in series with the coil and has the same current flow, the voltage signal across R _s is exactly proportional to the coil current and a closed loop for current control is provided. The inverting input for amplifier A ₃ maintains an exact reference derived by voltage divider from the reference voltage V ^ . The noninverting input of amplifier A ₃ tracks, with the exception of a small amount of positive feedback, the voltage developed across the current-sense resistor R _s . Without current control, the voltage across resistor R _s would rise in several coil time constants to a value of V _S (R _S /)R _S +R _C )) where R _c is the coil resistance. Setting the reference voltage applied to the inverting input of amplifier A ₃ anywhere below the maximum value achievable by the signal provided to the noninverting input will force current control because the oscillator will begin switching the voltage from the source through darlington Q _j , thereby limiting coil current. The current limit realized is dependent on the duty cycle of the oscillator and the duty cycle is defined by resistors R ₁₄ , R _n , R _s and the coil resistance R _c .

A calculation of the duty cycle is shown in equation 1.

Duty Cycle = Vref (R14)(Rs+Rc) EQ 1 Vs(R14+Rll)(Rs)

From equation 1, it is clear that the duty cycle increases as the coil resistance Rc increases and decreases as the source voltage V _s increases, both representing compensation in the correct direction. Operation of the current-controlled oscillator is best described with regard to FIGS. 8a, 8b and 8c. If the noninverting input signal V+ is lower than the inverting input signal V-, as indicated at time t _υ in FIG. 8b, the output of amplifier A ₃ is low, resulting in closure of switch Q,. Coil current begins to rise, resulting in a change in V+ sensed from resistor R _s . Subsequently, V+ rises exponentially with coil current toward the upper asymptote. At time t _v , the noninverting input signal V- exceeds the inverting signal input V-, triggering amplifier A ₃ to shut off, pulling up the output signal V _osc through resistor R ₁₇ . The speed of the switching action is increased by positive feedback leading

through capacitor C ₅ and sustained through resistor R ₁₆ . The sustained feedback shifts the noninverting input signal V+ above the inverting input signal V- by the ratio [V _s ][Rι ₅ /(R ₁₅ +R ₁₆ )]. where R ₁₆ > > R ₁₅ . The output of the amplifier V _osc being pulled high shuts off transistor Q _{l 5} opening the switch which ends the to _N interval. At time t _v , the coil current begins to decay through diode D ₅ . The coil current decays according to its time constant toward the lower asymptote, and the noninverting input of amplifier A ₃ receives the analog of this signal through resistor R ₁₅ as sensed from resistor R _s . At time t , the noninverting input signal V+ recedes below the inverting input signal V- causing amplifier A ₃ to go low (V _osc low), again turning on the darlington Q, , closing the switch and ending the to interval. Pulling the output of the amplifier low is again aided by leading the positive feedback through capacitor C ₅ sustained through resistor R ₁₆ . The present invention, as disclosed in this embodiment, maintains constant coil current independent of source voltage or temperature variations because feedback is realized as a direct analog of coil current. The operating frequency of the oscillator, as well as its duty cycle, varies with changing conditions in order to maintain constant average current in the coil. Coil current fluctuates somewhat as switching transistor Q, cycles on and off. Effects of varying source voltage from nominal 28 volts to 16 volts, as shown in FIG. 8a, is reflected by the changing duty cycle of the oscillator in FIG. 8b. Output signal from the oscillator Vos _C is shown in FIG. 8c, which clarifies the changing duty cycle. Note that in the PNP arrangement for darlington Q _j , V _osc low drives Q _j on.

Implementation of the fast drop-out switch through transistor Q ₂ is substantially identical to that described previously for the voltage-controlled chopper of FIG. 2.

Overall operation of the current-controlled embodiment of the invention is shown in FIG. 7, with coil current shown in trace 700, armature position shown in trace 710, and contact position shown in trace 720. Operation of the embodiment for the pick-up interval and economizing interval is substantially identical to that previously described with regard to FIG. 3.

A third embodiment of the invention is demonstrated in FIG. 11 for an additional voltage-controlled chopper. This embodiment of the invention is substantially identical to that disclosed and described with regard to FIG. 2, however, MOSFET's have been substituted for switching transistors Q _j and Q ₂ . Resistors Rg, R _u and R ₁₃ have been eliminated in this embodiment as unnecessary, and the inverting signal input for the gate amplifier A ₂ is provided directly from the reference voltage ^ _EF , eUminating resistors Ro, and R ₁₀ . An additional zener diode D ₇ has been added to the circuit to accommodate voltage regulation for the MOSFET operating as switch Q ₂ . Values for the various components in the circuit are identified on the drawing, and amplifiers implemented for A _1? A ₂ and A ₃ are 1/4 LM339's with pin inputs as identified in the drawing.

Having now described the invention in detail as required by the patent statutes, those skilled in the art will recognize modifications and substitutions to the specific embodiments disclosed to accommodate particular applications. Such substitutions and modifications are within the scope and intent of the present invention as defined in the following claims.