Locked Loop

BACKGROUND OF THE INVENTION

[0001] Phase, frequency and frequency derivative are parameters independent of energy of the input signal (i.e., non-energy parameters). There are known methods of estimating signal non-energy parameters based on processing of variables received from a phase-lock loop (PLL).

[0002] US Patent No. 7869554, entitled "Phase/frequency estimator-based phase locked loop", discloses an apparatus and methods described in use a PLL and provided a phase estimation of the input signal from which signal frequency is estimated by a derivative function and low pass filtering.

[0003] US Patent No. 3895294, entitled "Phase change measuring circuit", discloses a device for measuring the phase change of an input signal over a specified period comprising a phase-locked tracking filter including a high frequency voltage-controlled oscillator (VCO), a frequency divider to give local oscillator signal at the same frequency as the input signal and a counter counting cycles of the VCO whose phase change in any period is N times the input phase change to allow liN period resolution, N being an arbitrary integer. The phase change measuring circuit thus allows phase measurement with a resolution within a small fraction of one cycle.

[0004] US Patent No. 7222035, entitled "Method and apparatus for determining changing signal frequency", discloses a method and apparatus which include a PLL having a numerically controlled oscillator (NCO) and a filter of frequency estimates (FFE). The PLL tracks the changing signal frequency and outputs non-smoothed frequency estimates into the FFE. The FFE then smoothes noise in the signal to produce a more accurate smoothed frequency estimate of the input signal.

[0005] US Patent Publication No. 20140072084, entitled "Digital system and method of estimating quasi-harmonic signal non-energy parameters using a digital Phase Locked Loop", discloses a digital system and method of measuring (estimating) non-energy parameters of the signal (phase, frequency and frequency rate) received in additive mixture with Gaussian noise. The first embodiment of the measuring system consists of a PLL system tracking variable signal frequency, a block of NCO full phase computation (OFPC), a block of signal phase preliminary estimation (SPPE) and a first type adaptive filter filtering the signal from the output of SPPE. The second embodiment of the invention has no block SPPE, and NCO full phase is fed to the input of a second type adaptive filter.

[0006] A DPLL described in US Patent No. 4771250 generates signal phase which is an approximation of the phase of the received signal with a linear estimator. The effect of a complication associated with non-zero transport delays related to the DPLL is then compensated by a predictor. The estimator provides recursive estimates of phase, frequency, and higher order derivatives, while the predictor compensates for transport lag inherent in the loop.

[0007] However, the above references, as well as other conventional methods of measuring non- energy signal parameters using PLL are not adaptive to the jerking motion (when, for example, the acceleration varies linearly in time) of the receiver, or adapt to it by expanding the bandwidths of PLL or using smoothing filters. It is not possible to completely eliminate the dynamic measurement errors using conventional methods.

[0008] Unlike the methods above, the present invention enables to obtain accurate phase estimates of the input signal and its derivatives by correcting the preliminary estimates at sites with jerking motion and by an additional filtering of phase estimates at sites without such jerking motion.

[0009] The present invention can be used in receivers of various navigation systems, such as GPS, GLONASS and GALILEO, which provide precise measurements of signal phase at different rates of frequency change, as well as systems using digital PLLs for speed measurements.

Summary of the Invention

[0010] Accordingly, the present invention is related to a system of estimating quasi-harmonic signal non-energy parameters using a digital Phase Locked Loop that substantially obviates one or more of the disadvantages of the related art.

[0011] In one embodiment, a system for estimating parameters of an input signal includes (a) a digital phase locked loop (PLL) that tracks the input signal and includes:

i) a phase discriminator (PD) that determines a phase difference between the input signal and reference signals; ii) a loop filter (LF) with a control period T _c operating based on:

round (<p L ^L F ^t I KNCO

/A _y ), round (sf I Δ ^Ν ^° I T _c ), where cc ^LF , p n ^LF , γ„ ^L are constants,

d

Z _t is a PD output;

r

<Pi is a phase code for a Numerically Controlled Oscillator (NCO), f is a frequency code for the NCO, is a phase step size in the NCO,

A. NCO

is a frequency step size in the NCO, and

round (.) is an operation of numerical rounding; iii) the NCO having frequency and phase control using φ ₍ and /,· , wherein an output of the NCO is connected to a reference input of the PD;

(b) a calculation of full phase (CFP) block that inputs φ and f and that operates based

NCO NCO , r \NCO , rr NCO rp

on φ _ι = <Pi-i + <Pi · Δ + · Α _ω -T _c - (c) a low-pass filter (LPF) inputting Z _t ;

a preliminary estimation of signal parameters (PESP) block that inputs Ψ _ί " ^"0 , fi and ^s ₃ and outputs a preliminary estimate for a signal phase

*c,E ^c '^

Ψί and a preliminary estimate for a signal frequency ^ω based on: ώΓ ^£ =2»· ₁ ^Γ -*Γ (2·7 ); (d) a jerk-corrections of preliminary estimates (JCPE) block inputting Z _i , wherein the

^Λ c

JCPE block outputs an estimate for a signal phase Ψι and an estimate for a signal

Λ c

frequency based on:

CO;

C = \ - a ^LF /2 + β'" / l2 + r ^LI Ί24 _,

C„= (a ^LF - fi ^LF / 2 - y ^LF / 6) /T _c , _md

T _A is a threshold.

[0012] Additional features and advantages of the invention will be set forth in the description that follows, and in part will be apparent from the description, or may be learned by practice of the invention. The advantages of the invention will be realized and attained by the structure particularly pointed out in the written description and claims hereof as well as the appended drawings.

[0013] It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are intended to provide further explanation of the invention as claimed.

Brief Description of the Attached Figures

[0014] The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention.

[0015] In the drawings:

[0016] FIG. 1 is a block diagram for a first embodiment of the invention.

[0017] FIG. 2 is an example of an arctangent-type phase discriminator (PD).

[0018] FIG. 3 is a block diagram for a second embodiment of the invention.

[0019] FIG. 4 illustrates a jerk as a function of time. Detailed Description of the Preferred Embodiments

[0020] Reference will now be made in detail to the preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings.

[0021] In embodiments of the present invention, adaptation to the nature of the movement of the receiver is made not by changing the parameters of FAP, but by changing the algorithm for estimating the signal parameters. For this purpose, corrections that compensate for the dynamic measurement errors during jerking motion are produced.

[0022] FIG. 1 shows a block-diagram of the first embodiment of the invention. The measuring system shown in this figure is based on the use of variables of a PLL for calculating preliminary estimates of non-energy signal parameters (i.e., phase, frequency and frequency derivative (frequency drift)) and calculating the corrections when there is a spurt frequency caused by a receiver's jerking motion. The measuring system comprises a digital PLL (101) that has the following main components: a phase discriminator (PD) (102), loop filter (LF) (103), and numerically-controlled oscillator (NCO) (103) with frequency and phase control. Samples U™ ¹ of an analog process 1/" ^α {ί) at a sampling frequency f _s are fed to the discriminator input. An analog process

U ^ma (t) = U ^c (t) + U"(t) representing an additive mixture of quasi-harmonic signal

U ^c (t) and Gaussian noise U"(f) . Desired signal U ^c (t) is equal to where A ^c is the amplitude of the signal, φ ^€ (ί) = j f(t) - dt+ ⁽ p ₀ is the signal phase [in radians], of{t) is the signal frequency [in radian/s], <p ₀ is the initial signal phase [in radian].

[0023] Signal phase , signal frequency f(t) and frequency derivative of(t) should be estimated (measured).

[0024] A loop filter (LF) operates with a control period _c on the basis of recurrence equations:

φ; = round (<p ^LF /Δ* ^∞ ),

f,' = round (s? /A ^co /T _c ),

LF nLF LF

where Ct , p , γ are constant transfer coefficients,

Z _{ ^d is the PD output;

<Pi is the phase code for NCO, f is the frequency code for NCO,

K NCO

Ά _φ is the phase step size (radian) in the NCO, is the frequency step size (radian/s) in the NCO,

round 0 is the operation of a numerical rounding.

[0025] A numerically controlled oscillator (NCO) (104) has frequency and phase control. The phase input of the NCO is connected to the phase output φ of the loop filter (LF) and the frequency input of the NCO is connected to the frequency output f of the LF (103); wherein a complex output of a NCO connected to a reference input of a PD (102).

[0026] FIG. 2 shows an example of PD. Input samples U™ ^a are multiplied by quadrature samples (if _y Qf) from the NCO,

Γ? = A ^NC0 .cos{(p: ^>NCO ) 1

Qf = A ^NCO -s ((p ^>NCO )

[0027] where A ^NC0 is the sample amplitude, and φ ' ^Ν€ ° is the wrapped phase (i.e., 0 <φ ' ^Ν °° <+2π) of NCO in radians. Multiplication results

j mr jmix jref

n n n

Q ^mr zzz J ^mix . Q ^re f [0028] are fed to the input of low-pass filters, which are typically the reset accumulators ^ -l with frequency F _c « f _s . The reset frequency of the accumulators F _c is the control frequency in the PLL, for example, F _c = 50 Hz . . .1000 Hz; f _s = \0 MHz . . . 100 MHz. The outputs of the reset accumulators are

m=N _C m=N.

∑ η+(;-ΐ)·Λτ, and 2, ^_ ∑2 'm» +(i-\)-N _s

w=l m=l

The output of a phase discriminator is

[0029] : zf = arctan (g. / /. ) [in radians].

[0030] Further, the signal zf from the PD output is inputted to the loop filter (LF) (FIG. 2), which operates with a control period T _c = N f _s on the basis of recurrence equation:

. LF NCO

φ = round (<p ^Lt /A^ ), f = round {sf l ^ ^co I T _c ),

LF oLF LF

where CC , p , γ are constant transfer coefficients, Ί is the PD output;

⁽ Pi is the phase code for the NCO, f is the frequency code for the NCO,

\NCO

φ is the phase step size (radian) in the NCO,

_A NCO

ω is the frequency step size (radian/s) in the NCO, and

round (.) is the operation of a numerical rounding. [0031] Digital phase samples ψ,- are fed to the NCO phase control input and abruptly change its

, _N NCO A. NCO K NCO .

phase by the corresponding value ^ψ, — Ψι ' ^ _φ , where ^ _φ is the phase step size. Samples ft (frequency codes) are delivered to the NCO frequency input and determine its frequency "7 ^— J ' ^ _ω , where Δ _β is the frequency step size

[radian/s] in the NCO.

[0032] The measuring system (see FIG. 1) comprises also:

[0033] block (105) for calculation of full phase (CFP) of NCO, coupled with the LF outputs, operates on the basis of equation

[0034] block (106) - a low-pass filter (LPF) coupled with an output Z _{ of a PD;

[0035] block (107) - a block for preliminary estimation of signal parameters (PESP) coupled by its inputs with: the phase output ⁽ p? ^C ° of a block for CFP of a NCO, the frequency output f of the loop filter,

γ

the output S _i of the loop filter;

where a block for PESP operates on the basis of equations:

where

~ c, E

ψ _\ is the preliminary estimate for a signal phase [in radians], A ^C _' ^E is the preliminary estimate for a signal frequency [radian/s], ωtΐ ^c,E is the preliminary estimate for a signal frequency derivative [radian/s 2 ];

A T

[0036] block (108) is a threshold unit coupled with an output Z _t of a LPF; where an output J, of a threshold unit is given by the formula:

J _; = true, if zf > T _A ,

J _i = false, if zf ≤ T _A ,

[0037] here T _A is a threshold ; the threshold value is set equal to (3...5) · RMS(zf ) . [0038] block (109) - a block for jerk-corrections of preliminary estimates (JCPE) coupled with

A T

an output of a LPF and with an output J, of a threshold unit; where the block JCPE operates on the basis of equations:

where

^Λ c

(Pi is the estimate for a signal phase [in radians],

^Λ c

®i is the estimate for a signal frequency [in radian/s],

* C

i is the estimate for a signal frequency derivative [in radian/s ² ],

C ₉ = l- ^LF / 2 + fi ^LF / \2 + r ^LF 124 , C _a = {ct ^LF - p ^LF I 2 - Y ^LF I 6) I T _C ,

ο _ώ = β" ^ρ ιτ

[0039] FIG. 3 shows a block-diagram of the second embodiment of the invention. The measuring system is based on the use of variables of a PLL for calculating preliminary estimates of signal parameters (phase, frequency and frequency derivative) and calculating the corrections when there is a spurt frequency caused by a receiver jerking motion. When a frequency spurt is absent, the parameter estimates are obtained by 3rd order tracking filter, whose input is fed by a preliminary assessment of phase. The measuring system FIG. 3 comprises blocks (301), (302), (303), (304), (305), (306), (307), (308), (309). All of these blocks and their connection are the same as in the first embodiment respectively (101), (102), (103), (104), (105), (106), (107), (108), (109), except that the output of unit (308) does not feed unit (309). [0040] Block (309) for jerk-corrections of preliminary estimates (JCPE) coupled with an output Ζ,· of a LPF and operates on the basis of equations:

[0041] where are, respectively, estimates with jerk-corrections for a phase

[in radians], frequency [in radian/s] and frequency derivative [radian/s ] of a signal.

[0042] Block (309) for jerk-corrections of preliminary estimates (JCPE) reduces dynamic error of estimates due to frequency spurts, but it increases fluctuation errors of estimates. The measuring system, see FIG. 3, comprises a 3rd order tracking filter of phase (TFP) (310) to reduce the fluctuation errors in the absence of frequency spurt; wherein a TFP bandwidth is less than a PLL bandwidth. Block TFP (310) coupled with the outputs

~ c,E

Ψι of the PESP, operates on the basis of recurrence equations:

[0043] where CC ^T , β ^Τ , y ^T are constant transfer coefficients of the TFP.

[0044] Block (311) decides on which group of estimates for signal parameters should be taken; this block takes the estimates from the TFP block when there is no jerk, otherwise, it takes the estimates from the JCPE (when there is jerk), i.e. ,

where the estimate for a signal phase [in radians], the estimate for a signal frequency [in radian/s], is the estimate for a signal frequency derivative [in radian/s ].

[0045] FIG. 4 shows an example of motion with jerks. This jerking motion consists of: 0<t <ti and t > are states without movement, t <t <t ₂ and t ₃ < t <t ₄ are states with jerks, when the acceleration varies linearly, t ₂ < t <t ₃ is the state with a constant acceleration.

[0046] Having thus described a preferred embodiment, it should be apparent to those skilled in the art that certain advantages of the described apparatus have been achieved.

[0047] It should also be appreciated that various modifications, adaptations, and alternative embodiments thereof may be made within the scope and spirit of the present invention. The invention is further defined by the following claims.