INTRODUCTION AND BACKGROUND

This invention relates to a drive circuit and method for driving a transformer. More particularly the invention relates to a drive circuit and method for driving a transformer which forms part of an ignition system for an internal combustion engine.

In order to improve emissions in petrol internal combustion engines, the engine needs to be operated with a high exhaust gas recycling (EGR) or with lean air-fuel mixtures. However, when using current spark ignition systems, the combustion stability becomes unacceptable under these conditions. The reason for this is the small volume of gas that is ignited across the small (typically 0.8 mm) spark gap. Several corona ignition systems have been developed and it has been shown that these corona ignition systems have a much better performance, due to a larger volume of gas that is ignited. However, the complexity and cost of the drive circuit used in current corona systems is a barrier for this technology to enter the market. The drive circuit has to drive a corona resonator at high frequency (larger than 1 MHz) and high power (larger than 1 kW) at the correct resonance frequency and be able to cope with arcing. The circuit must also be able to perform reliably in harsh under-hood environment, must be cost effective and not occupy much more volume than the known systems. OBJECT OF THE INVENTION

Accordingly, it is an object of the present invention to provide a drive circuit and method with which the applicant believes the aforementioned disadvantages may at least be alleviated or which may provide a useful alternative for the known circuits and methods.

SUMMARY OF THE INVENTION

According to the invention there is provided a drive circuit for a transformer comprising a primary winding and a secondary winding which are connected in a primary winding circuit and a secondary circuit respectively, the secondary circuit comprising a variable secondary load impedance comprising a resistance and a capacitance, so that the secondary circuit has a secondary resonance frequency, the drive circuit comprising:

a DC current source;

a switching circuit comprising a first terminal, a second terminal and a switching arrangement between the first and second terminals which is switchable between a closed configuration and an open configuration by a signal applied to a control terminal of the switching arrangement;

the current source and the switching arrangement being connected in parallel with one another as well as with the primary winding circuit; and a feedback circuit having first and second input terminals which is also connected in parallel with the primary winding circuit and further comprising an output terminal which is connected to the control terminal of the switching arrangement, to cause the switching arrangement to switch between the open and closed configurations at a frequency equal to the secondary resonance frequency.

The current source may comprise any suitable current source including a voltage to current converter and a circuit comprising an inductor connected to a DC voltage source.

The primary and secondary windings of the transformer may be weakly coupled in that the coupling constant k is preferably such that k < 0.5, more preferably k < 0.4, even more preferably k < 0.3 and most preferably k < 0.2.

The secondary circuit preferably has a high quality factor Q, preferably higher than 50, when driven at resonance.

The feedback circuit may comprise passive components only. Passive components are components that cannot supply energy themselves, are incapable of power gain and of amplifying signals. Such passive components may comprise resistors, inductors, transformers, capacitors and diodes, including zener diodes.

The primary winding circuit may comprise a capacitor in series with the primary winding.

The switching arrangement may comprise at least one power insulated gate controlled switching device, such as a power MOSFET.

According to another aspect of the invention there is provided a method of driving a transformer comprising a primary winding and a secondary winding which are connected in a primary winding circuit and a secondary circuit respectively, the secondary circuit comprising a variable secondary load impedance comprising a resistance and a capacitance, so that the secondary circuit has a secondary resonance frequency, the method comprising:

utilizing a DC current source and a switch which are connected in parallel with one another and with the primary winding circuit;

operating the switch by opening and closing the switch under control of a control signal to cause the secondary circuit to resonate at the secondary resonance frequency;

deriving the control signal from a feedback circuit which is connected to the primary winding, to sense a voltage which is induced in the primary winding by a current in the secondary circuit which current is changing at the secondary resonant frequency, so that the switch is operated at the secondary resonance frequency.

BRIEF DESCRIPTION OF THE ACCOMPANYING DIAGRAMS

The invention will now further be described, by way of example only, with reference to the accompanying diagrams wherein:

figure 1 is a circuit diagram of a first example embodiment of a drive circuit for a transformer;

figure 2 is a diagrammatic representation of an ignition plug in the form of a spark plug comprising a transformer and the drive circuit;

figure 3 is a diagrammatic representation of an ignition plug in the form of a corona plug comprising a transformer and the drive circuit;

figure 4 show wave forms at various points in the circuit in figure 1 ; figure 5 is a diagram showing relevant parts of figure 1 with associated phasor diagrams;

figure 6 is a circuit diagram of a second example embodiment of the transformer and drive circuit;

figure 7 is a circuit diagram of a third example embodiment of the transformer and drive circuit;

figure 8 is another representation of part of the circuit in figure 7; and figure 9 is a circuit diagram of a fourth example embodiment of the transformer and drive circuit.

DESCRIPTION OF A PREFERRED EMBODIMENT OF THE INVENTION

A drive circuit for a transformer 12 is generally designated by the reference numeral 10 in figure 1 .

The transformer 12 comprises a primary winding 14 having an inductance L _p and a secondary winding 16 having an inductance L _s . The primary winding is connected into a primary winding circuit 18 and the secondary winding is connected in a secondary circuit 20 comprising a variable secondary load impedance 21 comprising a secondary capacitance C _s and a secondary resistance R _s . The secondary circuit has a resonance frequency which is referred herein as the secondary resonance frequency. The secondary capacitance may be provided by parasitic capacitance of the secondary winding 16 and in an example application wherein the circuit is used to drive an ignition plug in the form of a spark plug 24 (shown in figure 2) or a corona plug 26 (shown in figure 3) by capacitance between spaced electrodes 28, 30 of the plug. The resistance may be provided by losses in the secondary circuit and by losses in a gas in gap

32 between the electrodes. In the example embodiments of figures 2 and 3, the transformer 12 as well as a major part of the drive circuit 10 are mounted on the plug body 22. The drive circuit 10 comprises a DC current source 34. A switching circuit 36 comprises a first terminal 36.1 , a second terminal 36.2 and a switching arrangement 36.3 comprising at least one switching device between the first and second terminals. The arrangement 36.3 is switchable between a closed configuration and an open configuration by a signal applied to a control terminal 38 of the switching arrangement. The current source 34 and the switching arrangement 36.3 are connected in parallel with one another as well as with the primary winding circuit 18. A feedback circuit 40 comprises first and second input terminals 40.1 , 40.2 and is also connected in parallel with the primary winding circuit 18. An output terminal 40.3 of the feedback circuit is connected to the control terminal 38 of the switching arrangement 36.3, to cause the switching arrangement to switch between the open and closed configurations at a frequency equal to the secondary resonance frequency.

The current source 34 supplies current to the circuit through inductor 50. The purpose of capacitor 52 and inductor 50 is to filter high-frequency feedback from the circuit to the current source 34, so that the voltage across the current source stays almost constant. The current source 34, the switching arrangement 36.3 and the primary winding circuit 18 of the transformer are connected in parallel. A capacitor 54 having a capacitance of C _P is connected in series with the primary winding 14 in the primary winding circuit 18. The capacitor 54 blocks DC currents through the primary winding and when chosen such that C _P L _P ~ L _S C _S it ensures that the voltage across the primary circuit 1 8 and the current through the primary circuit are in phase at the secondary side resonance frequency, as explained in more detail below.

In the example embodiment of figure 1 , the switching arrangement comprises a single power MOSFET 36.3 having a gate providing the control terminal 38, a drain and a source. The drain may be connected to ground which may facilitate mounting of a heat sink for heat exchange, and avoid unwanted heat sink capacitance.

The feedback circuit 40 comprises an inductor 56 having an inductance Lfb, capacitor 58 having a capacitance C¾ and a capacitor 60 having a capacitance C _g . The components of the feedback circuit are all passive components that cannot supply energy themselves, are incapable of power gain and of amplifying signals. Such passive components may comprise any one or more of resistors, inductors, transformers, capacitors and diodes, including zener diodes. The input of the feedback circuit is connected across the switching arrangement and the primary winding circuit and the output 40.3 of the feedback circuit is connected to the gate 38 of the MOSFET 36.3, so that it drives the gate positive and negative. The feedback circuit is designed so that maximum feedback occurs at the secondary side resonance frequency (Lfb x C¾ « L _s x C _s ). The feedback circuit may also introduce a phase shift between its input and output, to compensate for a switching delay of the switching arrangement 36.3 (for example by making Lfb x Cfb slightly smaller than L _s x C _s ). A harmonic circuit 70 may be added in parallel with the switching circuit

36. When the MOSFET 36.3 is switched at the secondary side resonance frequency, it also generates higher-order harmonics. These harmonics are shaped by the harmonic circuit, as will be explained below. The graphs in figure 4 show the secondary output voltage 1 00 over R _s , drain-source voltage (VDS) 1 02 and source current 1 04 for the above circuit. At the resonance frequency, the output voltage 1 00 and the drain- source voltage 1 02 are in phase, while the source current (note the direction of the arrow 1 04 in figure 1 ) is 1 80 degrees out of phase, which means that conventional drain source current is also in phase with 1 00 and 1 02. The harmonic circuit 70 shapes the higher-order harmonics so that the drain-source voltage 1 02 is approximately a square wave. This limits the maximum drain-source voltage to be below the maximum rating of the MOSFET 36.3 used.

Referring to figures 1 and 5, the DC current source 34 and switching arrangement 36 cause an AC current l _P through the primary winding 1 4 which is in phase with the switching of the switching arrangement 36. The AC current l _P through the primary winding 14 causes a changing magnetic field, which induces a voltage V _p , _p over the primary winding 14 and a voltage V _s , _P over the secondary winding 1 6. On the primary side, the induced voltage V , is 90 degrees out of phase with the current l . The voltage V _S , _P induced over the secondary winding 1 6 gives rise to an AC current in the secondary winding. The closer the drive frequency is to the secondary resonance frequency, the larger the secondary current . When the secondary side is driven at its resonance frequency, the secondary side acts like a resistive element (the inductive and capacitive reactances cancel out at resonance) so that the secondary current is 180 degrees out of phase with V _s , _P and therefore also 90 degrees out of phase with the primary current l .

The AC secondary current induces a voltage V , _s over the primary winding which is 90 degrees out of phase with the secondary current and therefore in-phase with the primary current l .

The capacitor 54 serves to cancel the effect of the primary winding so that better feedback may be achieved. The same current l flows through the primary winding 14, but the voltages V , over the primary winding 14 and the capacitor V _c are 1 80 degrees out of phase and hence cancel one another. By choosing a proper value for C , the voltage V , may be cancelled so that the voltage over the primary circuit VDS is dominated by the voltage V _P , _S which is induced over the primary winding 1 4 by the AC current in the secondary circuit 20.

The feedback circuit 40 then uses this in-phase primary circuit voltage VDS to drive the switching arrangement 36. As stated above, a small phase shift may also be introduced by the feedback circuit 40 to compensate for the switching delay of the switching arrangement.

It is important to note that the circuit on the primary side is not resonant, but the secondary circuit 20 is. The primary winding 1 4 and series capacitor 54 do not resonate when driven by a current source. They would have formed a series resonant circuit were they driven by an AC voltage source. The oscillators used in most prior art corona ignition systems make use of a voltage power source to drive a resonator and make use of current feedback to lock the resonance frequency. A disadvantage of these systems is that the output power inherently increases when a spark is formed (that is when the secondary resistance decreases and the current increases), so that sophisticated control techniques and expensive circuitry are required to detect sparking and to control the power delivered. On the other hand, the drive circuit 10 uses a current power source 34 to drive the transformer 12 and voltage feedback from the primary winding to lock on the secondary resonance frequency. In the case of a corona plug 26 which is shown in figure 3, this has the advantage of the output power automatically decreasing when a spark is formed, so that no or little additional control may be necessary.

In the case of a spark plug 24 as shown in figure 2, the plug and high- voltage transformer 12 are driven at the resonant frequency in a high- power mode. Direct feedback from the primary is used to lock on the secondary side resonance frequency. When the secondary side becomes non-resonant due to a spark discharge, it inherently follows a more efficient lower energy transfer mode. In this low-power mode it is still self- oscillating and the high-voltage transformer becomes an AC current source (> 100 imA) that sustains the spark. When the secondary side becomes resonant again, it returns to the efficient, high power, resonant mode.

The example embodiment of the drive circuit 10 shown in figure 1 , comprises only a few inexpensive components. This drive circuit makes it possible to make a corona ignition system at about similar cost as a conventional spark ignition system and occupying about a similar volume. In the example embodiment of figure 1 , only one inexpensive switching device is used. By switching at the right phase, the switching circuit is highly efficient with very little switching loss. Only direct (analog) feedback is used and no digital control or integrated circuits (ICs) are used. This makes the drive circuit simple and reliable. The components of the circuit are selected to be able to reliably operate at under-hood conditions such as high temperatures, thermal cycling and vibrations.

In figure 6 these is shown another example embodiment of the drive circuit 10 and transformer 12 wherein the circuit is implemented in balanced configuration and wherein the same reference numerals as above are used for like parts. The drains of the MOSFET's may be connected to one another and to ground which may facilitate mounting of a heat sink for heat exchange purposes, and avoid unwanted heat sink capacitance. In the example embodiment of figure 7, two primary windings parts 14.1 and 14.2 are used to substitute the current source inductances 50.1 and 50.2 in figure 6. Although the two coupled windings 14.1 and 14.2 may be seen as in series with the current source 34, they have common-mode and differential-mode currents. The AC current through the windings are in the same direction (in-phase) and therefore common-mode currents. The common-mode currents do not flow through the current source. Both windings also have DC current components, which are in opposite directions and therefore the DC currents are differential-mode and in series with the current source. The common-mode and differential-mode currents may be separated by splitting the current source into two as shown at 80.1 and 80.2 in figure 8. This is equivalent to the circuit in figure 7, except that the primary windings do not have a differential current any more. In this case it becomes clear that for the AC current in the primary windings, the primary windings 14.1 and 14.2 are in parallel with the current source 34.

In another example embodiment of the drive circuit 10 and transformer 12 shown in figure 9, the primary winding 14 also serves as an inductor of the feedback circuit 40.