FIELD OF INVENTION The present invention relates to the recycling (i.e. the recovery and re-use) of electromagnetic field energy. Energy from a collapsing magnetic field in an inductive device is recovered and stored on a capacitor and, in a subsequent cycle, is returned to the inductive device, in addition to energy from a source, to establish a magnetic field at the inductive device. The invention is described with reference to switched reluctance motors.

BACKGROUND

A common aspect of conventional inductive devices, such as motors, linear actuators, solenoids, transformers and induction coils, is that they rely on the building of a magnetic field to perform motoring, transforming or inducing action, or a magnetic attraction or repulsion. The energy built up or contained within the magnetic field in these instances is substantial and significant energy remains even after work has been performed. Standard designs of motors, solenoids, linear actuators, pulse or fly-back transformers and induction coils do not incorporate, as a general rule, optimised field energy recovery on the primary or secondary windings.

The propensity of the magnetic field to remain once built up in inductive devices is often treated to some degree as a nuisance. Many control strategies are used to deplete or diminish the magnetic field, after it has been used to perform work, in a way that minimises damage to the inductive device or to other circuit components from excessive inductive voltage spikes and the like. Depletion of the magnetic field, sometimes referred to as 'de-fluxing', has been achieved by capacitor dump circuits, diode clamping, applying high reverse voltages, by use of DC link capacitors, and by other field control techniques.

In some cases, rather than merely dissipating the energy and to avoid potentially destructive voltages, the energy has been recovered for later re-use. Typically, energy from a collapsing magnetic field has been returned to a capacitor, such as a supply, DC link capacitor or supplementary capacitor, for re-use when demand is next placed on the supply. Some prior art magnetic field energy recovery circuits use a capacitor to store energy recovered from a collapsing magnetic field, and later re-use the energy stored on the capacitor to establish a magnetic field, however the capacitor is often relatively large and acts as an energy reservoir.

SUMMARY OF INVENTION

In broad terms a first aspect of the invention comprises a method of operating an inductive device having a winding connected to a controlled switching circuit, the method comprising repetitively configuring the controlled switching circuit successively in first, second and third configurations;

the controlled switching circuit, when configured in the first configuration, electrically coupling a source of electrical energy to the winding to establish a magnetic field in the winding; the controlled switching circuit, when configured in the second configuration, electrically coupling a capacitor to the winding to discharge the capacitor into the winding to maintain or augment the magnetic field established in the winding during the previous configuration of the switching circuit in the first configuration; and

the controlled switching circuit, when configured in the third configuration, electrically coupling the capacitor to the winding to charge the capacitor by a current induced in the winding by collapse of the magnetic field in the winding. Preferably, the switching circuit comprises a first transistor switch and a second transistor switch, the first configuration is established by turning on the first transistor switch to couple the source of electrical energy to the winding, the second configuration is established by turning on the second transistor switch to couple the capacitor to the winding, the third configuration is established by turning off both the first transistor switch and the second transistor switch simultaneously, and the current induced in the winding by the collapse of the magnetic field in the winding flows through semiconductor diodes which couple the capacitor to the winding.

Preferably the inductive device is a reluctance motor having a rotor and a stator. Preferably, the successive switching of the first transistor switch and the second transistor switch is repeated with a switching repetition interval synchronised with rotation of the rotor, the first transistor switch is turned on at a reference time dynamically synchronised to the rotor position, the second transistor switch is turned on at the end of a first controlled delay period after the reference time, both the first transistor switch and the second transistor switch are turned off at the end of a second controlled delay period after the reference time, the first controlled delay period is between 10% and 20% of the switching repetition interval, and the second controlled delay is between 35% and 45% of the switching repetition interval.

Preferably, the inductance of the winding is dependent on the rotational position of the rotor with respect to the stator, the inductance of the winding varying between a minimum value and a maximum value, and the first configuration is established when the inductance of the winding is substantially at the minimum value. More preferably, the third configuration is established before the inductance of the winding next reaches the maximum value.

Preferably, in the first configuration, the energy source is coupled to the motor winding through an inductive and capacitive impedance circuit which restricts delivery of energy from the source to the motor winding.

Preferably the inductive device is a reluctance motor having a rotor and a stator.

Preferably the successive switching of the first transistor switch and the second transistor switch is repeated with a switching repetition interval synchronised with rotation of the rotor, the first transistor switch is turned on at a reference time dynamically synchronised to the rotor position, the second transistor switch is turned on at the end of a first controlled delay period after the reference time, both the first transistor switch and the second transistor switch are turned off at the end of a second controlled delay period after the reference time. Preferably the first controlled delay period is between 10% and 20% of the switching repetition interval.

Preferably the second controlled delay is between 35% and 50% of the switching repetition interval.

Preferably the inductance of the winding is dependent on the rotational position of the rotor with respect to the stator, the inductance of the winding varying between a minimum value and a maximum value, and the first configuration is established when the inductance of the motor winding is substantially at the minimum value. Preferably the third configuration is established before the inductance of the winding next reaches the maximum value. Preferably the controlled switching circuit further comprises a second capacitor configured such that, in the second configuration, the capacitor and the second capacitor discharge in series.

Preferably in the third configuration, the switching circuit charges the second capacitor in a charge pump circuit configuration.

Preferably in the third configuration, the switching circuit charges the capacitor and the second capacitor by recovering energy from the magnetic field.

Preferably the method further comprises providing a controller configured to receive an input from the inductive device indicative of position and output a signal operable to change the state of the switching circuit.

Preferably in the first configuration, the energy source is coupled to the winding through an inductive and capacitive impedance circuit which restricts delivery of energy from the source to the winding.

Preferably the winding is part of at least one of a motor, linear actuator or solenoid.

In broad terms a second aspect of the invention comprises control system for operating an electrical circuit, the system comprising a controller configured to receive an input from one or more inductive devices and output one or more signals to a switching circuit to repetitively configure the switching circuit successively in first, second and third configurations;

the controlled switching circuit, when configured in the first configuration, operable to electrically couple a source of electrical energy to the winding to establish a magnetic field in the winding; the controlled switching circuit, when configured in the second configuration, operable to electrically couple a capacitor to the winding to discharge the capacitor into the winding to maintain or augment the magnetic field established in the winding during the previous configuration of the switching circuit in the first configuration; and the controlled switching circuit, when configured in the third configuration, operable to electrically couple the capacitor to the winding to charge the capacitor by a current induced in the winding by collapse of the magnetic field in the winding.

Preferably the switching circuit comprises a first transistor switch and a second transistor switch, the first configuration is established by turning on the first transistor switch to couple the source of electrical energy to the winding, the second configuration is established by turning on the second transistor switch to couple the capacitor to the winding, the third configuration is established by turning off both the first transistor switch and the second transistor switch simultaneously, and the current induced in the winding by the collapse of the magnetic field in the winding flows through semiconductor diodes which couple the capacitor to the winding.

Preferably the inductive device is a reluctance motor having a rotor and a stator.

Preferably the successive switching of the first transistor switch and the second transistor switch is repeated with a switching repetition interval synchronised with rotation of the rotor, the first transistor switch is turned on at a reference time dynamically synchronised to the rotor position, the second transistor switch is turned on at the end of a first controlled delay period after the reference time, both the first transistor switch and the second transistor switch are turned off at the end of a second controlled delay period after the reference time.

Preferably the first controlled delay period is between 10% and 20% of the switching repetition interval.

Preferably the second controlled delay is between 35% and 50% of the switching repetition interval.

Preferably the inductance of the winding is dependent on the rotational position of the rotor with respect to the stator, the inductance of the winding varying between a minimum value and a maximum value, and the first configuration is established when the inductance of the motor winding is substantially at the minimum value.

Preferably the third configuration is established before the inductance of the winding next reaches the maximum value. Preferably the controlled switching circuit further comprises a second capacitor configured such that, in the second configuration, the capacitor and the second capacitor discharge in series. Preferably in the third configuration the switching circuit charges the second capacitor in a charge pump circuit configuration.

Preferably in the third configuration the switching circuit charges the capacitor and the second capacitor by recovering energy from the magnetic field.

Preferably the controller is further configured to receive an input from the inductive device indicative of position and output a signal operable to change the state of the switching circuit.

Preferably in the first configuration the energy source is coupled to the winding through an inductive and capacitive impedance circuit which restricts delivery of energy from the source to the winding.

Preferably the winding is part of at least one of a motor, linear actuator or solenoid. In broad terms a second aspect of the invention comprises an appliance comprising an inductive device with a winding and a controlled switching circuit operable to drive the inductive device, the controlled switching circuit comprising first, second and third configurations;

the controlled switching circuit, when configured in the first configuration, operable to electrically couple a source of electrical energy to the winding to establish a magnetic field in the winding;

the controlled switching circuit, when configured in the second configuration, operable to electrically couple a capacitor to the winding to discharge the capacitor into the winding to maintain or augment the magnetic field established in the winding during the previous configuration of the switching circuit in the first configuration; and

the controlled switching circuit, when configured in the third configuration, operable to electrically couple the capacitor to the winding to charge the capacitor by a current induced in the winding by collapse of the magnetic field in the winding. Preferably the appliance further comprises a control system for operating the electrical circuit, the system comprising a controller configured to receive an input from the inductive device and output one or more signals to the switching circuit to repetitively configure the switching circuit successively in the first, second and third configurations.

Preferably the inductive device is a reluctance motor having a rotor and a stator.

Preferably the successive switching of the first transistor switch and the second transistor switch is repeated with a switching repetition interval synchronised with rotation of the rotor, the first transistor switch is turned on at a reference time dynamically synchronised to the rotor position, the second transistor switch is turned on at the end of a first controlled delay period after the reference time, both the first transistor switch and the second transistor switch are turned off at the end of a second controlled delay period after the reference time. Preferably the first controlled delay period is between 10% and 20% of the switching repetition interval.

Preferably the second controlled delay is between 35% and 50% of the switching repetition interval.

Preferably the inductance of the winding is dependent on the rotational position of the rotor with respect to the stator, the inductance of the winding varying between a minimum value and a maximum value, and the first configuration is established when the inductance of the motor winding is substantially at the minimum value.

Preferably the third configuration is established before the inductance of the winding next reaches the maximum value.

Preferably the controlled switching circuit further comprises a second capacitor configured such that, in the second configuration, the capacitor and the second capacitor discharge in series.

Preferably in the third configuration the switching circuit charges the second capacitor in a charge pump circuit configuration. Preferably in the third configuration, the switching circuit charges the capacitor and the second capacitor by recovering energy from the magnetic field. Preferably a controller configured to receive an input from the inductive device indicative of position and output a signal operable to change the state of the switching circuit.

Preferably, in the first configuration, the energy source is coupled to the winding through an inductive and capacitive impedance circuit which restricts delivery of energy from the source to the winding.

Preferably the appliance is at least one of a traction drive, air compressor, liquid pump, wheel loader, bulk handler, vacuum cleaner, washing machine, machine tool, mining equipment, air conditioning unit or aerospace device.

This invention may also be said broadly to consist in the parts, elements, features and method steps referred to or indicated in the specification of the application, individually or collectively, and any or all combinations of any two or more of said parts, elements, features, or method steps and where specific integers are mentioned herein which have known equivalents in the art to which this invention relates, such known equivalents are deemed to be incorporated herein as if individually set forth.

As used herein the term "and/or" means "and" or "or", or both. As used herein "(s)" following a noun means the plural and/ or singular forms of the noun.

The term 'inductor' where used in this specification means a passive component that is incorporated in a circuit primarily for its property of inductance. The term 'inductive device' where used in this specification means a device having inductance but which is incorporated in a circuit primarily for establishing a magnetic field to perform, for example, a motoring action, or a magnetic attraction or repulsion. Inductive devices include, but are not limited to, electromechanical devices such as electromagnetic motors, linear actuator coils and solenoids or, for example, pulse or fly-back transformers.

References herein to a current induced in an inductive device during collapse of a magnetic field can be understood as referring to a current that is driven by a voltage induced in the inductive device by collapse of the magnetic field through the winding inductance of the device.

BRIEF DESCRIPTION OF DRAWINGS The invention will be further described by way of example only and without intending to be limiting with reference to the following drawings, wherein:

Figure 1A shows a circuit illustrating a first embodiment of the invention. Figure IB shows the current waveform of the first embodiment of the invention.

Figure 1C shows the voltage waveform of the first embodiment of the invention.

Figure ID shows the switch timing of the first embodiment of the invention.

Figure 2A shows a circuit illustrating a second embodiment of the invention. Figure 2B shows the current waveform of the second embodiment of the invention. Figure 2C shows the voltage waveform of the second embodiment of the invention. Figure 2D shows the switch timing of the second embodiment of the invention. Figure 3A shows a circuit illustrating a third embodiment of the invention.

Figure 3B shows the current waveform of the third embodiment of the invention. Figure 3C shows the voltage waveform of the third embodiment of the invention. Figure 3D shows the switch timing of the third embodiment of the invention. DETAILED DESCRIPTION

The invention seeks to improve and optimise the use of recovered electromagnetic field energy. Optimisation of the recovery of field energy from a reluctance type motor winding has been found to depend on maximising the power transfer of the winding inductance and the energy recovery capacitor in line with the principles of the Maximum Power Transfer theorem; the right sizing and operation of the capacitor on which the recovered energy is stored. The efficiency of the recovery is improved by sizing the recovery capacitor so that it meets the conditions of maximum power transfer and begins the recovery cycle with a low voltage and charges up to a high voltage which is often advantageously higher than the supply voltage. As the voltage on the recovery capacitor rises near the end of the recovery period, the capacitor presents a high reverse voltage to the voltage across the motor winding, therefore steepening the fall of the recovery current. This efficiently recovers field energy and creates rapid motor de-fluxing which is desirable, especially at high motor speeds. It also eliminates or minimises a generation current "tail" during the period of falling motor winding inductance that would otherwise produce undesirable back torques in the motor.

Motor efficiency improvements are achieved from optimising the recovery of electromagnetic field energy from the motor winding inductance and then the re-introduction of this energy back into the motor at the most appropriate point in the motors magnetic cycle. Whereas other methods of motor efficiency improvement focus on loss reduction (reducing Joule heating) or from improving motor magnetic materials (reducing hysteresis losses) which would typically be expected to give a 2-3% increase in motor efficiency.

The invention relates to operation of controlled switching circuits for driving electromagnetic devices, and particularly for driving devices that exhibit a change in inductance during operation of the devices. For example, a switched reluctance motor or a solenoid-driven actuator or pump presents a winding inductance that varies over a particular range during operation. The invention relates to such circuits incorporating recovery of energy from a collapsing magnetic field, the storage of that recovered energy as charge on a capacitor, the re-establishment of the magnetic field using energy drawn from an energy source, and the subsequent use of the stored recovered energy being delivered into the winding inductance to maintain and augment the established magnetic field. The controlled switching circuit effectively connects a capacitor and a winding in various circuit configurations to carry out the energy transfers.

Energy recovered from the motor winding inductance is re-used when the winding is next magnetised. On each magnetisation cycle, the winding is initially magnetised by energy from a supply. The supply energy is delivered to the winding through a restricting circuit to limit or control the amount of energy drawn from the supply. After the magnetic field has been established using energy from the supply, energy recovered from the previous magnetic cycle is injected into the winding to augment the established magnetic field at the winding. Field energy of the winding is then recovered and stored ready for use in the next following injection cycle.

It has further been discovered that to optimise the driving of a reluctance motor, field energy recovered at high voltage is best re-injected into the motor winding when the back electro-motive force (BEMF) from the winding inductance change is at its highest, and near the optimum torque position of the rotor. Injection of recovered energy when the BEMF in the motor winding is high makes best use of the high voltage achieved across the recovery capacitor. Also, injection of recovered energy when the motor rotor is near the optimum torque position, typically when the winding inductance is approximately 1/3 up the rising slope of the winding inductance waveform, makes best use of the high energy on the recovery capacitor to produce a higher winding current and/ or consequently aid in the generation of additional motor torque or at least maintaining motor torque with less energy input. It has also been discovered that to aid the efficient use of the recovered energy and reduce current drawn from the energy source supplying the motor, the supply which provides the first part of the motor current pulse can be choked or restricted. This may be done by suitably sizing a DC link reservoir capacitor connected across the supply and by coupling the supply to the motor winding through an impedance network. Significant reductions in motor supply current typically ranging from 5-15% or more can be achieved.

In applications of the current invention driving a variable reluctance motor, the configuration of the switching circuit to magnetise a winding of the motor is preferably commenced at a synchronisation time derived from a pick-up or sensor device monitoring the angular position of the rotor of the motor or from sensor-less position detection.

The duration or period that the switching circuit maintains the magnetising configuration is actively controlled by controlled switches.

Semiconductor diodes are used in the current invention to make automatic changes to the switching circuit configurations. For example, semiconductor diodes are used in the configuration of the switching circuit in a recovery configuration by which current induced by collapse of the magnetic field at a motor winding is directed to charge a recovery capacitor for storage of energy recovered from the collapsing magnetic field.

Drawing conventions

It should be noted that in the accompanying figures the connection between wires is shown with a dot. Wires that intersect but have no dot at the intersection are not connected.

In general, circuit components labelled similarly throughout the following description and in the accompanying figures provide corresponding functions. For example, in each of the three embodiments described, the transistors Q2A and Q2B control the configuration of the switching circuit to establish a magnetic field at respective motor windings that are shown in the figures as inductances L2A and L2B. Similarly, the controlled switches Q1A and Q1B control the configuration of the switching circuit to re-use energy stored on respective recovery capacitors C2A, C3A, and C2B, C3B by injecting the stored energy into the respective motor windings A and B.

The motors used to perform experimental analysis of each of the three embodiments described herein have two phase windings, A and B, each winding comprising two stator pole windings. The circuit components associated with each of the windings are respectively identified with the suffixes A and B. Suffixes A and B have been omitted from references to corresponding components where description applies to each of the circuits driving the two motor windings A and B. The descriptions are to be construed as applying similarly to each circuit. Transistor switches and switching controller

The controlled switches in the circuits shown in the accompanying figures are transistor switches controlled by any suitable switching controller. For example, the controller may be a microprocessor, microcontroller or other suitable digital logic or programmable device that can provide the switching transistors with control pulses or signals of the required amplitude and timing.

One controller, particularly suitable for motor development and efficiency testing purposes, is based on a Parallax Propeller 32 bit microcontroller, which has 8 independent processors that communicate via a central hub, each having 2 counter/ timer systems, and each able to directly drive any of 32 input-output ports. In practice, there are many microcontrollers, such as the ST7MC, PIC18FXX31, TMS320C24, and HT46R47-H, that would be suitable for motor switching control once the optimum motor performance parameters have been determined from real-time testing.

Software configures the ports either as outputs to drive the transistor switches via opto-couplers, or as inputs to sense the rotor position via an optical sensor. A software system sequences the motor through each revolution, stepping through the transistor switching events to achieve correct operation. The software provides a table-based set of control events that are used to provide the desired control sequence. These are dynamically adjusted using a Propellor™ microcontroller test platform to achieve the maximum motor timing and performance and can then be compiled into a table-based software program. The gate control signals provided to the switches by the controller will be responsive to one or more operating conditions of the motor. For example, the timing of the control signals provided to the switches may be responsive to the rotational speed or position of the rotor of the motor, or of the motor shaft, or the inductance of the stator winding for a sensor less motor, or a component driven by the motor.

In the embodiments described, the switching control sequence for each revolution is timed relative to a datum point provided by the switching edge of a slotted rotor disc and a suitable optical sensor, such as the OPB625, that detects the rotor position. Switching of the transistor switches occurs relative to this datum point. The timing of switching may be varied with motor speed, preferably being advanced at higher rotor speeds. This advance is typically applied progressively and lineady above a transition speed for high speed motors (for example the 81,000 rpm Dyson Airblade™ hand dryer motor operated by the circuit shown in Figure 2). The transition is typically chosen in high speed motors to be at half the motor's full speed. The switching of the transistor switches is repeated for each pole to complete a full rotor revolution.

The controlled transistor switches may be any switch suitable for the currents and voltages encountered, and have suitable switch characteristics such as switching speed, low 'on' or closed resistance, and high 'off or open resistance. Hybrid insulated gate bipolar transistors (IGBTs), for example the Power Mos APT15GP90 from Advanced Power Technology, and metal oxide semiconductor field effect transistor (MOSFET) switches, for example the IRG4PSH71K and IRFK4HE50 from International Rectifier, have been found suitable for some applications of the circuits and methods described below. Other suitable solid state switching devices may be used.

In practical circuits according to some embodiments of the invention, and particularly circuits operating at higher switching frequencies, the controlled switches are preferably matched transistors having closely similar switching speeds, i.e. rise and fall times and switching turn-on and turn-off delay times.

MOSFET gate drivers

The power MOSFETs are driven by gate drivers that deliver optically isolated gate control signals that are synchronised to the position of the rotor. The control signal is connected to the MOSFET gate through an optically isolated opto-coupler, for example HCPL-3120 from Hewlett Packard. The opto-coupler is powered from a 15 volt supply derived from an electrically isolated DC to DC converter. One suitable converter is the NME1215S from C&D Technologies which provides 2kV isolation and supplies 1 watt at 15 volts output for a 12 volt input.

The 12 volt supply for the electrically isolated converter can be derived from the AC mains supply by an AC to DC converter. TRECO supplies one suitable converter that converts 50 Hz 240 volts AC to the 12 volts DC for supply to the electrically isolated converter. The controlled transistor switches are coupled to the controller by any suitable means. In the embodiments described below, MOSFET switches are coupled to a switch controller,by opto- couplers, for example HCPL-3120 from Hewlett Packard, with gate drivers powered by isolated converter supplies, for example from C & D Technologies.

Diodes

Some of the switching devices of the embodiments described are semi-conductor diodes.

RURP1560 (600v 30Amp) and RHRG75120 (1200 volt, 75 Amp), or similar hyper-fast diodes, are suitable. The semiconductor diodes require a small forward bias voltage to make the diodes conductive. This requirement has generally been ignored in the following description to simplify the explanation of circuit operation.

Capacitors

The recovery capacitors described in the following embodiments are preferably "low-loss" pulse rated capacitors, i.e. capacitors having low equivalent series resistance and low equivalent series inductance. Suitable recovery capacitors are metallised polypropylene pulse rated capacitors, or metallised polypropylene foil-film pulse rated capacitors for applications generating high voltages on the recovery capacitors. The circuit of each embodiment described below includes a capacitor that temporarily stores energy recovered from the collapsing magnetic field of a motor winding. These recovered energy storage capacitors are, for convenience, generally referred to in this specification by the briefer term "recovery capacitor", to help distinguish the function of these capacitors from those of other capacitors, for example filter capacitors or power supply DC link reservoir capacitors.

Motors

The motors used in each of the three example embodiments described below are two phase 2/ 4 switched reluctance motors, i.e. each 2/4 motor has a two pole rotor driven by a four pole stator. The stator carries four windings comprising two windings for alternately magnetising each respective phase pair of the stator poles. The inductance of each winding varies according to the rotational position of the rotor with respect to the stator. The inductance of each winding is at a minimum when the rotor poles are approximately midway between the poles of the respective winding, and at a maximum when the rotor poles are in alignment with the poles of the respective winding.

Windings

It should also be noted that, for simplicity, the motor windings are shown in the accompanying circuit diagrams as idealised inductances. In practice, these windings also comprise winding resistance, core losses, and in some instances small inter-winding capacitance, which have not been shown in accompanying figures nor described further in the accompanying description, to simplify the explanation of circuit operation. The winding parameters may be optimised to maximise the power transfer into and out of the winding inductance in line with the principles of the Maximum Power Transfer theorem.

Earthing/ Grounding

The circuits shown in the figures have a bottom rail that is earthed or grounded. The earthing or grounding of this rail is optional and, although useful in many practical motor applications, is not a necessary part of the invention.

EMBODIMENTS SHOWN IN THE FIGURES Each of the circuit diagrams of Figures 1 A, 2A and 3A show two phase windings of a switched reluctance motor powered by a two-phase switching circuit from a 50 Hz power supply VI, via a filter inductor LI, bridge rectifier BR1 and a supply DC link reservoir capacitor CI. Switched reluctance motor examples all run at a frequency much higher than the 50Hz AC supply such that the supply input current is modulated with a high frequency input ripple. High power factor is achieved by suitable choice of the input filter inductance LI and capacitance CI.

Common features shared by the three circuits of Figures 1 A, 2A and 3A are described below. In each circuit, one phase winding L2A is driven by a respective A phase circuit comprising MOSFETs Q1A and Q2A, semiconductor diodes suffixed A, and a recovery capacitor C2A, connected as shown in the figures, to put the A phase circuit into the successive first, second and third configurations. The other phase winding L2B is driven by a respective B phase circuit comprising MOSFETs QIB and Q2B, semiconductor diodes suffixed B, and recovery capacitor C2B, connected as shown in the figures, to put the B phase circuit into the successive first, second and third configurations. The two windings L2A and L2B are driven in quadrature, i.e. synchronised to respective rotor positions that are 90 degrees apart, to reflect the orthogonal arrangement of the two pairs of stator poles. The MOSFETs are driven by a switching controller through opto-coupled gate drivers. A rotor position signal derived from a rotor position sensor is coupled to the switching controller to synchronise the switching of the MOSFETs with the rotational position of the motor rotor with respect to the motor stator. The configuration of the switching circuits in the successive first, second and third configurations are repeated twice for each revolution of the rotor to power the windings and thereby operate the motor.

The circuits use energy recovered from the collapsing magnetic field at each stator winding to more effectively drive the motor. The delivery of energy from the supply is throtded by the inductive/ capacitive impedance circuits comprising inductors LI and L3, and capacitors CI and C4. As the delivery of energy through these impedance circuits reduces and the back EMF at the winding increases, a high voltage on the recovery capacitor C2 is used to augment or maintain the magnetic field at the winding. The amplitude and/ or duration of the motors magnetic field may be increased. In some cases, the voltage of the recovery capacitor C2 is much higher than that across the supply reservoir capacitor CI, making the augmentation of the magnetic field possible even when the back EMF at the winding has increased.

First configuration

The MOSFET Q2 is turned on (i.e. made conductive) when the rotor poles are not in alignment with the stator poles to be magnetised by a respective motor winding L2 and the inductance of winding L2 is at a minimum. Turning MOSFET Q2 on puts the switching circuit that powers the motor winding L2 into a first configuration. In this first, or magnetising, configuration, current begins to flows from the supply reservoir capacitor CI, through the winding L2, and through the MOSFET Q2. This current establishes a magnetic field at the winding L2. Second configuration

The MOSFET Ql is turned on at the end of a delay period following the turning on of MOSFET Q2. MOSFET Q2 remains on. At this time the rotor poles have moved toward, but have not reached, full alignment with the stator poles being magnetised by motor winding L2, and the winding inductance has increased. The turning on of MOSFET Ql puts the circuit that drives the motor winding L2 into a second configuration.

In the second configuration, current begins to flows from recovery capacitor C2 (charged during configuration of the circuit in the third configuration of the preceding cycle), through MOSFET Ql, the winding L2 and the MOSFET Q2. This current maintains or augments the magnetic field at the winding L2, already established during the first configuration.

In the second configuration, the circuit recycles energy, previously recovered from the magnetic field at the winding L2 and stored on recovery capacitor C2, by injecting the energy back into the winding thereby maintaining or augmenting the magnetic field established by the first configuration.

The switching controller, preferably programmed using a look-up table, synchronises the turn-on of MOSFET Ql with the turn-on of MOSFET Q2 when the motor speed is below a transition speed, and progressively and linearly delays the turn-on of MOSFET Ql after turn-on of Q2 for speeds above a transition speed. The delay is dependent on motor speed above the transition speed but is typically half the rated full speed of the motor.

Third configuration

The MOSFETs Ql and Q2 are turned off (i.e. made non-conductive) when the rotor poles have moved further toward full alignment with the stator poles being magnetised by the respective motor winding L2, and the winding inductance has increased further but has still not reached its maximum. The turning off of MOSFETs Ql and Q2 puts the circuit that drives the motor winding L2 into a third configuration. The circuit is automatically put into the third

configuration by semiconductor diodes D3 and D4 which turn on to provide a conductive path for current induced in the winding on collapse of the magnetic field when the MOSFETs Ql and Q2 are turned off simultaneously. In the third, or recovery, configuration, the collapsing magnetic field at winding L2 induces a current to flow from the winding L2, through diode D3 and into recovery capacitor C2, and back to the winding L2 through diode D4. In this configuration, the circuit recovers and stores energy from the magnetic field. The maximum recovered energy and recovery current flow is achieved in the third configuration when the capacitor is depleted to as low a voltage as possible after the end of the second configuration. Figure 2A shows the use of an appropriately sized auxiliary 'boost' capacitor C3 linked to ground below the recovery capacitor C2 which assists, in a similar way to a charge pump capacitor, in achieving the maximum voltage differential on the recovery capacitor and a boost in recovery current delivery by discharging in series with C2 during the second configuration period.

The recovery circuit component values are chosen, taking into account the laws of maximum power transfer between a capacitor and inductor or through empirical testing if necessary, to optimise the level of recovered energy and consequently recovery capacitor voltage so that the winding current drops to zero before the circuit is next configured in the first configuration.

FIGURE 1A EMBODIMENT

Figure 1A is a circuit diagram illustrating a first embodiment of the invention driving a switched reluctance motor from a Shop-Vac Corporation industrial vacuum cleaner. The motor is powered by a two-phase switching circuit from a 120 volt 50 Hz AC power supply VI, via a filter inductor LI, bridge rectifier BR1 and a supply reservoir capacitor CI.

Each motor winding L2 is driven by a respective phase of the switching circuit, comprising a pair of MOSFETs Ql and Q2, semiconductor diodes, and a recovery capacitor C2, connected as shown in the figures. The MOSFETs are driven by a switching controller and opto-coupled gate drivers. A rotor position signal derived from a rotor position sensor is coupled to the switching controller to synchronise the repetitive switching of the MOSFETs with the rotational position of the motor rotor with respect to the motor stator.

First configuration

In the first configuration, further to the above description referring to the common features of Figures 1A, 2A and 3A, current begins to flow from the supply reservoir capacitor CI, via diode Dl and through the winding L2, through the MOSFET Q2, and back to the reservoir capacitor CI at the earth or ground rail. This current establishes a magnetic field at the winding L2.

Figure IB shows the typical current waveform of one phase of the Dyson motor during the peak of the 50Hz AC cycle. Figure 1C shows one phase of the voltage cycle on the recovery capacitor C2 relative to ground. Figure ID shows the switch timings for Ql and Q2 for the first, second and third configurations starting at t _l5 1,, t ₃ respectively. Figures IB, 1C, ID are vertically time aligned to show the synchronisation of the switch timing sequence with the motor winding current and recovery capacitor voltage for one phase over two cycles of the various

configurations .

Second configuration

In the second configuration, further to the above description referring to the common features Figures 1A, 2A and 3A, current begins to flow from the charged recovery capacitor C2, via MOSFET Ql and through the winding L2, through the MOSFET Q2 and diode D2 back to the recovery capacitor C2.

Third configuration

The third configuration is as described above with reference to the common features of Figures 1A, 2A and 3A.

Specific embodiment

One specific embodiment of the circuit shown in Figure 1A has the following component types and circuit values:

MOSFETs Q1A, Q1B, Q2A and Q2B: IRG4PSH71K

Diodes D1A, DIB, D2A, D2B, D3A, D3B, D4A and D4B: RHRG75120

Supply voltage: 120 volts 50 Hz AC

CI: 18.8 μΡ

Ll: 1.5 mH

Motor winding inductance L2A and L2B:

varies between minimum of 1.5 mH and maximum of 7.2 mH

MOSFET switching frequency: 666 Hz at motor speed of 20,000 rpm Duration of first configuration: 177 μ5 at motor speed of 20,000 rpm Duration of second configuration: 442 μ8 at motor speed of 20,000 rpm.

The total period, in each cycle, during which current is supplied to each phase winding of the motor is the total duration of the first and second configurations, i.e. the delay before commencement of the third configuration. At a rotor speed of 20,000 rpm, this period is 619μ8 which is about 41% of the 1.5 mS repetition period of the MOSFET switching.

The duration of the first configuration, i.e. the delay before commencement of the second configuration, is 177 μ8 which represents about 12% of the 1.5 mS repetition period of the

MOSFET switching, at a rotor speed of 20,000 rpm. This delay before commencement of the second configuration is about 28.6% of the total period, in each cycle, during which current is supplied to each phase of the motor winding. In this specific embodiment the motor winding current has a peak value of about 27 Amperes at the peak of the AC supply cycle.

The following Table 1 shows a comparison of a standard Shop-Vac 2/ 4 switched reluctance motor, and the same motor driven by the circuit shown in Figure la with the specific values and parameters set out above. Test data was obtained using an ASTM specified plenum chamber with a 23mm diameter sharp edge orifice and results recorded with an Agilent DSO7034 digital sampling oscilloscope, a Voltech PM1000+, high precision, high bandwidth wattmeter, a precision digital tachometer and digital vacuum gauge.

Table 1: Comparison of performances of Shop-Vac motor

Table 1 shows the circuit described above decreased supply current by over 10%, reduced power consumption of 3 %, and increased the power factor from 0.90 to 0.98 without substantially dropping the speed and vacuum performance, when compared to the Shop-Vac motor driven in an unmodified mode. The power saving figure of 3% would improve to nearer a 10% saving if the power factor of the standard ShopVac motor were corrected to meet the power factor of the converted motor.

FIGURE 2A EMBODIMENT

Figure 2A is a circuit diagram illustrating a second embodiment of the invention driving a switched reluctance motor of a type used in a Dyson Airblade hand dryer. The motor is powered by a two-phase switching circuit from a 120 volt 50 Hz AC power supply VI, via a filter inductor LI, bridge rectifier BR1 and a supply reservoir capacitor CI.

Each motor winding L2 is driven by a respective phase of the switching circuit, comprising a pair of MOSFETs Ql and Q2, three semiconductor diodes Dl, D3 and D4, an auxiliary boost capacitor C3, and a recovery capacitor C2, connected as shown in Figure 2A. The MOSFETs are driven by opto-coupled gate drivers which are responsive to a rotor position sensor to synchronise the repetitive switching of the MOSFETs with the rotational position of the motor rotor with respect to the motor stator. Capacitor C3 provides further improvements to energy- recovery and power savings.

Figure 2B shows the typical waveform current of one phase of the Dyson motor during the peak of the 50Hz AC cycle. Figure 2C (upper trace) shows in the voltage cycles on the recovery capacitor C2 of one phase relative to ground. Figure 2C (lower trace) shows the voltage cycles on the auxiliary boost capacitor C3 relative to ground. Figure 2D shows the switch timings for Ql and Q2 for the first, second and third configurations starting at tl, t2, t3 respectively. Figures 2B, 2C, 2C are drawn vertically aligned in correct time register to show the synchronisation of the switch timing with the motor winding current and recovery capacitor voltage for one phase over two cycles of the various configurations.

First configuration

In the first configuration, further to the above description referring to the common features of Figures 1A, 2A and 3A, current begins to flow from the supply reservoir capacitor CI, via diode Dl and through the winding L2, through the MOSFET Q2, and back to the reservoir capacitor CI at the earth or ground rail. This current establishes a magnetic field at the winding L2.

Second configuration

In the second configuration, further to the above description referring to the common features of Figures 1A, 2A and 3A, current begins to flow from the series combination of charged capacitors, being recovery capacitor C2 and auxiliary boost capacitor C3, via MOSFET Ql and through the winding L2, through the MOSFET Q2 to ground or earth. This current maintains or augments the magnetic field at the winding L2, already established during the first

configuration. In this configuration, the circuit recycles energy previously recovered from the magnetic field at the winding L2.and stored as a charge on the recovery capacitors C2 and C3.

Third configuration

In the third configuration, further to the above description referring to the common features of Figures 1A, 2A and 3A, the turning off of MOSFETs Ql and Q2 puts the circuit that drives the motor winding L2 into a third configuration. The circuit is automatically put into the third configuration by semiconductor diodes Dl and D3 which turn on to provide a conductive path for current induced in the winding on collapse of the magnetic field when the MOSFETs Ql and Q2 are turned off. In this third, or recovery, configuration, the collapsing magnetic field at winding L2 induces a current to flow from the winding L2, through diode D3, recovery capacitor C2 and auxiliary boost capacitor C3 to ground and back through the supply at supply reservoir capacitor CI, and diode Dl back to the winding L2. This energy recovery current does not flow through diode D4. Diode D4 snubs current spikes.

In the third configuration, the circuit recovers and stores energy from the magnetic field by simultaneously charging the recovery capacitor C2 and the auxiliary boost capacitor C3. After initial run-up of the motor, from rest or a low speed, the voltage across auxiliary boost capacitor C3 swings above and below ground or earth potential. This has the effect of boosting the recovery cap to gain a greater voltage and therefore more recovered energy. This is similar in action to that of a charge pump circuit. The discharge voltage (lowest point on trace) of the auxiliary boost capacitor C3 cycles between roughly Ov at peak of AC cycle to -lOOv during the lowest point in the AC cycle. Figure 2C shows only the waveform at the peak of the AC voltage cycle. The auxiliary boost capacitor C3 maintains a bigger differential across capacitor C2. During the second configuration, capacitors C2 and C3 discharge and will drive a higher winding current because of the higher voltage potential.

Run-up

During initial run-up of the motor, from rest or a low speed, a small current flows from the supply reservoir capacitor CI, through the series circuit of diode Dl, winding L2, diode D3, recovery capacitor C2, and auxiliary boost capacitor C3. This current flows between

configuration of the circuit in the third configuration and the next following first configuration, during run-up before the voltage across the combination of the recovery capacitor C2 and boost capacitor C3 has built up to exceed the voltage across the supply reservoir capacitor CI. Specific embodiment

One specific embodiment of the circuit shown in Figure 2 has the following component types and circuit values:

MOSFETs Q1A, Q1B, Q2A and Q2B: IRG4PSH71K Diodes D1A, DIB, D3A, D3B, D4A and D4B: RHRG75120

Supply voltage: 120 volts AC

CI: 4.7 μΡ

C2A and C2B: 12.7 μΡ

C3A and C3B: 8.0 μΡ

Ll: 3 mH

Motor winding inductance L2A and L2B:

varies between minimum of 0.38 mH and maximum of 2.6 mH

MOSFET switching frequency: 2.701 Hz at motor speed of 81,030 rpm

Duration of first configuration: 63^S at motor speed of 81,030 rpm

Duration of second configuration: 69 μ≤ at motor speed of 81,030 rpm

The total period, in each cycle, during which current is supplied to each phase winding of the motor is the total duration of the first and second configurations, i.e. the delay before

commencement of the third configuration. At a rotor speed of 81,030 rpm, this period is 132 μ≤ which is about 36% of the 370 μ8 repetition period of the MOSFET switching.

The duration of the first configuration, i.e. the delay before commencement of the second configuration, is 63 ≤ which represents about 17% of the 370 μ≤ repetition period of the MOSFET switching, at a rotor speed of 81,030 rpm. This delay before commencement of the second configuration is about 48% of the total period, in each cycle, during which current is supplied to each phase of the motor winding.

In this specific embodiment the motor winding current has a peak value of about 34 Amperes at · the peak of the AC supply cycle.

Table 2 shows a comparison of a standard Dyson 120 volt 2/4 switched reluctance motor, of a type used in a Dyson Airblade hand dryer, and the same motor driven by the circuit shown in Figure 2 with the values and parameters set out above. Test results were obtained with an Agilent DSO7034 digital sampling oscilloscope, a Voltech PM1000+ high precision, high bandwidth wattmeter and precision digital tachometer.

Table 2: Comparison of performances of Dyson Airblade motor Dyson Airblade motor

Standard Dyson Airblade

driven by circuit

motor

described above.

Supply voltage at 50 Hz 120.0 V rms 120.5 volts

Supply current 12.15 A rms 11.45 A rms

Power 1444 W 1378 W

Power factor 0.99 0.99

Loaded speed 81,150 rpm 81,450 rpm

From Table 2 it can be calculated that the circuit shown in Figure 2A provided a decrease in supply current of over 5%, and a decrease in power consumption of over 4.5%, while increasing motor speed, when compared to the Dyson Airblade motor driven in an unmodified mode.

FIGURE 3A EMBODIMENT

Figure 3A is a circuit diagram illustrating a third embodiment of the invention driving an Ametek Lamb Electric switched reluctance motor from an industrial motor-blower, designed by Nidec SRDrives UK. The motor is powered by a two-phase switching circuit from a 230 volt AC power supply VI, via filter inductors LI and L3, filter capacitor C4, bridge rectifier BR1, supply reservoir capacitor CI, and auxiliary reservoir network comprising a series connection of capacitors C5 and C6 and diode D5.

Each motor winding L2 is driven by a respective phase of the switching circuit, comprising a pair of MOSFETs Ql and Q2, three semiconductor diodes D2, D3 and D4, and a recovery capacitor C2, connected as shown in Figure 1A. The MOSFETs are driven by opto-coupled gate drivers which are responsive to a rotor position sensor to synchronise the repetitive switching of the MOSFETs with the rotational position of the motor rotor with respect to the motor stator.

Figure 3B shows a typical waveform current of one phase of the Ametek motor during the peak of the 50Hz AC cycle. Figure 3C shows one phase of a voltage cycle on the recovery capacitor C2 relative to ground. Figure 3D shows the switch timings for Ql and Q2 for the first, second and third configurations starting at t _l5 1 ₂ , t ₃ respectively. Figures 3B, 3C, and 3D are vertically time aligned to show the synchronisation of the switch timing sequence with the motor winding current and recovery capacitor voltage for one phase over two cycles of the various

configurations.

First configuration

In the first configuration of the A phase circuit, further to the above description referring to the common features of Figures 1 A, 2A and 3A, current driving winding L2A begins to flow through the winding L2A and the MOSFET Q2A to establish a magnetic field at the winding L2A.

There are two parts to this current. A first part current flows from the supply reservoir capacitor CI, through the winding L2A and MOSFET Q2A, and back through a first auxiliary reservoir capacitor C6 to the earth or ground rail. This part current discharges reservoir capacitor CI, at least partially, and charges auxiliary reservoir capacitor C6.

A second part current flows from a second auxiliary reservoir capacitor C5, through the winding L2A and MOSFET Q2A, and back through diode D5 to the a second auxiliary reservoir capacitor C5. This second part current discharges the second auxiliary reservoir capacitor C5, at least partially.

In the first configuration of the B phase circuit, further to the above description referring to the common features of Figures 1A, 2A and 3A, current driving winding L2B begins to flow through the winding L2B and the MOSFET Q2B to establish a magnetic field at the winding L2B.

There are two parts to this current. A first part current flows from the supply reservoir capacitor CI, through the second auxiliary reservoir capacitor C5 and winding L2B and MOSFET Q2B to the earth or ground rail. This part current discharges the reservoir capacitor CI, at least partially, and charges the second auxiliary reservoir capacitor C5.

A second part current flows from the first auxiliary reservoir capacitor C6, through diode D5, the winding L2B and MOSFET Q2B, and back to the earth or ground rail. This second part current partially discharges the first auxiliary reservoir capacitor C5. The first auxiliary capacitor C6 is charged and the second auxiliary capacitor C5 is discharged during magnetisation of the A phase winding L2A, and the first auxiliary capacitor C6 is discharged and the second auxiliary capacitor C5 is charged during magnetisation of the B phase winding L2B.

Second configuration

In the second configuration, current begins to flows from the charged recovery capacitor C2, via MOSFET Ql and through the winding L2, through the MOSFET Q2 and diode D2 back to the recovery capacitor C2. This current maintains or augments the magnetic field at the winding L2, already established during the first configuration. In this configuration, the circuit recycles energy previously recovered from the magnetic field at the winding L2.

Third configuration

The third configuration is as described above with reference to the common features of Figures 1A, 2A. However, the Figure 3A embodiment receives a voltage boost from the operation of the third configuration of the opposite phase. The voltage boost is due to the diode linked auxiliary reservoir capacitors C5 and C6 the recovery capacitors C2 of both phases. Figure 3C shows the voltage boost as a voltage step in Figure 3C just prior to the closing of switch Q2. The voltage boost produces an increased voltage on the recovery capacitors and more recovery energy.

Specific embodiment

One specific embodiment of the circuit shown in Figure 3A has the following component types and circuit values:

MOSFETs Q1A, Q1B, Q2A and Q2B: IRG4PSH71K

Diodes D2A, D2B, D3A, D3B, D4A, D4B and D5: RHRG75120

Supply voltage: 230 volts 50 Hz AC

CI: 1.0 Ρ

C2A and C2B: 2.2 μΡ

C4: 25 μΡ

Ll : 2.5 mH

Motor winding inductance L2A and L2B: varies between minimum of 1.86 mH and maximum of 12.2 mH L3: 2.5 mH

MOSFET switching frequency: 732 Hz at motor speed of 21,960 rpm

Duration of first configuration: 256μ8 at motor speed of 21,960 rpm

Duration of second configuration: 331 μδ at motor speed of 21,960 rpm

The total period, in each cycle, during which current is supplied to each phase winding of the motor is the total duration of the first and second configurations, i.e. the delay before commencement of the third configuration. At a rotor speed of 21,960 rpm, this period is 587 μ8 which is about 43% of the 1.366 mS repetition period of the MOSFET switching.

The duration of the first configuration, i.e. the delay before commencement of the second configuration, is 256 μδ which represents about 19% of the 1.366 mS repetition period of the MOSFET switching, at a rotor speed of 21,960 rpm. This delay before commencement of the second configuration is about 43.6% of the total period, in each cycle, during which current is supplied to each phase of the motor winding.

In this specific embodiment the motor winding current has a peak value of about 27 Amperes at the peak of the AC supply cycle.

Tables 3 and 4 show a comparison of a standard Ametek Lamb Electric 2/ 4 switched reluctance motor, and the same motor driven by the circuit described above with the values and parameters set out above. Test data was obtained using an ASTM specified plenum chamber with a 23mm diameter sharp edge orifice and results recorded with an Agilent DSO7034 digital sampling oscilloscope, a Voltech PM1000+, high precision, high bandwidth wattmeter, a precision digital tachometer and digital vacuum gauge.

Table 3: Comparison of performances of Ametek Lamb motor

Ametek Lamb Electric

Standard Ametek Lamb

motor driven by circuit

Electric motor

described above. Supply voltage at 50 Hz 230.5 V rms 230.4 volts

Supply current 6.95 A rms 5.91 A rms

Power 1384 \V 1305 W

Power factor 0.86 0.96

Loaded speed 21,950 - 22,000 rpm 22,000 - 22,040 rpm

Vacuum 123 - 126 cm H ₂ 0 125 - 128 H ₂ 0

ASTM Orifice 23 mm diameter 23 mm diameter

Temperature 22 - 23 °C 22 - 23 °C

From Table 3 it can be calculated that the circuit described above decreased supply current by about 15%, decreased power consumption 5.7% and increased the power factor from 0.86 to 0.96, while marginally increasing the motor speed and vacuum performance. If the power factor of the standard Ametek motor, were corrected to meet the power factor of the converted motor, then the power saving figure from using the invention, would improve from 5.7% to near the 15% current saving value.

Table 4: Second comparison of Ametek motor performances as Tested at UL Labs Auckland , New Zealand on 14 ^th April 2011 .

Ametek Lamb Electric

Standard Ametek Lamb

motor driven by circuit

Electric motor

described above.

Supply voltage at 50 Hz 230.3 V rms 230.9 volts

Supply current 7.11 A rms 5.93 A rms

Power 1410 1370 W

Power factor 0.86 0.96 Loaded speed 21,935 - 2 ,970 rpm 21,750 - 21,890 rpm

Vacuum 128 - 130 cm H ₂ 0 125-127 H ₂ 0

ASTM Orifice 23 mm diameter 23 mm diameter

Temperature/Humidity 21.1 °C @ 48% 22.7 °C @ 48%

From Table 4 it can be calculated that the circuit described above decreased supply current by about 16%, decreased power consumption by 7% and increased the power factor from 0.86 to 0.96 while only marginally decreasing the motor speed and vacuum performance. If the power factor of the standard Ametek motor were corrected to meet the power factor of the converted motor then the power saving figure from using the invention, would improve from 7% to nearer the 16% current saving value.

APPLICATIONS OF THE CURRENT INVENTION

The current invention has been described with reference to driving switched reluctance motors. Other applications include but are not limited to, for example, linear actuators and solenoids. The current invention has application to single and multi-phase devices, including, but not limited to, one phase, two phase and three or more phase devices.

Examples of end uses for switched reluctance motors include traction drives, air compressors, liquid pumps, wheel loaders, bulk handlers, vacuum cleaners, washing machines, machine tools, mining equipment, air conditioning and aerospace applications. Examples of end uses for linear actuators include but are not limited to, solenoids including distributing pumps, linear actuators, linear generators, solenoid valves, and solenoid actuators.

Application of the invention to switched reluctance motors has showed performance improvements by reducing input current by 6% to 15% and improving power factor by up to 11%. The consequent increase in overall efficiency ranged from 2% to 7% however f the power factor of the standard motors, where applicable, were corrected to meet the power factor of the converted motors then the power saving figure from using the invention, would improve to nearer the current saving figure. Some converted motors will naturally provide better power saving than others because of different construcdons and operating speeds.

Motors that are specifically constructed so as to optimise the recovery of field energy can be expected to out-perform the efficiency gains achieved by these conversions of standard switched reluctance commercial motors.