TECHNICAL FIELD

The present disclosure relates generally to the field of transceivers. More particularly, it relates to a full-duplex transceiver for wireless communication.

BACKGROUND

Typically, wireless communication systems applies either time division duplex (TDD) or frequency division duplex (FDD) to avoid cross talk between transmission and reception in a transceiver device.

However, to achieve efficient use of the available communication resources (thereby potentially achieving high effective data-rates and/or low delays) it would be beneficial to enable full-duplex communication, with simultaneous transmission and reception within the same frequency range.

A full-duplex transceiver typically needs to mitigate leakage of the transmitted signal into the received signal. One approach for doing so relates to use of an antenna isolator. Even so, the isolation that is possible to achieve between transmitter and receiver paths of the transceiver is typically limited (e.g., due to variations in antenna impedance).

Therefore, there is a need for alternative approaches to mitigate leakage of the transmitted signal into the received signal in a full-duplex transceiver.

SUMMARY

It should be emphasized that the term "comprises/comprising" (replaceable by "includes/including") when used in this specification is taken to specify the presence of stated features, integers, steps, or components, but does not preclude the presence or addition of one or more other features, integers, steps, components, or groups thereof. As used herein, the singular forms "a", "an" and "the" are intended to include the plural forms as well, unless the context clearly indicates otherwise. Generally, when an arrangement is referred to herein, it is to be understood as a physical product; e.g., an apparatus. The physical product may comprise one or more parts, such as controlling circuitry in the form of one or more controllers, one or more processors, or the like.

Also generally, when full-duplex communication is referred to herein, it is to be understood as encompassing any simultaneous transmission and reception within the same frequency range.

Transmission and reception within the same frequency range is meant to encompass transmission and reception that completely (or partially) overlap in the frequency domain (e.g., transmission and reception having the same carrier frequency, or having the same center frequency).

Furthermore, transmission and reception within the same frequency range is meant to also encompass transmission and reception that do not overlap in the frequency domain (e.g., transmission and reception not having the same carrier frequency, or not having the same center frequency), but that still falls within the same frequency range (e.g., the same frequency band). Thus, transmission and reception within the same frequency range may include transmission and reception in frequency intervals that are adjacent to each other, or otherwise in proximity to each other in the frequency domain.

An example scenario of transmission and reception within the same frequency range relates to communication wherein either or both of the transmission and reception involves frequency hopping (within the frequency range; having a bandwidth BW). In such scenarios, the frequency distance between transmission and reception will vary over time; typically in an interval ranging from the negative of the frequency range bandwidth (-BW) to the positive of the frequency range bandwidth (+BW) and including zero.

It is an object of some embodiments to solve or mitigate, alleviate, or eliminate at least some of the above or other disadvantages.

A first aspect is a full-duplex transceiver arrangement connectable to an antenna. For example, the full-duplex transceiver arrangement may be for connection to the antenna, and/or may be connected to the antenna. The transceiver arrangement comprises a signal modulator arranged in a transmitter path and configured to provide a radio frequency transmission signal to the antenna and mixer circuitry arranged in a receiver path and configured to down-convert a radio frequency reception signal provided from the antenna.

The signal modulator is further configured to provide the radio frequency transmission signal to the mixer circuitry, and the mixer circuitry is further configured to down-convert the radio frequency reception signal by mixing it with the radio frequency transmission signal. Thereby, a part of the radio frequency transmission signal leaked into the radio frequency reception signal is provided as a direct current signal component of the down-converted reception signal.

The transceiver arrangement also comprises control circuitry configured to cause suppression of the direct current signal component of the down-converted reception signal.

In some embodiments, the control circuitry is configured to cause suppression of the direct current signal component by causing passing of the down-converted reception signal through a high-pass filter.

In some embodiments, the control circuitry is configured to cause suppression of the direct current signal component by causing estimation of the direct current signal component based on a digitized down-converted reception signal and removal of the estimated direct current signal component from the digitized down-converted reception signal.

In some embodiments, the estimation of the direct current signal component is further based on one or more of; an amplitude of the radio frequency transmission signal, and a control signal for the amplitude of the radio frequency transmission signal.

In some embodiments, the control circuitry is further configured to cause de-rotation of the down-converted reception signal. Thereby, a phase modulation imposed on the radio frequency reception signal by mixing it with the radio frequency transmission signal may be counteracted.

In some embodiments, a duplex frequency distance between the radio frequency transmission signal and the radio frequency reception signal is variable in an interval comprising a zero frequency distance. In some embodiments, the control circuitry is further configured to cause selection of a bandwidth range of a receiver path filter for the down-converted reception signal based on the duplex frequency distance.

In some embodiments, the control circuitry is further configured to cause selection of a sampling frequency of a receiver path analog-to-digital converter for the down-converted reception signal based on the duplex frequency distance.

In some embodiments, the full-duplex transceiver arrangement further comprises an isolator for connection of the transmitter and receiver paths to the antenna. The isolator may be configured to provide initial mitigation of signal leakage of the radio frequency transmission signal into the radio frequency reception signal, and the control circuitry may be further configured to cause estimation of in-phase and quadrature parts of the direct current signal component and tuning of the isolator based on the estimated in-phase and quadrature parts of direct current signal component.

A second aspect is a full-duplex transceiver comprising the full-duplex transceiver arrangement of the first aspect.

In some embodiments, the full-duplex transceiver is a Bluetooth transceiver.

A third aspect is a wireless communication apparatus comprising the full-duplex transceiver arrangement of the first aspect and/or the full-duplex transceiver of the second aspect.

A fourth aspect is a method of controlling a full-duplex transceiver arrangement connectable to an antenna.

The method comprises providing (by a signal modulator arranged in a transmitter path of the full-duplex transceiver arrangement) a radio frequency transmission signal to the antenna and to mixer circuitry arranged in a receiver path of the full-duplex transceiver arrangement.

The method also comprises down-converting (by the mixer circuitry) a radio frequency reception signal provided from the antenna by mixing it with the radio frequency transmission signal, thereby providing a part of the radio frequency transmission signal leaked into the radio frequency reception signal as a direct current signal component of the down-converted reception signal. The method also comprises suppressing the direct current signal component of the down- converted reception signal.

A fifth aspect is a computer program product comprising a non-transitory computer readable medium, having thereon a computer program comprising program instructions. The computer program is loadable into a data processing unit and configured to cause execution of the method according to the fourth aspect when the computer program is run by the data processing unit.

In some embodiments, any of the above aspects may additionally have features identical with or corresponding to any of the various features as explained above for any of the other aspects.

An advantage of some embodiments is that approaches are provided to mitigate leakage of the transmitted signal into the received signal in a full-duplex transceiver.

Another advantage of some embodiments is that efficient use of the available communication resources may be achieved (potentially leading to high effective data-rates and/or low delays).

BRIEF DESCRIPTION OF THE DRAWINGS

Further objects, features and advantages will appear from the following detailed description of embodiments, with reference being made to the accompanying drawings. The drawings are not necessarily to scale, emphasis instead being placed upon illustrating the example embodiments.

Figure 1A is a schematic block diagram illustrating an example arrangement according to some embodiments;

Figure IB is a schematic block diagram illustrating an example arrangement according to some embodiments;

Figure 2A is a schematic block diagram illustrating an example arrangement according to some embodiments;

Figure 2B is a schematic block diagram illustrating an example arrangement according to some embodiments; Figure 3 is a schematic block diagram illustrating an example arrangement according to some embodiments;

Figure 4 is a flowchart illustrating example method steps according to some embodiments;

Figure 5 is a collection of simulation plots illustrating example signals according to some embodiments;

Figure 6 is a collection of simulation plots illustrating example signals according to some embodiments;

Figure 7 is a schematic diagram illustrating an example isolator arrangement according to some embodiments;

Figure 8 is a plot illustrating parametric sweep of isolation resistance and capacitance according to some embodiments; and

Figure 9 is a schematic drawing illustrating an example computer readable medium according to some embodiments.

DETAILED DESCRIPTION

As already mentioned above, it should be emphasized that the term "comprises/comprising" (replaceable by "includes/including") when used in this specification is taken to specify the presence of stated features, integers, steps, or components, but does not preclude the presence or addition of one or more other features, integers, steps, components, or groups thereof. As used herein, the singular forms "a", "an" and "the" are intended to include the plural forms as well, unless the context clearly indicates otherwise.

Embodiments of the present disclosure will be described and exemplified more fully hereinafter with reference to the accompanying drawings. The solutions disclosed herein can, however, be realized in many different forms and should not be construed as being limited to the embodiments set forth herein.

In the following, embodiments will be described for simultaneous transmission and reception within the same frequency range, where approaches are provided to mitigate leakage of the transmitted signal into the received signal in a full-duplex transceiver. Generally, it should be noted that signals described herein may (according to some embodiments) comprise in-phase and quadrature (I/O.) components; even if that is not explicitly mentioned herein.

Figure 1A schematically illustrates an example full-duplex transceiver arrangement 100 according to some embodiments. The arrangement is connectable to an antenna 135, e.g., via an antenna port 130.

The full-duplex transceiver arrangement 100 is for a full-duplex transceiver (e.g., a Bluetooth transceiver). Thus, the full-duplex transceiver arrangement 100 may be comprisable, or comprised, in a full-duplex transceiver.

The full-duplex transceiver and/or the full-duplex transceiver arrangement 100 may be for a wireless communication apparatus (e.g., a network node - NWN, an access point - AP, a user equipment - UE, a station - ST A, etc.). Thus, the full-duplex transceiver and/or the full-duplex transceiver arrangement 100 may be comprisable, or comprised, in a wireless communication apparatus.

The full-duplex transceiver arrangement 100 comprises a signal modulator (MOD; e.g., signal modulation circuitry or a signal modulation module) 115 arranged in a transmitter path of the transceiver arrangement, a mixer (MIX; e.g., mixer circuitry or a mixer module) 145 arranged in a receiver path of the transceiver arrangement, and a controller (CNTR; e.g., control circuitry or a control module) 165.

In various embodiments, the full-duplex transceiver arrangement 100 may also comprise one or more of an isolator (IS) 125 for connection of the transmitter and receiver paths to the antenna, one or more filters (FILT) 150, an analog-to-digital converter (ADC) 155, a digital part of the transmitter path (D-TX) 110 and a digital part of the receiver path (D-RX) 160.

The controller 165 may be separate from the digital part of the receiver path 160 as illustrated in Figure 1A, or may be (fully or partly) comprised in the digital part of the receiver path 160 and/or may be (fully or partly) comprised in the digital part of the transmitter path 110.

The signal modulator 115 is configured to provide a radio frequency (RF) transmission signal to the antenna and to the mixer 145. The signal modulator may apply any suitable approach to provide the RF transmission signal. The signal modulator may, for example, provide the RF transmission signal by modulating a digital signal sequence provided from the digital part 110 of the transmitter path.

In some embodiments, the signal modulator is implemented using a phase-locked loop (PLL) arrangement and/or a frequency synthesizer arrangement according to any known or future approach. For example the modulator 115 be implemented as a PLL-based frequency synthesizer supporting modulation. Examples include an analog PLL with two-point modulation, an all-digital variant such as an all-digital PLL (ADPLL), and a constant frequency PLL combined with a modulator (e.g., a mixer).

The RF transmission signal may be a constant amplitude (or constant envelope) signal, or may have varying amplitude (or envelope). For example, a signal from a frequency synthesizer may be used to generate a constant envelope signal RF transmission signal, and a polar architecture may be used to impose an amplitude modulation in addition to the frequency (phase) modulation.

It should be noted that the terms envelope and amplitude will be used interchangeably herein, and that features and advantages described in connection to constant/non-constant envelope may be equally applicable in relation to constant/non-constant amplitude, and vice versa.

The mixer 145 is configured to down-convert a radio frequency (RF) reception signal provided from the antenna by mixing it with the RF transmission signal. Since the RF reception signal is mixed with the RF transmission signal, any part of the RF transmission signal that has leaked into the RF reception signal will be provided as a direct current (DC) signal component after down-conversion (since it is mixed with a scaled copy of itself), which may facilitate mitigation of the RF transmission signal leakage.

Thus, leakage of the transmission signal into the reception signal is handled by mixing the leaked signal to a DC signal in the receiver path. This approach may make it easier to remove the leaked signal part as will be exemplified in the following. Furthermore, this approach may minimize (or at least decrease) the effects of distortion and/or phase noise on receiver performance. The controller 165 is configured to cause suppression of the DC signal component of the down-converted reception signal, thereby mitigating signal leakage of the radio frequency transmission signal into the radio frequency reception signal.

Suppression of the DC signal component may be achieved by estimation of the DC signal component based on a digitized down-converted reception signal and cancellation/removal of the estimated DC signal component from the digitized down-converted reception signal. The estimation and cancellation/removal may, for example, be implemented by the digital part 160 of the receiver path.

The estimation of the DC signal component may be in accordance with any suitable approach. For example, the DC signal component may be estimated as an average of the digitized down- converted reception signal.

When the leaked transmission signal has constant envelope it will typically appear as a pure DC signal component at the output of the mixer. When the leaked transmission signal has an amplitude modulation (i.e., has varying envelope) it will typically appear, at the output of an in-phase/quadrature (I/O.) mixer, as an in-phase/quadrature vector with constant phase and a magnitude that follows the amplitude modulation.

When the RF transmission signal also comprises amplitude variations (e.g., when the modulation conveys information in the amplitude of the RF signal), the estimation of the DC signal component may be further based on the amplitude of the RF transmission signal and/or on a control signal for the amplitude of the radio frequency transmission signal (e.g., in the form of the digital signal sequence provided from the digital part of the transmitter path).

When the transmission signal is not constant envelope, the magnitude of the down-converted reception signal (e.g., the length of an in-phase/quadrature vector) will typically not be constant, but will typically vary proportionally to the varying envelope of the transmission signal. However, the phase of the down-converted reception signal will typically be constant also in this scenario. For non-constant envelope transmission signals, the phase of the down- converted reception signal (and its average) can be estimated based on the DC offset(s) at the mixer output, and when the modulation information is known, the transmission signal leakage can be subtracted from the down-converted reception signal in the digital domain. For such scenarios, it may be pre-estimated (via measurements on known digital transmission signal sequences) how much leakage is caused by each relevant amplitude. It may also be pre estimated what the DC signal component is without any transmission signal at all. During operation, the information regarding pre-estimated leakage and DC signal component may be used together with the digital transmission signal sequences provided from the digital part 110 of the transmitter path to estimate the momentary amplitude of the DC signal component.

The cancellation/removal of the estimated DC signal component may also be in accordance with any suitable approach. For example, the cancellation/removal may comprise subtracting the estimated DC signal component from the down-converted reception signal.

Alternatively or additionally, suppression of the DC signal component may be achieved by filtering. For example, a high-pass filter 150 may be provided after the mixer 145. The high- pass filter should preferably cancel out the DC signal component (or suppress it as much as possible) while passing the desired part of the down-converted reception signal as unaffected as possible. In some embodiments, a high-pass DC notch filter is applied. Although illustrated as performed in the analog domain in Figure 1A, the suppression filtering may, additionally or alternatively, be performed in the digital domain (e.g., in the digital part 160 of the receiver path).

The duplex frequency distance between the radio frequency transmission signal and the radio frequency reception signal may be fixed (to zero or to a non-zero value), or may be variable (e.g., for the scenario of frequency hopping) in an interval; wherein the interval typically comprises both negative and positive values, as well as zero.

Suppression by estimation and cancellation/removal may be particularly suitable when there is a zero frequency distance between the RF transmission signal and the RF reception signal (a zero duplex distance).

Suppression by high-pass filtering may be particularly suitable when there is a non-zero frequency distance between the RF transmission signal and the RF reception signal (a non-zero duplex distance). In some embodiments, suppression by high-pass filtering may be combined with suppression by estimation and cancellation/removal. Such an approach may be particularly suitable when there is a non-zero frequency distance between the RF transmission signal and the RF reception signal (a non-zero duplex distance).

In some embodiments, the controller 165 may be configured to dynamically select suppression method based on the duplex distance (zero or non-zero), and cause application of the selected suppression method.

Alternatively or additionally, the controller 165 may be configured to cause selection of (e.g., select) a bandwidth range of the receiver path filter 150 based on the duplex frequency distance, and cause application of the selected bandwidth range in the filter.

For example, a low-pass filter may be selected when the duplex distance is zero and a band pass filter (e.g., tuned to a center frequency at duplex distance) may be selected when the duplex distance is non-zero. For non-zero duplex distance, the filter 150 may comprise a band pass filter and a high-pass DC notch filter according to some embodiments. The latter approach may have the benefit of reducing the dynamic range requirements of the ADC at low duplex frequency distances.

Alternatively or additionally, the controller 165 may be configured to cause selection of (e.g., select) a sampling frequency of the ADC 155 based on the duplex frequency distance, and cause application of the selected sampling frequency in the ADC.

For example, a default sampling frequency may be selected when the duplex distance is zero and a sampling frequency which is higher than the default sampling frequency may be selected when the duplex distance is non-zero. According to some embodiments, the sampling frequency is an increasing function of the magnitude (i.e., the absolute value) of the duplex distance.

Application of an adjustable ADC sampling rate may provide a power efficient approach where power consumption may be decreased for low duplex frequency distances. In an implementation with analog de-rotation and DC-removal, the desired received signal may be down-converted to baseband before the ADC, which may reduce the power consumption of the ADC for high duplex frequency distances. Alternatively or additionally, the controller 165 may be configured to cause tuning of (e.g., tune) the isolator 125 based on the DC signal component (e.g., as estimated by the digital part 160 of the receiver path). Typically, the tuning may be based on estimated in-phase and quadrature parts of the DC signal component. It should be noted that the controller 165 typically needs to acquire information regarding the DC signal component to cause isolator tuning. For example, when the DC signal component is removed by high-pass filtering, separate low pass filtering and low speed ADC may be used (e.g., arranged as a parallel receiver path lacking the high-pass filtering) to provide information regarding the DC signal component to the controller. The isolator is typically configured to provide initial mitigation of signal leakage of the radio frequency transmission signal into the radio frequency reception signal. Typically, the tuning aims at achieving initial mitigation that is as good as possible (e.g., that suppresses the leaked transmission signal as much as possible while leaving the desired reception signal unaffected to as large extent as possible). The initial mitigation and/or the tuning of the isolator may be implemented in accordance with any suitable known or future approach. For example, tuning may comprise balancing of paths in the isolator by varying impedance values of one or more isolator components.

Alternatively or additionally, the controller 165 may be configured to cause de-rotation of the down-converted reception signal. The de-rotation may, for example, be implemented by the digital part 160 of the receiver path, or in the analog domain.

The de-rotation is for counteracting a phase modulation (rotation) imposed on the RF reception signal by mixing it with the RF transmission signal.

Information regarding the phase of the RF transmission signal (e.g., in the form of the digital signal sequence provided from the digital part 110 of the transmitter path and a model of the modulator 115, or in the form of an estimated phase of the RF transmission signal provided from the digital part 110 of the transmitter path) may be used by the controller 165 to determine the de-rotation phase. Figure IB schematically illustrates an example full-duplex transceiver arrangement 100' according to some embodiments. The full-duplex transceiver arrangement 100' may be particularly beneficial when the RF transmission signal also comprises amplitude variations.

The full-duplex transceiver arrangement 100' may be seen as a variation of the full-duplex transceiver arrangement 100 of Figure 1A, wherein the modulator 115 of Figure 1A (which may or may not be a constant amplitude modulator as mentioned above) is replaced by two modulators; a constant amplitude modulator (MOD) 115' and an amplitude modulator (AM) 116. Together, the constant amplitude modulator 115' and the amplitude modulator 116 are configured to provide the RF transmission signal to the antenna.

The constant amplitude modulator 115' is configured to provide its output to the mixer 145 for mixing with the radio frequency (RF) reception signal provided from the antenna to down- convert the RF transmission signal. Thereby, an amplitude modulated RF transmission signal may be used while amplitude variations are avoided in the mixing signal.

Figure 2A schematically illustrates an example full-duplex transceiver arrangement 200 according to some embodiments. The arrangement is connectable to an antenna 235, e.g., via an antenna port 230.

The full-duplex transceiver arrangement 200 is for a full-duplex transceiver (e.g., a Bluetooth transceiver). Thus, the full-duplex transceiver arrangement 200 may be comprisable, or comprised, in a full-duplex transceiver.

The full-duplex transceiver and/or the full-duplex transceiver arrangement 200 may be for a wireless communication apparatus (e.g., a network node - NWN, an access point - AP, a user equipment - UE, a station - ST A, etc.). Thus, the full-duplex transceiver and/or the full-duplex transceiver arrangement 200 may be comprisable, or comprised, in a wireless communication apparatus.

The full-duplex transceiver arrangement 200 comprises a signal modulator (MOD; e.g., signal modulation circuitry or a signal modulation module) 215 arranged in a transmitter path of the transceiver arrangement, two mixers (for in-phase and quadrature, respectively; e.g., mixer circuitry or a mixer module) 245, 246 arranged in a receiver path of the transceiver arrangement, and a controller (CNTR; e.g., control circuitry or a control module) 265. The full-duplex transceiver arrangement 200 also comprises an isolator (IS) 225 for connection of the transmitter and receiver paths to the antenna, two filters (for in-phase and quadrature, respectively; FILT) 250, 251, two analog-to-digital converters (for in-phase and quadrature, respectively; ADC) 255, 256, a digital part of the transmitter path (D-TX) 210 and a digital part of the receiver path (D-RX) 260 comprising the controller 265.

The controller 265 may be comprised in the digital part of the receiver path 260 as illustrated in Figure 2A, or may be separate from the digital part of the receiver path 260.

Furthermore, the full-duplex transceiver arrangement 200 also comprises a power amplifier (PA) 220 arranged in the transmitter path and a low noise amplifier (LNA) 240 arranged in the receiver path.

The signal modulator 215 is configured to provide an RF transmission signal to the antenna via the power amplifier 220, and to the mixers 245, 246 (in in-phase and quadrature versions; achieved by a p/2 rotation illustrated as 247 or in any other suitable manner, e.g., using a quadrature frequency divider or a quadrature oscillator to generate the quadrature signal). The signal modulator may apply any suitable approach to provide the RF transmission signal. The signal modulator may, for example, provide the RF transmission signal by modulating a digital signal sequence provided from the digital part 210 of the transmitter path.

The mixers 245, 246 are configured to down-convert an RF reception signal provided from the antenna via the low noise amplifier 240, by mixing it with the RF transmission signal in in- phase and quadrature versions, respectively. Since the RF reception signal is mixed with the RF transmission signal, any part of the RF transmission signal that has leaked into the RF reception signal will be provided as a direct current (DC) signal component after down- conversion, which may facilitate mitigation of the RF transmission signal leakage. There will typically be one DC signal component from each of the two mixers; whereby the in phase/quadrature vector provides both magnitude and phase information about the transmission signal leakage.

The controller 265 is configured to cause suppression of the DC signal component of the down-converted reception signal, thereby mitigating signal leakage of the radio frequency transmission signal into the radio frequency reception signal. Suppression of the DC signal component may be achieved by estimation of the DC signal component based on a digitized down-converted reception signal and cancellation/removal of the estimated DC signal component from the digitized down-converted reception signal. The estimation and cancellation/removal may, for example, be implemented by the digital part 260 of the receiver path.

Alternatively or additionally, suppression of the DC signal component may be achieved by filtering. For example, high-pass filters 250, 251 may be provided after the mixers 245, 246. The high-pass filters should preferably cancel out the DC signal component (or suppress it as much as possible) while passing the desired part of the down-converted reception signal as unaffected as possible. In some embodiments, a high-pass DC notch filter is applied.

In some embodiments, the controller 265 may be configured to dynamically select suppression method based on the duplex distance (zero or non-zero), and cause application of the selected suppression method.

Alternatively or additionally, the controller 265 may be configured to cause selection of (e.g., select) a bandwidth range of the receiver path filters 250, 251 based on the duplex frequency distance, and cause application of the selected bandwidth range in the filter.

For example, low-pass filters may be selected when the duplex distance is zero and band-pass filters may be selected when the duplex distance is non-zero. For non-zero duplex distance, each of the filters 250, 251 may comprise a band-pass filter and a high-pass DC notch filter according to some embodiments.

Alternatively or additionally, the controller 265 may be configured to cause selection of (e.g., select) a sampling frequency of the ADCs 255, 256 based on the duplex frequency distance, and cause application of the selected sampling frequency in the ADCs.

For example, a default sampling frequency may be selected when the duplex distance is zero and a sampling frequency which is higher than the default sampling frequency may be selected when the duplex distance is non-zero. According to some embodiments, the sampling frequency is an increasing function of the magnitude (i.e., the absolute value) of the duplex distance. Alternatively or additionally, the controller 265 may be configured to cause tuning of (e.g., tune) the isolator 225 based on the DC signal component (e.g., as estimated by the digital part 260 of the receiver path). Typically, the tuning may be based on estimated in-phase and quadrature parts of the DC signal component.

Alternatively or additionally, the controller 265 may be configured to cause de-rotation of the down-converted reception signal. The de-rotation may, for example, be implemented by the digital part 260 of the receiver path.

It should be noted that the controller 265 typically needs to acquire information regarding the DC signal component to cause isolator tuning. For example, when the DC signal component is removed by high-pass filtering, separate low pass filtering and low speed ADC may be used (e.g., arranged as a parallel receiver path lacking the high-pass filtering) to provide information regarding the DC signal component to the controller.

Figure 2B schematically illustrates an example full-duplex transceiver arrangement 200' according to some embodiments. The full-duplex transceiver arrangement 200' may be particularly beneficial when the RF transmission signal also comprises amplitude variations.

The full-duplex transceiver arrangement 200' may be seen as a variation of the full-duplex transceiver arrangement 200 of Figure 2A, wherein the modulator 215 of Figure 2A (which may or may not be a constant amplitude modulator as mentioned above) is replaced by two modulators; a constant amplitude modulator (MOD) 215' and an amplitude modulator (AM) 216. Together, the constant amplitude modulator 215' and the amplitude modulator 216 are configured to provide the RF transmission signal to the antenna via the PA 220.

The constant amplitude modulator 215' is configured to provide its output to the mixers, 245, 246 (in in-phase and quadrature versions) for mixing with the radio frequency (RF) reception signal provided from the antenna via the LNA 240, to down-convert the RF transmission signal. Thereby, an amplitude modulated RF transmission signal may be used while amplitude variations are avoided in the mixing signal.

It should be noted that the full-duplex transceiver arrangements 200 and 200' may be seen as implementations of the full-duplex transceiver arrangements 100 and 100', respectively. Thus, features explained in connection with Figures 1A and IB may be equally applicable to Figures 2A and 2B.

Figure 3 schematically illustrates an example arrangement that may be used as digital part of the receiver path (D-RX) 360 (compare with 160, 260 of Figures 1A, IB, 2A, 2B). The arrangement comprises a controller (CNTR) 365 (compare with 165, 265 of Figures 1A, IB, 2A, 2B). The controller 365 is configured to cause suppression of a DC signal component of a down-converted reception signal as explained above.

For suppression of the DC signal component by cancellation/removal of an estimated DC signal component from the digitized down-converted reception signal, the arrangement may comprise a DC canceller (DCC; e.g., DC cancellation circuitry or a DC cancellation module) 361 and an estimator (EST; e.g., estimation circuitry or an estimation module) 367, which may or may not be comprised in the DC canceller. The estimator may be configured to estimate the DC signal component and the DC canceller may be configured to cancel/remove the estimated DC signal component from the digitized down-converted reception signal (which is illustrated in Figure 3 in the form of in-phase and quadrature parts 372, 373).

As mentioned above, the controller 365 may also be configured to dynamically select suppression method based on the duplex distance (zero or non-zero), and cause application of the selected suppression method, and/or to control one or more of isolator tuning, filtering, and ADC sampling rate, which is illustrated by control signals 375, 376, 377 in Figure 3.

For de-rotation of the down-converted reception signal, the arrangement may comprise a multiplier (MULT; e.g., multiplication circuitry or a multiplication module) 362 and a de-rotator (DR; e.g., de-rotation circuitry or a de-rotation module) 366. The de-rotator 366 may be configured to generate a de-rotation signal based on information 371 regarding the phase of the RF transmission signal (e.g., in the form of the digital signal sequence provided from the digital part of the transmitter path), and the multiplier 362 may be configured to multiply the digitized down-converted reception signal with the de-rotation signal. The de-rotation signal should be suitable to counteract the phase modulation (rotation) imposed on the RF reception signal by mixing it with the RF transmission signal. Digital down-conversion (unless the duplex frequency distance is zero), sharp digital filtering, and demodulation may then be performed in the digital domain.

In some embodiments, the arrangement may comprise one or more digital filters (FILT) 363 and/or a mapper (MAP; e.g., mapping circuitry or a mapping module) 364 as illustrated in Figure 3. The mapper may be configured to map the digital (in-phase and quadrature) values to symbols of a digital signal sequence. Digital down-conversion is omitted in Figure 3 for simplicity.

Figure 4 illustrates an example method 400 according to some embodiments. The method 400 is for controlling a full-duplex transceiver arrangement connectable to an antenna. The arrangements illustrated in Figures 1A, IB, 2A, 2B and 3 may, for example, be configured to perform one or more of the steps of the method 400.

As illustrated by step 420, the method comprises providing (by a signal modulator arranged in a transmitter path of the full-duplex transceiver arrangement) a radio frequency transmission signal to the antenna and to mixer circuitry arranged in a receiver path of the full-duplex transceiver arrangement.

As illustrated by step 440, the method also comprises down-converting (by the mixer circuitry) a radio frequency reception signal provided from the antenna by mixing it with the radio frequency transmission signal. As explained above, this provides a part of the radio frequency transmission signal leaked into the radio frequency reception signal as a direct current signal component of the down-converted reception signal.

As illustrated by step 450, the method also comprises suppressing the direct current signal component of the down-converted reception signal. Suppressing the direct current signal component may comprise passing the down-converted reception signal through a high-pass filter, as illustrated by optional sub-step 451. Alternatively or additionally, suppressing the direct current signal component may comprise estimating the direct current signal component based on a digitized down-converted reception signal and removing the estimated direct current signal component from the digitized down-converted reception signal, as illustrated by optional sub-steps 452 and 453. As illustrated by optional step 460, the method may also comprise de-rotating the down- converted reception signal (thereby counteracting a phase modulation imposed on the radio frequency reception signal by mixing it with the radio frequency transmission signal).

As illustrated by optional step 410, the method may also comprise selecting a bandwidth range of a receiver path filter for the down-converted reception signal and/or a sampling frequency of a receiver path analog-to-digital converter for the down-converted reception signal; based on the duplex frequency distance.

As illustrated by optional step 430, the method may also comprise providing (by an isolator) initial mitigation of signal leakage of the radio frequency transmission signal into the radio frequency reception signal.

As illustrated by optional step 470, the method may also comprise tuning of the isolator based on the estimated direct current signal component.

An example of the applicability of some embodiments will now be described in the context of Bluetooth transceivers. Bluetooth devices typically use time division duplex (TDD), whereby a device cannot receive and transmit simultaneously. An aim of wireless communication systems is typically to provide good performance, e.g., in terms of effective data rates and delays. Use of TDD limits the performance compared to if a device could receive and transmit at the same time. However, it is cumbersome to avoid TDD in Bluetooth using prior art approaches. This is because the transmission and reception occur in the same frequency band.

Furthermore, frequency hopping is generally used for Bluetooth; with different hopping patterns for reception and transmission. Thus, the duplex frequency distance will typically vary over time in a pseudo-random pattern (possibly with both positive and negative values); and it may be equal to, or close to, zero at some moments in time. For example, if the 80MHz ISM band at 2.45GHz is used, the duplex frequency distance will typically vary in the interval ranging from -80MHz to 80MHz.

When the duplex frequency distance is varying in a pseudo-random pattern - including duplex frequency distances equal to, or close to, zero - using a duplex filter to separate reception and transmission (as is common for frequency division duplex - FDD) is generally not helpful or even possible. For handling of zero duplex frequency distance, a radio frequency isolation device based on cancellation may be used to connect the receiver and transmitter to the antenna. However, the isolation that can be achieved is limited (especially in view of variations in antenna impedance etc.). Thus, to achieve transmit power level and receiver sensitivity for communication over channels with realistic path loss, mitigation of the remaining part of the transmission signal leaked into the reception signal is desired. Typically, distortion due to the transmission signal also needs to be handled (preferably eliminated from the reception signal) to achieve useful performance. Furthermore, to be beneficial for Bluetooth, a full-duplex transceiver should be able to handle both zero and non-zero duplex frequency distances.

Some embodiments provide a flexible duplex transceiver fulfilling one, more, or all or the above requirements for a Bluetooth transceiver. For example, the transceiver of some embodiments is configured to flexibly adapt to varying duplex frequency distances.

The transceiver architecture according to some embodiments may provide improved performance in terms of delay and/or effective data rate; e.g. compared to TDD systems such as conventional Bluetooth.

Figures 5 and 6 are collections of simulation plots illustrating example signals according to some embodiments.

A setup similar to that of Figure 2A was used, with 1 MHz 4 ^th order Butterworth filters at 250, 251, and a 11-bit l/Q ADC at 255, 256.

The digital part 260 of the receiver path comprises a peak detector DC canceller (clocked at 25 MHz, using a first in first out, FIFO, depth of 500 and a latency of 20 ps, and outputting 6-bit I/O. samples), followed by a digital multiplier (clocked at 25-320 MHz, and multiplying the I/O. cos # —sin # , wherein Q is the transmit phase information provided from 210), sin # cos # an 800 kHz 4 ^th order infinite impulse response (HR) Butterworth filter, and a phase domain mapping with 16 phase regions.

Bluetooth Low Energy (BLE) transmission signals were used, with Gaussian frequency shift keying (GFSK) modulation and a 2MHz channel spacing. The duplex distance was zero. The received signal was at the sensitivity limit (-90dBm). The 1 ^st adjacent channels were 20dB stronger than the received signal (-70dBm), the 2 ^nd adjacent channels were 30dB stronger (- 60dBm), and the 3 ^rd adjacent channels were 40dB stronger (-50dBm). The leakage of the transmission signal into the receiver was at -30dBm; i.e. 60dB stronger than the desired received signal. This setup typically requires at least 11 effective bits in the ADC.

The spectrum of the signal entering the receiver (at the input of the LNA) is shown in part (a) of Figure 5. Part (b) of Figure 5 illustrates the desired received signal entering the receiver. Parts (a) and (b) of Figure 5 illustrate RF spectrum and the peaks values 501 and 502 differ by approximately 60dB.

The signal of part (a) of Figure 5 is mixed with the frequency modulated transmission signal as described above. The signal after mixing is shown in part (c) of Figure 5, wherein the DC signal component can be seen at 504. The signal of part (c) is shown after low-pass filtering in part (d) of Figure 5, wherein the value 505 is approximately 55dB lower than the value 503 and the DC signal component can be seen at 506. Parts (c) and (d) of Figure 5 illustrate baseband spectrum.

The signal of part (d) is then AD-converted and the DC component is detected and cancelled, and the resulting signal spectrum is shown in part (e) of Figure 5. Part (f) shows the signal spectrum after de-rotation. It can be seen (e.g., using the dashed vertical line) that the spectrum is slightly narrowed after de-rotation (removal of the transmit phase modulation). Since the signal bandwidth is no longer widened by transmit modulation, sharp digital filtering can be applied before de-modulation, and part (g) shows the spectrum of the filtered signal. Interference reduction may be seen, e.g., by comparing the value 507 with the value 508, the latter being approximately 30dB lower. Parts (e), (f) and (g) of Figure 5 illustrate baseband spectrum.

After filtering, the signal is de-modulated. Part (a) of Figure 6 shows the de-modulated bit stream in time domain and part (b) shows the transmitted bit stream. By comparison, it can be seen that the signal is correctly demodulated; i.e., the de-modulated bit sequence matches what was transmitted - with some delay.

More time domain waveforms are also shown in Figure 6: part (c) illustrates l/Q.-signals at the output of the analog low-pass filter, part (d)) illustrates l/Q.-signals after ADC and DC cancellation, part (e) illustrates l/Q.-signals after de-rotation, part (f) illustrates l/Q.-signals after baseband digital filtering, and part (g) illustrates the de-modulated signal (compare with part (a))·

The simulations show an example wherein, by using the approach of converting the transmission signal leakage into a DC signal component, adequate signal reception performance is achieved despite strong transmission signal leakage and strong adjacent channels.

Figure 7 is a schematic diagram illustrating an example isolator arrangement according to some embodiments. The isolator of Figure 7 was simulated to investigate the isolation versus tuning parameter characteristics. The inductors were modelled with a finite quality factor and the antenna port had a reactive mismatch. The following parameter values were used: Li=5.2 nH, 1. ₂ =10.4 nH, Ci=0.65 pF, C ₂ =1.3 pF, 100 W TX/RX port impedance, 80+j20 W antenna port impedance, and inductor Q.=10.

The result of the simulation of the isolator of Figure 7 is illustrated in Figure 8 by plotting a parametric sweep of isolation resistance and capacitance, showing real and imaginary parts of transmission signal leakage into the receiver path through the isolator (represented by the y- axis in Figure 7; in the range -0.070 to 0.170).

The isolation was simulated at 2GHz, sweeping the isolation impedance ( Rj _S0 , Gso, Ci _SO c) . In the simulation a single variable, Cc, was used to control Ci and Cj _S oc- For positive Cc, Ci is equal to Cc and Cj _SO c is equal to zero. For negative Cc, Cj _SO c is instead equal to -Cc and Ci is equal to zero. The variable Cc was swept from -400 fF to 400 fF (represented by the x-axis in Figure 7). The isolation resistance Ri was swept parametrically, using 15 values in a +/-20% range, with the centre value of 106 W, which is represented by corresponding curves in Figure 8.

If the excess phase shifts due to signal routing of the transmission signal are equal when arriving at the two ports of the receiver path mixer, the real and imaginary parts of the isolation in this simulation correspond to in-phase and quadrature DC components at the reception path mixer output. If the excess phase shifts are not equal, there will be a rotation of the I/O. DC vector, which can be mitigated by e.g. the tuning scheme described below.

It can be observed in Figure 8 that the real and imaginary parts of isolator transfer (in the range shown) are monotonously dependent on both Ri and Cc, and that both real and imaginary parts can be tuned separately by changing Ri _S0 and Cc. In the example shown, there is a rather steep dependence of the imaginary part on Cc, whereas the real part depends more on Riso, so tuning of the real part can be performed mainly on Ri _S0 and tuning of the imaginary part can be performed mainly on Cc.

Various algorithms may be envisioned for bringing the DC vector towards the origin of the l/Q.- plane. One possible approach involves taking several steps towards the origin (the smaller the steps - the lower risk for mistakes, but the longer the time to reach the origin). Performing a step may comprise determining of direction (e.g., by applying two different change vectors to the current Ri _S0 Cc vector), applying the determined direction, observing the resulting change in the DC vector, and using the observation for determining the direction of the next step.

Generally, when an arrangement is referred to herein, it is to be understood as a physical product; e.g., an apparatus. The physical product may comprise one or more parts, such as controlling circuitry in the form of one or more controllers, one or more processors, or the like.

The described embodiments and their equivalents may be realized in software or hardware or a combination thereof. The embodiments may be performed by general purpose circuitry. Examples of general purpose circuitry include digital signal processors (DSP), central processing units (CPU), co-processor units, field programmable gate arrays (FPGA) and other programmable hardware. Alternatively or additionally, the embodiments may be performed by specialized circuitry, such as application specific integrated circuits (ASIC). The general purpose circuitry and/or the specialized circuitry may, for example, be associated with or comprised in an apparatus such as a wireless communication apparatus.

Embodiments may appear within an electronic apparatus (such as a wireless communication apparatus) comprising arrangements, circuitry, and/or logic according to any of the embodiments described herein. Alternatively or additionally, an electronic apparatus (such as a wireless communication apparatus) may be configured to perform methods according to any of the embodiments described herein.

According to some embodiments, a computer program product comprises a computer readable medium such as, for example a universal serial bus (USB) memory, a plug-in card, an embedded drive or a read only memory (ROM). Figure 9 illustrates an example computer readable medium in the form of a compact disc (CD) ROM 900. The computer readable medium has stored thereon a computer program comprising program instructions. The computer program is loadable into a data processor (PROC; e.g., data processing circuitry or a data processing unit) 920, which may, for example, be comprised in a wireless communication apparatus 910. When loaded into the data processor, the computer program may be stored in a memory (MEM) 930 associated with or comprised in the data processor. According to some embodiments, the computer program may, when loaded into and run by the data processor, cause execution of method steps according to, for example, any of the methods as illustrated in Figure 4 or otherwise described herein.

Generally, all terms used herein are to be interpreted according to their ordinary meaning in the relevant technical field, unless a different meaning is clearly given and/or is implied from the context in which it is used.

Reference has been made herein to various embodiments. However, a person skilled in the art would recognize numerous variations to the described embodiments that would still fall within the scope of the claims.

For example, the method embodiments described herein discloses example methods through steps being performed in a certain order. However, it is recognized that these sequences of events may take place in another order without departing from the scope of the claims. Furthermore, some method steps may be performed in parallel even though they have been described as being performed in sequence. Thus, the steps of any methods disclosed herein do not have to be performed in the exact order disclosed, unless a step is explicitly described as following or preceding another step and/or where it is implicit that a step must follow or precede another step.

In the same manner, it should be noted that in the description of embodiments, the partition of functional blocks into particular units is by no means intended as limiting. Contra rily, these partitions are merely examples. Functional blocks described herein as one unit may be split into two or more units. Furthermore, functional blocks described herein as being implemented as two or more units may be merged into fewer (e.g. a single) unit(s). Any feature of any of the embodiments disclosed herein may be applied to any other embodiment, wherever suitable. Likewise, any advantage of any of the embodiments may apply to any other embodiments, and vice versa.

Hence, it should be understood that the details of the described embodiments are merely examples brought forward for illustrative purposes, and that all variations that fall within the scope of the claims are intended to be embraced therein.