HIGH DATA RATE TRANSMITTER AND RECEIVER

1. Field of the Invention

The invention relates generally to communications, and more particularly to systems and methods for high data rate communications.

2. Background

Wireless communication systems are proliferating at the Wide Area Network (WAN), Local Area Network (LAN), and Personal Area Network (PAN) levels. These wireless communication systems use a variety of techniques to allow simultaneous access to multiple users. The most common of these techniques are Frequency Division Multiple Access (FDMA), which assigns specific frequencies to each user, Time DivisionMultiple Access (IDMA), which assigns particular time slots to each user, and Code Division Multiple Access (CDMA), which assigns specific codes to each user. But these wireless conmiunication systems and various modulation techniques are afflicted by a host of problems that limit the capacity and the quality of service provided to the users. The following paragraphs briefly describe a few of these problems for the purpose of illustration

One problem that can exist in a wireless communication system is multipath interference. Multipath interference, or multipath, occurs because some of the energy in a transmitted wireless signal bounces off of obstacles, such as buildings or mountains, as it travels from source to destination The obstacles in effect create reflections of the transmitted signal and the more obstacles there are, the more reflections they generate. The reflections then travel along their own transmission paths to the destination (or receiver). The reflections will contain the same information as the original signal; however, because of the differing transmission path lengths, the reflected signals will be out of phase with the original signal As a result, they will often combine destructively with the original signal in the receiver. This is referred to as fading. To combat fading, current systems typically try to estimate the multipath effects and then compensate for them in the receiver using an equalizer. In practice, however, it is very difficult to achieve effective multipath compensation.

A second problem that can affect the operation of wireless communication systems is interference from adjacent communication cells within the system In FDMA/TDMA systems, this type of interference is prevent Ihrough a frequency reuse plan. Under a frequency reuse plan, available communication frequencies are allocated to communication cells within the communication system such that the same frequency will not be used in adjacent cells. Essentially, the available frequencies ae split into groups. The number of groups is termed the reuse factor. Then the communication cells are grouped into clusters, each cluster containing the same number of cells as there are frequency groups. Each frequency group is then assigned to a cell in each cluster. Thus, if a frequency reuse factor of 7 is used, for example, then a particular communication frequency will be used only once in every seven communication cells. Thus, in any group of seven communication cells, each cell can only use 1/7 of the available frequencies, i.e., each cell is only able to use 1/7 of the available bandwidth.

In a CDMA communication system, each cell uses the same wideband communication channel, In order to avoid interference with adjacent cells, each communication cell uses a particular set of spread spectrum codes to differentiate

coiTimiuiications within the cell from those originating outside of the cell Thus, CDMA systems preserve ihe bandvvicrth in Hie sense fliat tiiey avoid reuse plaining. But as will be discussed, there are other issues that limit Hie bandwidth in CDMA systems as welL Thus, in overcoming interference, system bandwidth is often sacrificed Bandwidth is becoming a very valuable commodity as wireless communication systems continue to expand by adding more and more users. Therefore, trading off bandwidth for system performance is a costly, albeit necessary, proposition to is inherent in all wireless communication systems.

The foregoing are just two examples of the types of problems that can affect conventional wireless communication systems. The examples also illustrate that toe are many aspects of wireless commuracatmsys(miperfoinτanceύiatcanbe improved through systems aid methods that, for example, reduce interference, increase baidwidth, or both. Not only ae conventional wireless communication systems effected by problems, such as those described in the preceding paragraphs, bin also different types of systems ae effected in different ways and to different degrees. Wireless communication systems cai be split into three types: 1) line-of-sight systems, which can include point-to-point or poiit-to-multipoint systems; 2) indoor non-line of sight systems; and 3) outdoor systems such as wireless WANs. Une-of-sight systems ae least affected by die problems described above, while indoor systems ae more affected, due for example to signals bouncing off ofbuilding walls. Outdoor systems are by far the most affected of the three systems. Because these types of problems are limiting factors in die design of wireless tansmitters and receivers, such designs must be tailored to die specific types of system in which it will operate. In practice, each type of system implements unique communication standards that address the issues unique to die particular type of system. Even if an indoor system used die same communication protocols and modulation techniques as an outdoor system, for example, die receiver designs would still be different because multipath and other problems ae unique to a given type of system and must be addressed with unique solutions. This would not necessarily be die case if cost efficient aid effective mediodologies can be developed to combat such problems as described above diat build in programmability so that a device can be reconfigured for different types of systems and still maintain superior performance.

SUMMARY OF THE INVENTION

Ih order to combat die above problems, a high data rate transmitter and receiver ae provided M one embodiment, a transmitter comprises abasebaid processor structured to receive data and to convert die data into amultipliάty ofhigh and low signal values, with each high and low signal value having a first timing interval A local oscillator generates a clock signal at a second timing interval aid a digital circuit combines the high aid low signal values witii die clock signal to produce atransmission sigial drtectlyatatraismissionfiequαxy.

The radio frequency used for transmission may range up to 11 Giga-Hertz, aid μυduction of the transmission signal directly at die tøansmission frequency is possible by use of a high-speed oscillator.

A receiver is structured to receive die communication signal, which in one embodiment, may have a fractional baidwiddi that may range between approximately 20 percent aid approximately 200 percent The receiver includes a highspeed aialog to digital converter configured to directly convert die radio frequency signal into a data signal These aid other features and advantages of die present invention will be appreciated from review of the following Detailed Description of the

Prefened Embodiments, along with the accompanying figures in which like reference numerals are used to desαibe the same, similar or corresponding pals in Ihe sevaal views of the drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

Preiared αnbodiments of the present inventions taught herein are illustrated by way of example, and not by way oflimitation, in tie figures of the accompanying drawings, in which:

HG. IA is adagrømilliisira&ganemφlea channels in accoidance with the invaition;

FIG. IB is a diagram illustrating the effects of multipath in a wireless communication system;

HG 2 is a diagram illustrating anoiher example embodimait of a wideband communication channel divided into aplurality of aib-channels in accoidance with theinvaition;

FIG.3 isadiagramffiiistratingiteappKcati∞ofai^^^ 1 and2;

FIG. 4A is a diagram illustrating Ihe assignment of sub-channels for a wideband communication channel in accoidance wilh the invaition;

FIG. 4B is a diagram illustrating the assignment of time slots for a wideband communication channel in accoidance with the invention;

FIG. 5 is a diagram illustrating an example embodiment of a wireless communication in accordance with the invaition;

FIG.6 is a diagram illustrating Ihe use of synclironization codes in the wireless communication system of figure 5 in accoidance with the invaώon;

FIG. 7 is a diagram illustrating a corolaior that can be used to correlate synclnOnization codes in Ihe wireless communication systan of figure 5;

FIG.8 is adiagram iUustrating synchronization code conelation in accoidance withthe invention;

FIG.9 is a diagram illustrating the cros&correlation propaties of syrclτronization codes configured in accoidance with the invention;

FIG. 10 is a diagram illustrating anotha" example embodiment of a wireless communication system in accoidance with the invention;

FIG. HA is a diagram illustrating how sub-channels of a wideband communication channel according to the present invention can be grouped in accoidance with die presait invention;

FIG. 1 IB is a diagram illustrating the assignmait of Ihe groups of sub- channels of figure 1 IA in accordance with tile invention;

FIG. lZisadiagiianiillustiBtingfliegOupassigtTinentsoffigure llBinthetimedomain;

FIG. 13 is a flow chart illustrating the assignment of sub-channels based on SIR measurements in the wireless coimniimcationsystan of figure 10 in accordance with the invention;

FIG. 14 is a logical block diagram of an example aiibodiment of transmitter configured in accoidance with the

invention;

HG. 15 is a logical block diagram of an example emtodiment of a modulator configured in accordance with the presentiiwentionfcruseinftietønsitiittEroffigure 14;

FIG. 16 is a diagram illustrating an example embodiment of a rate controller configured in accordance with the inventionfcrυseinttiemodulatDroffigure 15;

KG. 17 is a diagram iUustrating another example embodiment of a rate controller configured in accordance with theinveMαiforiisemthemodulatDroffigure 15;

HG. 18 is a diagram iUusfccating an example anbodiment of a fequeπey encoder configured in accordance with theiiVQlionfbruseinthemodulatoroffigure 15;

HG. 19 is a logical block diagram of an example errjbodiment of a TDMZFDM block configured in accordance withtheinverjiimfeuseinthemodulatDroffigure 15;

HG. 20 is a logical block diagram of another example embodiment of a TDMZFDM block configured in accordanrewilhftiehveMonfouseintemodulatDrof figure 15;

HG.21 isalogicalblcck diagram of an example embαliment of aifequer^^ thekventionforuseinthemodulatDrof figure 15;

HG.22isalogicalblodcdiagramofarecdvffc^

HG.23 isalogicalblockdiagramof an example embodiment ofademcxMatorcx-Mguredin accordance withthe invention for useinihexeceiver of figure 22;

HG.24 is a logical block diagram of an example embodiment of an equalizer configured in accordance with the present invention feruse in the demodulatorof figure 23;

HG.25 is a logical block diagram of an example embodiment of a wireless communication device configured in accordance wiihthe invention;

HG.26isanfflusfrationofdifia^cccnmum(^onmelhods;

HG.27 is aiillustrationoftwo ultra-widebandpulses;

HG.28 is a chart of ultra-wideband emission limits as established by the Federal Communications Cbmrnission 0*1^22,2002;

HG.29 illustrates atransrrώts: consistent with one aiiwdimentof thepresent invention;

HG.30 illustrates atimitig diagram of various signals;

HG.31 ffluEilratesafianeαmsistjentwrftioneembodm

HG.32aillustrates one embodiment ofadigital circuit enployed in the transmitter ofHG.29;

HG.32b illustrates asecorxlmiboclimertf 29;

HG 32cillus(rates athMembodimentof adigMcbxdterrpbyedinthetransrritterofHG.29;

HG.33 illustrates a data stream consistentwith one embodimentof thepresentinvention;

HG.34 illustrates areoavαccmsientwithcceerri^^

HG.35 fflustrafesaschematicofafeporti^^

EIG.36iUustiatesasdiernaticofase∞rκIp^^^ 34;

FIG.37iUiistratesoneaiixx3unatf ofaply-pl:^ 36;

FIG.38 illustrates another embodiment of apoly-phase filter employed in the basebandpracessor ofFIG.36; HG.39 iEusttates another timing diagnii^^

FIG.40 illustrates one embodiment of an equalizer consistent with the present invention; FIG.41 illustrates an exemplary EEC encoder and exemplary FEC decoder,

FIG. 42 illustrates an example EEC encoder configured in accordance with one embodiment of the present invention;

FIG.43 illustrates a EEC encoder configured to generate a code word from input data in accordance with one embodiment;

FIG.44 illustrates the encoder ofHG.42 inmoie detail; FIG.45 illustrates further detail for the encoder ofFIG.42;

FIG. 46 illustrates ai example parity node processor that can be included in a decoder in accordance with one embodiment;

FIG.47 illustiatesonenodeoftiTepaiityiTodepiOcessor ofFIG.45;

FIG.48 illustraiestlieparitynodeprocessor ofFIG.45 in more detail; and

HG49illιιstratesaparitynodeprcressorc^

It will be recognized that some or all of the Figures are schematic representations for purposes of illustration and do not necessarily depict the actual relative sizes or locations of the elements shown. The Figures are provided forthe purpose of illustrating one or more embodiments of the invention with the explicit understanding that they will not be used to limit the scope or the meaning of the claims.

DETAILED DESCRIPTION OF THE PREEEMlEP EMBODIMENTS

1. EittTxluction

Bi the following paragraphs, the present invention will be desαibed in detail by way of example with reference to the attached drawings. While this invention is capable of en±xxlrment in many different forms, there is shown in the drawings and will herein be desαibed in detail specific embodiments, with the understanding that the present disclosure is to be considered as an example of the principles of the invention and not intended to limit the invention to the specific embodiments shown and described That is, throughout this description, the embodiments and examples shown should be considered as exemplars, rather than as limitations on the present invention As used herein, the "present invention" refers to any one of Hie embodiments of Hie invention desαibed herein, and any equivalents. Furthermore, reference to various

featιire(s) of tiie'^^saitinvoition" liTOu^iTout this document does mtmeantoaU claimed aiibodhBontsormdhodsniust indide the referenced features).

In oider to improve wireless communication system pαformance and allow a single device to move from one type of system to another, while still maintaining superior performance, the systems and methods described herein provide various communication methodologies that enhance pαformance of transmitters and receives with regard to various common problems that afflict aich systems and that allow die transmitters and/or receivers to be reconfigure! for optimal performance in a v&iety of systems. Accordingly, die systems and melhods described herein define a channel access protocol that uses a common wideband communication channel for all communication cells. The wideband channel, however, is then divided into a plurality of sub-channels. Different sub-channels are thai assigned to one or more usas within each cell. But the base station, or service access point, within each cell transmits one message that occupies the entire bandwidth of the wideband channel. Each user's communication device receives the entire message, but only decodes those portions of the message that reside in sub-channels assigned to the user. For a point-to-point system, for example, a single user ^* may be assigned all sub-channels and, therefore, lias the full wide baid channel available to them, Si a wireless WAN, on the other hand, the sub-channels maybe divided among aplurality of users.

In the ascriptions of example embodiments that follow, implementation differences, or unique concerns, relating to different types of systems will be pointed out to the extent possible. But it should be understood that Hie systems and melhods described herein are applicable to any type of communication systems. In addition, terms such as communication cell, base station, service access point, etc. are used interchangeably to refer to the common aspects of networks at these different levels. To begin illustrating the advantages of die systems and methods described herein, one can start by loolcmg at the mullipath effects for a sύigle wideband co^ 100 ofbandwidth5 as shown in figure IA

Communication sent ova channel 100 in a tadrtional wireless communication system will comprise digital data bits, or symbols, that are encoded and modulated onto a RF carrier diat is centered at frequency f _c and occupies bandwidth B. Generally, die widdi of die symbols (or the symbol duration) T is defined as IfB. Thus, if die bandwidth B is equal to 100MHz, tiien the symbol duration Tis defined by die following equation: T= IfB = 1/100 megahertz (MHZ) = 10 nanoseconds (ITS). (1) λVhen a receiver receives die communication, demodulates it, and then decodes it it will recreate a stream 104 of datasymbols 106 as illustrated in figure lB.RittheiecdvffwiUalsoreceivemiiltipathveisioiis lOSof thesamedatastream. Because multipadi data streams 108 are delayed in time relative to die data stream 104 by delays dl, d2, d3, and d4, for example^ diey may combine destructively with data stream 104.

A delay spread d _s is defined as die delay from reception of data stream 104 to the reception of the last multipadi data stream 1 OS diat interferes with the reception of data stream 104. Thus, in die example illustrated in figure IB, die delay spread d _s is equal to delay d4. The delay spread 4 will vary for different environments. An environment with a lot of obstacles will ατsatealotofmultipathieflecte type environments, die delay spread 4 can be as long as 20 microseconds. Using the 10ns symbol duration of equation (1),

this translates to 2000 symbols. Thus, with a very large bandwidth, such as 100MHz, multipath interference can cause a significant amount of interference at the symbol level for which adequate compensation is difficult to achieve. This is true even for indoor environments. For indoor LAN type systems, the delay spread d _s is significantly shorter, typically about 1 microsecond. For a 10ns symbol duration, this is equivalent to 100 symbols, which is more manageable but still significant By segmenting the bandwidth B into a plurality of sub-channels 202, as illustrated in figure 2, and generating a distinct data stream for each sub-cbanneL the multipath effect can be reduced to a much more manageable leveL For example^ if fee bandwidth b of each sub-channel 202 is 500KHz, then the symbol duration is 2 miαoseconds. Thus, the delay spread d _s for each sub-channel is equivalent to only 10 symbols (outdoor) or half asymbol (indoor). Thus, bybrealάngup amessage that occupies tine entire bandwidth B into discrete messages, each occupying the bandwidth b of sub-channels 202, a very λvideband signal that suffers fiom relatively minor multipath effects is created

Before discussing further features and advantages of using a widebaid communication channel segmented into a plurality of sub-channels as described, certain aspects of the subchannels will be explained in more detail Referring back to figure 2, the overall bandwidth B is segmented into TVsub-cliannels center at frequencies^ tofi _f -i. Thus, the sub-channel 202 that is immediately to the right of/e is offset ϋxxnfc by b/2, where b is the bandwidth of each sub-channel 202. The next subchannel 202 is oftset by 3b/2, the next by 5b/2, and so on To the left oϊfc, each sub-channel 202 is offset by -b/2, -3b/2, - 5V2, etc. Referably, sub-channels 202 arerrai-overiapping as this allows each sub-chamelfobepitx^ssedindependeiitlyiii the receiver. To accomplish this, a roll-off factor is preferably applied to the signals in each sub-channel in a pirlse-shaping step. The effect of such a pulse-shaping step is illustrated in figure 2 by the non-rectangular shape of the pulses in each subchannel 202. Thus, the bandwidth b of each sub-channel can be represented by an equation such as the following: b = (1+rVT; (2)

Where r = die roll-off factor, and T = the symbol duration ^" Without Hie røll-ofϊfactor, Le., 6 = IfT, the pulse shape would be rectangular in the frequency domain, which corresponds to a (sai x)ά function in the time domain. The time domain sigial for a (sόi x)ά signal 400 is shown in figure 3 in order to illustrate the problems associated with a rectangular pulse shape and the need to use a roU-offfactor. As canbe seen, main lobe 402 comprises almost all of signal 400. But some of the signal also resides in side lobes 404, which stretch out indefinitely in both directions from main lobe 402. Side lobes 404 make processing signal 400 much more difficult, which increases Ihe complexity of the receiver". Applying a roll-off factor /; as in equation (2), causes signal 400 to decay faster, reducing the number of side lobes 404. Thus, increasing the roll- off factor decreases the length of signal 400, i.e., signal 400 becomes shorter in time. But including the roll-off factor also decreases the available bandwidth in each sub-channel 202. Therefore, r must be selected so as to reduce the number of side lobes 404 to a sufficient number, e.g, 15, wMestilhiiaxkήzingthe available bm Thus, the overaUbandwidthiϊforcαrmiiiracationctø^

For efficiency purposes related to transmitter desigi, it is preferable that r is chosen so that A/in aquation (S) is an integer. Choosing r so that M is an integer allows for more efficient transmitters designs using, for example. Inverse Fast FomieiTiBrώfomi(lFFT)tecMques. SinceM= N+ N^), andN:is always aniritege^ that iV(r) is an integer. Generally, it is preferable for r to be between 0.1 and 0.5. Therefore, if iVis 16, for example, then 0.5 could be selected for r so tτatN(r) is an integer. Alternatively, if a value for r is chosen in ttie above exanple so that JV(/;) is not anintegα",i?canbemadesHghflyvλ^ is approximately an integer.

2. F^anipleBiτboo^τieiitofaWirdessConτrτiuiτicationS>^5 tern

WMi the above in mind, figure 5 illustrates an exanple communication system 600 comprising aplurality of cells 602 that each use a common wideband communication channel to communicate with communicatioii devices 604 within each cell 602. The common communication channel is a wideband communication channel as described above. Each communication cell 602 is defined as the coverage area of a base station, or service access point, 606 within the cell. One such base station 606 is shown for illustration in figure 5. For purposes of this specification and the claims to follow, the term base station will be used generically to refer' to a device that μ-ovides wireless access to the wireless communication system for aplurality of commiuiication devices, whether the system is a line of sight, indoor, or outdoor system. Because each cell 602 uses the same communication channel, signals in one cell 602 must be distinguishable from signals in adjacent cells 602. To differentiate signals from one cell 602 to another, adjacent base stations 606 use different synchronization codes according to a code reuse plan. Li figure 6, system 600 uses a synchonization code reuse factor of 4, although die reuse factor can vary depending on the application Preferably, the synclironization code is periodically inserted into a cαimiunication from a base station 606 to a cornmunication device 604 as illustrated in figure 6. After a predetermined number of data packets 702, in this case two, the particular" synchronization code 704 is inserted into the information being transmitted by each base station 606. A synclτronization code is a sequence of data bits known to both the base station 606 and any communication devices 604 with which it is communicating. The synchronization code allows such a communication device 604 to synchronize its liming to that ofbase station 606, which, in turn, allows device 604 to decode the data properly. Thus, in cell 1 (see lightly shaded cells 602 in figure 6), for exanpH synchronizatiαi code 1 (SYNCl) is inserted into data stream 706, which is generated by base station 606 in cell 1, after" every two packets 702; incell2 SYNC2 is inserted after" every two packets 702; in cell 3 SYNC3 is inserted; and in cell 4 SYNC4 is inserted. Use of the synchonization codes is discussed in more detail below.

In figure 4A, an example wideband communication channel 500 for use in communication system 600 is divided into 16 sub-channels 502, centered at frequencies/) tofe A base station 606 at the center of each communication cell 602 transmits a single packet occupying die whole bandwidth B of wideband chanriel500.Suchapacketisillustiϊϊtedbypacket 504 in figure 4B. Packet 504 comprises sub-packets 506 ftiat are encoded with a frequency offset (^responding to one of sub-channels 502. Sub-packets 506 in effect define available time slots in packet 504. Similarly, sub-channels 502 can be

said to define available fiequency bins in communication channel 500. Therefore, the resources available in communication cell 602 ae time slots 506 and fiequency bins 502, which can be assigned to different communication devices 604 within each cell 602. Thus, for example, fiequency bins 502 and time slots 506 can be assigned to 4 different commutication devices 604 within a cell 602 as shown in figure 5. Each communication device 604 receives the entire packet 504, but only processes those fiequency bins 502 and/or timeslots 506 lhat ae assigned to it Preferably, each device 604 is assigned non- adjacent fiequency bins 502, as in figure 4A This way, if interference corrupts the infcuTiation. in a portion of communication channel 500, then the effects are spread across all devices 604 within a cell 602. Hopefully, by spreading out the effects of mterference in this manner the effects areminimized and the entire infόrmationsentto each device 604 can still be recreated from 1he unaffected information received in other fiequency bins. For example, if interference, such as fading, coniipted the information in bins j£r$ then each user 1-4 loses one packet of data But each user potentially receives three unaffected packets from the other bins assigned to them. Hopefully, the unaffected data in the other Ihree bins provides enough infonnation to recreate the entire message for each user. Thus, fiequency diversity can be achieved by assigning non-adj acent bins to each of multiple users.

Ensuring that fee bins assigied to one user are separated by more than the coherence bandwidth ensures frequency diversity. As discussed above, the coherence bandwidth is approximately equal to lά& For outdoor systems, where ώ is typically 1 microsecond, l/d _s = 1/1 microsecond = 1 MegaHαtz (MHz). Thus, the non-adjacent frequency bands assigned to a lisa." ae preferably separated by at least IMHz. It is even more preferable, however, if the coherence bandwidth plus some guard band to ensure sufficient fiequency diversity separate the non-adjacent bins assigned to each user. For exanple, it is preferable in certain implementations to ensure that at least 5 times the coherence bandwidth, or 5MHz in the above exaiple, sepaates the non-adjacent bins. Another way to provide fiequency diversity is to repeat blocks of data in fiequency bins assigned to a paticular user that ae separated by more thai Ihe coherence baidwidth. Bi other words, if 4 sub-channels 202 are assigned to a user, then data block a can be repeated in the first and thud sub-channels 202 aid data block b can be repeated in the second and fourth sub-channels 202, provided the sub-channels ae sufficiently separated in fiequency. Li this case, Ihe system can be said to be using a diversity length factor of 2. The system can similarly be configured to ύiplemαitoύiα diversity lengths, e.g, 3, 4,..., /.

It should be noted that spatial diversity can also be included depending on the embodiment Spatial diversity can comprise transmit spatial diversity, receive spatial diversity, or both In transmit spatial diversity, the transmitter uses apluraliry of separate transmitters and a plurality of separate aitermas to transmit each message. In other woids, each transmitter transmits the same message in parallel The messages ae then received from die trarismitters and combined in the receiver. Because the parallel tiansmissions travel different paths, if one is affected by fading, the oftiers will likely not be affected. Thus, when they ae combined in ftte receiver, the message should be recoverable even if one or more of the other teaisrnission paths experienced severe fading. Receive spatial diversity uses a plurality of sepaate receivers and apluraliry of separate antennas to receive a single message. If an adequate distance separates the aitennas, then the transmission path for the signals received by the antennas will be different Again, this difference in the transmission paths will provide

imperviousness to lading when the signals from Hie receivers ae combined Transmit and receive spatial diversity can also be combined within a system such as system 600 so that two antennas ate used to transmit and two antennas ate used to receive. Thus, each base station 606 transmitter can include two antennas, for transmit spatial diversity, aid each comnimiicalion device 604 receiver can include two antennas, for receive spatial diversity. If only transmit spatial diversity is implemented in system 600, then it can be implemented in base stations 606 or in communication devices 604. Similarly, if only receive spatial diversity is included in system 600, then it can be implemented in base stations 606 or communication devices 604. The number of communication devices 604 assigned frequency bins 502 and/or time slots 506 in each cell 602 is preferably programmable in real time. Ih other words, Hie resource allocation within a communication cell 602 is preferably programmable in the face of varying external conditions, i.e, multipath or adjacent cell interference, and varying requirements, i.e, baidwidth requirements for various users within the celL Thus, if user 1 requires the whole bandwidth to download a large video file, lor example, then the allocation ofbins 502 can be adjust to provide user 1 with more, or even all, ofbins 502. Once user 1 no longer requires such large amounts of bandwidth, the allocation ofbins 502 can be readjusted among all of users 14. It should also be noted that all of the bins assigned to a particular user can be used for bodi die fcαwad and reverse link. Alternatively, some bins 502 can be assigned as die forward link and some can be assigned for use on die reverse link, depending on die implementation. To increase capacity, die entire bandwidfli B is preferably reused in each communication cell 602, with each cell 602 being differentiated by a unique synchronization code (see discussion below). Thus, system 600 provides increased immunity to multipadi and lading as well as increased baidwidth due to the elimhiationofirequeixyimisereqiiirements. 3. Synchronization

Figure 6 illustrates an exaiple onbodiment of a syr-chrαnization code correlator 800 (shown in figure 7). When a device 604 in cell 1 (see figure S), for example, receives an incoming communication fiom die cell 1 base station 606, it compares die incoming data with SYNQ in correlator 800. Essentially, die device scans die incoming data trying to conflate die data with die known synchronization code, in this case SYNQ Once correlator 800 matches die incoming data to SYNQ it generates a correlation peak 804 at die output Multipatii versions of die data will also generate correlation peaks 806, aldiough diese peaks 806 are generally smaller tiian correlation peak 804. The device can dien use die coirelationpeaks to perform, enamel estimation, which allows die device to adjust for die multipath using ai equalizer. Thus, in cell 1, if correlator 800 receives a data stream comprising SYNCl, it will generate correlation peals 804 aid 806. If, on die other hand, the data stream comprises SYNC2, for example, dien no peaks will be generated and die device will essentially ignore die incoming communication.

Even diough a data sfrean diat comprises SYNC2 will not create any conization peaks, it can create noise in correlator 800 diat can prevent detection of correlation peaks 804 aid 806. Several steps can be taken to prevent tiiis from occuuing One way to minimize die noise created in correlator 800 by signals from adjacent cells &X>, is to configure system 600 so diat each base station 606 transmits at die same time. This way, die synefoOnization codes caipreferably be generated in such a manna" tiiat only die synchronization codes 704 of adjacent cell data streams, e.g., streams 708, 710, and 712, as

opposed to packets 702 wilhin those steams, will interfere with detection of the correct synchronization code 704, e.g, SYNCl . The syrchronization codes can then be further configured to eliminate or reduce the interference. For example, the noise or interference caused by an inconect synchronization code is a function of the cross correlation of feat synchronization code with respect to the correct code. The better the cross correlation between the two, the lower the noise leveL When Hie cross conelation is ideal, then the noise level will be virtually zero as illustrated in figure 8 by noise level 902. Therefore, a preferred artbodiment of system 600 uses synchiOnization codes feat exhibit ideal cross correlation, Le., zero. Preferably, the ideal cross conelation of fee synchronization codes covers a period / that is sufficient to allow accurate detection of multipatti 906 as well as multipafe correlation peals 904. This is important so that accurate channel estimation and equalization can take place. Outside of period /, fee noise level 908 goes up, because fee data in packets 702 is random aid will exhibit low cross correlation wife fee synchiOnization code, e.g, SYNCl. Preferably, paiod / is actually slightly longer then fee multipafe length in order to ensure feat fee multipafe can be detected a SvnclTiOnizationcode generation

Conventional systems use orthogonal codes to achieve cross conelation in conelator 800. In system 600 for example, SYNCl, SYNC2, SYNC3, and SYNC4, coneεponding to cells 1-4 (see lightly shaded cells 602 of figure S) respectively, will all need to be generated in such a manner feat they will have ideal cross conelation wife each other. In one embodiment if fee data streams involved comprise high aid low data bits, then fee value ' ¹ I " can be asagnedtothe high data bits aid "-1" to fee low data bits. Cαfeogonal data sequences ae then those that produce a "0" output when they ae exclusively ORed (XORed) together in conelator 800. The following exaτiple illustrates this point for orthogonal sequences I and2:

Thus, when fee lesultsofXORing each bit pair ae added, fee result is "0".

But in system 600, for example, each code must have ideal, or zero, cross correlation wife each of fee ofeer codes used in adjacent cells 602. Thaefore, in one exanple anbodiment of a method for generating synchronization codes exhibiting fee propaties described above, fee process begins by selecting a ' "pafect sequence' ' to be used as fee basis for fee codes. A pafect sequence is one feat when conelated wife itself produces a number equal to fee number of bits in fee sequence. For example:

But each time a perfect sequence is cyclically shified by one bit, fee new sequence is orthogonal wife fee original sequence. Thus, for example, if perfect sequence 1 is cyclically shifted by one bit and then conelated wife fee original, fee correlationproduces a "0" as in fee following exanple;

If Hie perfect sequence 1 is again cyclically shifted by one bit, and again conelated with the original, Ihen it will produce a "0". In general, you can cyclically shift aperfect sequence by any number ofbits up to its length and correlate Ihe shifted sequence with Ihe original to obtain a "0". Once a perfect sequence of the correct lenglh is selected, the first synditonizalion code is preferably generated in one embodiment by repeating the sequence 4 times. Thus, if perfect sequence 1 isbdiigiise^teiafii^s^hOnizidmcodevwouldbethefollowiiig y=l l-l l 11-11 11-11 11-11. Otmgene∞iαm ^f y=:^C^X For asequenceoflengthL: y=x(0)x(l)....X(L)X(0)X(L)...x(L)x(0)x(i)....x(L)x(0)x(l)... x(L)

Repeating Hie perfect sequence allows conelator 800 a bettei' opportunity to detect Ihe synchonization code and allows generation of other uricorrelated frequencies as welL Repeating lias Ihe effect of sampling in Ihe frequency domain. This effect is illustratedby the graphs in figure 9. Thus, in TRACE 1, which con^spoiTdstosiαTchiOnizalioncode^asaiωple 1002 is generated every fourth sample bin 1000. Each sample bin is separated by 1/(4LxT), where T is the symbol duration. Thus, in the above example, where L ~4, each sample bin is separated by 17(16κT) in the frequency domain TRACES 2-4 illustrate the next three syncfaonization codes. As can be seen, the samples for each subsequent synchronization code are shifted by one sample bin relative to the samples for the previous sequence. Therefore, none of fee sequences interfere with each other. To generate the subsequent sequences, corresponding to TRACES 2-4, sequence v must be shifted in frequency. Thiscaiibeacconplishedusingthe following equation:

lor r = 1 to L (# of sequences) and m = 0 to 4*IA (time); aid where: I(m) = each subsequent sequence;;^ = the first sequence; and n = the number of times the sequence is repeated It will be understood that multiplying by an eφ(j2-π(r*m/N)) factor, where TV is equal to Ihe number of times the sequence is repeated n multiplied by the length of Ihe underlying perfect sequence L, in the time domain results in a shift in the frequency domain. Equation (6) results in the desired shift as illustrated in figure 9 for each of synchronization codes 24, relative to synchOnizalion code 1. The final step in generating each synchronization code is to append the copies of the last A/samples, whereλ/is the length of the multipath, to the front of each code. This is done to make the convolution with the multipath cyclic and to allow easier detection of the mulφath. It should be noted that synchronization codes can be generated from mem than one perfect sequence using the same methodology. For example, a perfect sequence can be generated aid repeated four times and then a second perfect sequence can be generated and repeated four times to get a n factor equal to dght The resulώ]gsequQκ:e can Ilien be shifted as described above to create the sjαichiuiization codes. b. Signal Measurements Using Svnclτronization Codes

Therefore, when a communication device is at the edge of a cell, it will receive signals from multiple base stations and, therefore, will be decoding several synchtanization codes at Hie same time. This can be illustrated with the help of figure 10, which illustrates another example embodiment of a wireless oommunication system 1100 comprising communication

cells 1102, 1104, aid 1106 as well as communication device 1108, which is in communication withbase station 1110 of cell 1102 but also receiving communication fora base stations 1112 aid 1114 of cells 1104 aid 1106, respectively. If communications from base station 1110 comprise synchronization code SYNCl and communications fiυm base station 1112 and 1114 comprise SYNC2 and SYNC3 respectively, then device HOS will effectively receive the sum of these Ihree synchronization codes. This is because, as explained above, base stations 1110, 1112, and 1114 are configured to transmit at the same time. Also, the synchronization codes arrive at device 11OS at almost the same time because they are generated in accordaice with the description above. Again as described above, Ihe syjxfoOnizarion codes SYNCl, SYNC2, and SYNC3 exhibit ideal cross coiieMon Therefore, when device 1108 correlates the sumx of codes SYNCl, SYNC2, aid S YNC3, Ihe latter two A vill not interfere with proper detection of S YNC 1 by device 1108. Inpoitaitly, the aim x can also be used to determine important signal characteristics, because the stun x is equal to the sum of the synchiOnization code signal in acooidaxewiththeMo\vύigeqLiatioi

ηierefore,when SYNCl is removed, tiiesim of SYNC2ardSYNC3 Meft, as shownin the following:

The energy computed from the sum (SYNC2 + SYNC3) is equal to the noise or interference seen by device 1108. Since the purpose of conelating the synchronization code in device 1106 is to extract the energy in SYNC 1, device 1108 also lias the energy in the signal from base station 1110, ie., the energyrepresented by SYNCl. Therefore, device 1106 can use the energy of SYNCl aid of (SYNC2 + SYNC3) to perfonn a signal-to-interference measurement for the communication enamel over which it is communicating with base station 1110. The result of the measurement is preferably a sigiial-tc^mterference ratio (SIR). The SIR measurement cai then be communicated back to base station 1110 for purposes thatwillbediscussedbelow. The ideal cross correlation of the synchronization codes, also allows device 1108 to perform extremely accurate detenninations of the Channel Impulse Response (CIR), or channel estimation, from the correlation produced by αxrelator 800. This allows for highly accurate equalization using low cost, low complexity equalizers, thus ovαcoming a sigificant draw back of conventional systems. 4. Sub-Channel Assignments

As mentioned, the SIR as determined by device 1108 can be communicated back to base station lllO forusein the assignment of channels 502. In one embodiment, due to the fact that each aib-chamel 502 is processed independently, the SIR for each aib-charmel 502 can be meaaired aid communicated back to base station 1110. Ih such an embodiment, therefore, sub-channels 502 can be divided into groups aid a SIR measurement for each group cai be sent to base station 1110. This is illustrated in figure 1 IA, which shows a wideband communication channel 1200 segmented into aib-chamels o tofis Sub-channels/? tof ₁₅ ae then grouped into 8 groups Gl to G8. Thus, in one embodiment, device 1108 and base station 1110 cαrimunicate ova a channel such as channel 1200.

Sub-chamels in the same group are preferably separated by as many sub-chamels as possible to ensure diversity. In figure 1 IA for exanple, sub-chamels within the same group ae 7 sub-channels apart, e.g., group Gl comprises^ stndβ. Device 1102 reports a SIR measurement for each of the groups Gl to GS. These SIR measurements ae preferably

compared with a threshold value to detemine which sub-channels groups ae useable by device 1108. This comparison can occiu" in device 1108 or base station 1110. If it occurs in device 1108, then device 1108 can simply report to base station 1110 wliichsub-channels groups are useable by device 1108.

SIR reporting will be simultaneously occurring for a plurality of devices within cell 1102. Thus, figure 1 IB illustrates the situation where two communication devices ∞nespσnding to User 1 and User 2 report SIR levels above the threshold for groups GL ₁ G3, G5, and G7. Base station 111 Opreferably then assigns sub-channel groups to User 1 and User 2 based on the SIR reporting as illustrated in Figure 1 IB. When assigning the ' 'good' ' sub-channel groups to User ¹ 1 and User 2, base station 1110 also preferably assigns them based on flieprirdplesoffequαxy diversity. Ih figure HB, therefore. User 1 and User 2 are alternately assigned every other "good" sub-channeL The assignment of sub-channels in the frequency domain is equivalent to the assignment of time slots in the time domain Therefore, as illustrated in figure 12, two users, User 1 and User 2, receive packet 1302 tansmitred ova communication cliannel 1200. Figure 12 also illustrated the sub-channel assignment of figure 1 IB. While figures 11 and 12 illustrate sub-channelλime slot assignment based on SIR for two users, the principles illustrated can be extended for any number of users. Thus, a packet within cell 1102 can be received by 3 or more users. Although, as the number- of available subchannels is reduced due to high SIR, so is the available bandwidth In other words, as available channels ae reduced, the number of users that can gain access to communication channel 1200 is also reduced.

Poor SIR can be caused for a variety of reasons, but frequently it results from a device at the edge of a cell receiving communication signals from adjacent cells. Because each cell is using the same bandwidth B, the adjacent cell signals will eventually raise the noise level and degrade SIR for certain sub-channels. In certain embodiments, therefore, sub-channel assignment can be coordinatedbetween cells, such as cells 1102, 1104, and 1106 hi figure 10, in order- to prevent interference from adjacent cells. Thus, if communication device 1108 is near the edge of cell 1102, and device 1118 is nearthe edge of cell 1106, then the two can interfere with each other. As a result, the SIR measurements that device 1108 and 1118 report back to base stations 1110 and 1114, respectively, will indicate that the interference level is too high. Base station 1110 can then be configured to assign only the odd groups, i.e., Gl, G3, G5, etc., to device 1108, while base station 1114 can be configured to assigi the even groups to device 1118. The two devices 1108 and 1118 will then not interfere with each other due to the coordinated assignment of sub-channel groups.

Assigning the sub-channels in this manner- reduces the overall bandwidth available to devices 1108 aid 1118, respectively. In this case the baxfwidth is reduced by a factor of two. But it should be remembered that devices operating closer to eadibase station 1110 aid 1114, respectively, will still be able to use all cliaπnelsifneeded. Thus, it is only devices, such as device 1108, that ae nea- the edge of a cell that will have the available bandwidth reduced Contrast this with a CDMA system, for example, in which the bandwidth for all users is reduced, due to the spreading techniques used in such systems, by approximately a factor of 10 at all times. It can be seen, therefore, that the systems and methods for wireless communication over a wide bandwidth channel using a plurality of sub-channels not only improves the quality of service, but can also increase the available baidwidth significantly. When there ae fee devices 1108, 1118, andlU6nea-the edge

oftlαeir respective adjacent cells 1102, 1104, and 1106, the aib-channels can be divided by three. Thus, device 1108, for example, can be assigned groups Gζ G4, etc., device 1118 can be assigned groups G2, G5, etc., aid device 1116 can be assigned groups G3, G6, etc. In this case the available bandwidth for these devices, i.e,, devices near Ihe edges of cells 1102, 1104, and 1106, is reducedby a factor of 3, but this is still better than a CDMA system, fcr example.

The manna.- in which such a coordinated assignment of sub-channels can work is illustrated by the flow chart in figure 13. Frist instep 1402, acommunication device, such as device 1108, reports the SIR for all sub-channel groups Gl to G8. The SIRs reported are then compared, in step 1404, to a threshold to determine if the SIR is sufficiently low for each group. Alternatively, device 1108 can make the determination aid simply report which groups are above or below the SIR threshold. Iftiie SIR levels are good for each group, then base station 1110 can make each group available to device 1108, in step 1406. Periodically, device 1108 preferably measures the SIR level and updates base station 1110 in case the SIR as cleteriorateά For example, device 1108 may move from near the cento.- of cell 1102 towad the edge, where inteiterence from an adjacent cell may affect the SIR for device 1108. Ifflie comparison in step 1404 reveals that the SIR levels ae not good, then base station 1110 can be rjieprøgrammed to assign eiftαer the odd groups or die even groups only to device 1108, which it will do in step 1408. Device 1108 then reports the SIR measurements for Ihe odd or evai groups it is assigned in step 1410, aid they are again compared to aSR threshold in step 1412. It is assumed thatthe poor SIRlevel is duetothe fact that device 1108 is operating at the edge of cell 1102 andistliereforebeinginteifei«iwithbyadesdce such as device 1118. But device 1108 will be interfering with device 1118 at the same time. Therefore, the assignment of odd or even groups in step 1408 μeferably corresponds with the assignment of the opposite groups to device 1118, by base station 1114. Accordingly, when device 1108 reports the SIR measurements for whichever groups, odd or even, ae assigned to it, the comparison in step 1410 should reveal that the SIR levels ae now below the threshold level Thus, base station 1110 makes the assigned groups available to device 1108 in step 1414. Again, device 1108 preferably periodically updates the SIR measuremaits by returning to step 1402.

It is possible for the conparison of step 1410 to reveal that the SIR levels ae still above the threshold, which should indicate that a third device, e.g., device 1116 is still interfering with device 1108. In this case, base station 1110 can be preprogrammed to assign every thid group to device 1108 in step 1416. This should correspond with the corresponding assignments of non-interfering chamels to devices 1118 and 1116 by base stations 1114 aid 1112, respectively. Thus, device 1108 should be able to operate on the sub-chaτnel groups assigned, Le., Gl, G4, etc., without undue mla'ference. Again, device 1108 preferably periodically updates Ihe SIR measurements by returning to step 1402. Optionally, a third comparison step (not shown) can be implemaited after step 1416, to ensure that the groups assigned to device 1408 posses an adequate SIR level for proper operation Moreover, if thae aremore adjacent cells, Le,, if it is possible for devices in a4 or even a 5 ^th adjacent cell to interfere with device 1108, then the process of figure 13 would continue and the sub-channel groups would be divided even further to ensure adequate SIR levels on the sub-channels assigned to device 1108. Even though theprocess of figure 13 reduces the bandwidth available to devices atthe edge of cells 1102, 1104, aid 1106,theSIR measirrements can be used in such a manner as to increase the data rate aid therefore restore or even increase bandwidth. To

ac∞mpMi1his _> 1hetiHτsrrώtøarrii^^ 1102, 1104, and 1106, and in devices in communication therewith, e.g, devices 11OS, 1114, and 1116 respectively, must be capable of dynamically changing the symbol mapping schemes used for some or all of the sub-chaπneL For example, in some embodiments, the symbol mapping scheme can be dynamically changed among BPSK, QPSK, 8PSK, 16QAM, 32QAM, etc. As the symbol mapping scheme moves higher, i.e., toward 32QAM, the SIR level required for proper operation moves higher, ie, less and less interference can be wilhstood Therefore, once the SIR levels are deteariined for each group, Hie base station, e.g, base station 1110, can then determine what symbol mapping scheme can be supported for each sub-channel group and can change the modulation scheme accordingly. Device 1108 must also change Hie symbol mapping scheme to correspond to that of the base stations. The change can be effected for all groups uniformly, or it can be effected for individual groups. Moreover, the symbol mapping scheme canbe changed on just the forward link, just the reverse link, or both, depending on the embodiment Thus, by maintaining the capability to dynamically assign sub-channels and to dynamically change the symbol mapping scheme used for assigned aib-channels, the systems and methods described herein provide the ability to maintain higher available bandwidths with higher performance levels than conventional systems. To My realize the benefits described, however, the systems and methods described thus far must be capable of implementation in a cost effect and convenient manner. Moreover, the implementation must include i^oiifigurability so that a single device can move between different types of communication systems and still maintain optimum perfonnance in accordance with the systems and methods described herein. The following descriptions detail example high level embodiments of hardware implementations configured to operate in accordance with Ihe systems and methods described herein in such a manner as to provide the capability just described above. 5. SarripleTransrnittei-Eiiibodiments

Figure 14 is logical block diagram illustrating an example embodiment of a transmitter 1500 ccαifigured for wireless communication in accordance with the systems and methods described above. The transmitter could, for example be within a base station, e.g, base station 606, or within a communication device, such as device 604. Transmitter 1500 is provided to illustrate logical components that can be included in a tansmitter configured in accordance with the systems and methods described herein. 1 is not intended to limit die systems and methods for wireless communication over a wide bandwidth channel using a plurality of sub-channels to any particular transmitter configuration or any particular wireless committiication system. With this in mind, it can be seen that transmitter 1500 comprises a serial-to-patallel converter 1504 configured to receive a serial data stream 1502 comprising a data rate R Serial-to-parallel converter 1504 converts data steam 1502 into iVparallel data streams 1504, whereNis the number of sub-channels 202. Ir should be ∞ted that while the discussion that follows assumes that a single serial data stream is used, more than one serial data stream can also be used if required or desired In any case, thedatarateof each parallel datastream 1504 is then&M Each data stream 1504 is then sent to a scrambler, encoder, and interieaver block 1506. Scrambling encoding and interleaving ae common techniques implemented inmany wireless communication transmitters andhelp to provide robust, secure communication. Examples of these techniques will be briefly explained for illustrative purposes.

Scrambling breaks up the data to be transmitted in an effort to smooth out the spectral density of the transmitted data For example, if the data, comprises a long siring of Ts, there will be aspike in the specttal density. This spilce can cause greater interference within the wireless communication system. By breaking up the data, Hie spectral density can be smoothed out to avoid any such peals. Often, scrambling is achieved by XORing the data with a random sequence. Encoding, or coding, the parallel bit streams 1504 can, for example, provide Forward Enυr Correction (FEQ. The purpose ofFECistoinprovethecapadtyofaconunuracalicttichame^ the data being transmitted through the channel. The process of adding this redundant information is known as channel coding, ConvoMonal coding and block coding are the two major forms of channel coding ConvoMonal codes operate on serial data, one or a few bits at a time. Block codes operate on relatively large (typically, up to a couple of hundred bytes) message blocks. There ae a variety of useful convolutional and block codes, and a variety of algorithms for decoding the received coded information sequences to recover the original data For example, convolutional encoding or turbo coding with Viteώi decoding is a FEC technique that is particularly suited to a channel in which the transmitted signal is corrupted mainly by additive white gaussian noise (AWGN) or even a channel that simply experiences lading

Convolutional codes ae usually described using two paameters: the code rate and the constraint length. The code rare, Mi, is expressed as aratio of the number ofhits into the convolutional encoder (k) to the number of channel symbols (h) output by the convolutional encoder in a given encoder cyde. A common code rate is 1/2, which means to 2 symbols ae produced for every 1-bit input into the coda". The constraint length parameter, K, denotes the 'length" of the convolutional encoder, i.e. how many kbit stages are available to feed the combinatorial logic to produces the output symbols. Closely related toUTisthe parameter n ι, which indicates how many encoder cycles an input bit is retained and used for encoding after it first appears at the input to the convolutional encoder. The m parameter can be thought of as the memory length of the encoder. Interleaving is used to reduce the effects of fading. Interleaving mixes up the order of the data so that if a fade interferes with a portion of the transmitted signal, the overall message will not be effected. This is because once the message is de-interleaved aid decoded in the receiver, the data lost will comprise non-contiguous portions of the overall message. In other words, the fade will interfere with a contiguous portion of the interleaved message, but when die message is de- interleaved, the interfered with portion is spread fliroughout the overall message. Using techniques such as FEC, the missing mfomτaticnαωthenbe:ffled-n,OTtte

After blocks 1506, each parallel data streaυ 1504 is sent to symbol mappers 1508. Symbol mappers 1508 apply Hie requisite symbol mapping, e.g, BPSK, QPSK, etc., to each parallel data stream 1504. Symbol mappers 1508 are preferably programmable so to the modulation applied to parallel data streams can be changed, for example, in response to fiie SIR reported for each sub-channel 202. It is also preferable, to each symbol mapper 1508 be separately programmable so that the optimum symbol mapping scheme for each sub-channel can be selected aid applied to each parallel data streari 1504. After symbol mappers 1508, parallel datastreams 1504 are sent to modulators 1510. Important aspects and features of exarrple embodiments of modulators 1510are described below. Aflermodulators 1510, parallel datastreams 1504 are sent tosummer 1512, whidi is configured to sijmttetrørafld data streams and tb^ 1518

comprising each of the individually processed parallel data steams 1504. Serial data stream 1518 is thai sent to radio module 1512, where it is modulated with an RF carrier, amplified, and transmitted via antenna 1516 according to known techniques.

The transmitted signal occupies the entire bandwidthi? of ∞niiiunication channel 100 and comprises each of the discrete parallel data streams 1504enccdedontotiieirrespectivesul>chamiels 102 wifl±i bandwidth 5. Encoding parallel datastreams 1504 onto tiieφpropriatesub-chainels 102 requires that each parallel data stream 1504beshMedinfiτ£quency by an appropriate offset This is achieved in modulator 1510. Figure 15 is a logical block diagram of an example embodiment of a modulator 1600 in accordance with the systems and methods described herein. Lnpoitantly, modulator 1600 takes parallel data streams 1602 performs Time Division Modulation (TDM) or Frequency Division Modulation (FDM) on each data stream 1602, filters Ihem using filters 1612, and then shifts each data stream in frequency using frequency shifter 1614 so that they occupy the appropriate sub-channel. Filters 1612 apply the required pulse shaping, Le., they apply the roll-off factor described in section 1. The frequency shifted parallel data streams 1602 are then summed and transmitted. Modulator 1600 can also include rate controller 1604, frequency encoder 1606, aid interpolators 1610. All of die components shown in figure 15 ae described in more detail in the following paragraphs aid in conjunction with figures 16-22.

Figure 16 illustrates one example anbodiment of a rate controller 1700 in accordance with ihe systems and methods described herein Rate control 1700 is used to control the data rate of each parallel data stream 1602. Bi rate controller 1700, the data rate is halved byrepeating data steams d(0) to d(I), for example, producing streams α(0) to α(15) in which Ct(O) is die sane as α(8), α(l) is the sane as α(9), etc. Figure 16 also illustrates that the effect of repeating Hie data streans in this mamer is to take the data streaiis that ae encoded onto the first 8 sub-channels 1702, and duplicate them on flienext 8 sub-channels 17O2.Ascaibeseen,7sub-chainelssepaateaib-channels 17CGooi'nprisingthesaiie _i ordiplicate, data streams. Thus, if fading effects one sub-enamel 1702, for example, ihe other sub-channels 1702 carrying the same data will likely not be effected, Le., there is frequency diversity between the duplicate data streams. So by sacrificing data rate, in this case half the data rate, more robust tansmission is achieved Moreover, the robustness provided by duplicating the data streams d(0) to d(7) cai be further enhanced by applying scrambling to the duplicated data streams via scramblers 170S. It should be noted that the data rate can be reduced by more than harζ e.g, by foui-ormore. Alternatively, die dataiate can also be reduced by an anount other thai half For example if information from n data strεan is encoded onto m sub-channels, where m >n. Thus, to decrease the rate by 2/3, information from one data strean can be encoded on a first sub-channel, infomiation from a second data stream can be encoded on a second data channel, aid the sum or clifference of the two data steams can be encoded on a third chainel Si which case _> proper scaling will need to be applied to the power in the third channeL Otherwise, for example, the power in the third chamel can be twice Hie power in the first two. Preferably, rate controller 1700 is programmable so that the data rate can be changed responsive to certain operational factors. For example, if the SIR reported for sub-channels 1702 is low, then rate controller 1700 can be programmed to provide more robust transmission via repetition to ensure that no data is lost due to interference. Additionally, clifferent types of wireless

conminicaiion system, e.g,, indoor, outdoor, lhe-of-sight, may require varying degrees of robustness. Thus, rate controller 1700 can be adjusted to provide the minimum required robustness for the particular type of commiuiication system This type of programmability not only ensures robust commiuiication, it can also be used to allow a single device to move between communication systems and maintain superior performance.

Figure 17 illustrates an alternative example embodiment of a rate controller 1800 in accoitlance with the systems and methods described. Si rate controller ISOO the data rate is increased instead of decreased. This is accomplished using serial-to- parallel converters 1802 to convert each data steams d(0) to d(15), for example, into two data streams. Delay circuits 1804 then delay one of Hie two data streams generated by each serial-to-parallel converter 1802 by 1/2 a symbol. Thus, data streams d(0) to d(15) ae frarisfcmied into data streams a(0) to aβl). The data streams generated by a particular serial-to-paallel converter 1802 aid associate delay circuit 1804 must then be summed and encoded onto the appropriate sub-channeL For example, data streams a(0) and a(l) must be summed and encoded onto the first subchannel. Preferably, Hie data streams ae armmed subsequait to each data stream being pulsed shaped by a filter 1612. Thus, rate controller 1604 is preferably programmable so that the data rate can be increased, as in rate controller 1800, or decreased, as in rate controller 1700, as required by a paticular type of wireless communication system, or as required by the communication channel conditions or sub-channel conditions. Ih the event that Hie data rate is increased, filters 1612 are also preferably programmable so that they can be configured to apply pulse shapping to data streams a(0) to a(31), for exaiiple, and then sum Hie appropriate streams to generate Hie appropriate number of parallel data streams to send to frequency shifter 1614. The advantage of increasing the data rate in Hie manner illustrated in figure 17 is that higher symbol mapping rates can essentially be achieved, wiHiout changing Hie symbol mapping used in symbol mappers 1508. Qnce Hie data streams are summed, Hie summed streams are shifted in frequency so that Hiey reside in Hie appropriate sub-channeL But because Hie number ofbits per each symbol has been doubled, Hie symbol mappingrate lias been doubled. Thus, for example, a4QAM symbol mapping can be converted to a 16QAM symbol mapping, even if Hie SIR is too high for 16QAM symbol mapping to oHierwise be applied. fiioHier-woitHpiOgrBmmingiatecontrOllei- 1800 to increase Hie datarate in Hie manner illustrated in figure 17 can increase Hie symbol mapping even when channel conditions would otherwise not allow it, which in turn can allow a commiuiication device to maintain adequate or even superior performance regardless of Hie type of communication system The draw back to increasing Hie data rate as illustrated in figure 17 is fliat interference is increased, as is receiver complexity. The foniier is due to Hie increased amount of data. The latter is due to Hie fact that each symbol cannot be processed independently because of Hie 1/2 symbol overlap. Thus, Hiese concerns must be balanced against HK increase symbol mapping ability when irnplanenting a rate controller such as rate controller 1800.

Figure 18 illustrates one example embodiment of a frequency encoder 1900 in accordance with Hie systems and methods described herein. Similar to rate encoding, frequency encoding is preferably used to provide increased corimuinication robustness. In frequency encoder ^* 1900 the sum or difference of multiple, data streams are encoded onto each sub-channeL This is accomplished using adders 1902 to sum data streams d(0) to d(7) wifli data streams d(8) to d(15), respectively, while adders 1904 subtract data streams dφ) to d(7) from data streams d(8) to d(15), respectively, as shown

Tlius, data streams a(0) to a(15) generated by adders 1902 and 1904 comprise information related to more thai one data streams d(0) to d(15). For example, a(0) comprises the sum of d(0) and d(8), i.e, d(0) + d(8), while a(8) comprises d(8) - d(0). Therefore, if either a(0) or a(8) is not received due to fading, for example, then both of data streams d(0) and d(8) can still be retrieved fiαn data streamα(S).

Essentially, tlie relationship between data stream d(0) to d(15) and a(0) to a(15) is a matrix relationship. Thus, if the receiver knows the correct matrix to apply, it can recover the sums and differences oϊd(0) to d(15) fiiαm a(0) to a(15). Preferably, frequency encoder 1900 is programmable, so to it can be enabled and disabled in order to provided robustness whenreqirøl Preferable, adders 1902 and 1904a^progιwimτablealsosotlτatαiπQ^nBMc^(^nbeφpKedtθ( ^)to d(15). After frequency encoding, if it is included, data steams 1602 ae sent to TDMFDM blocks 1608. TDMFDM blocks 1608 perform TDM or FDM on the data streams as required by the particular embodiment. Figure 19 illustrates an exarηpfeembodime^ is provided to illustrate the logical components that can be included in aTDMFDM block configured to perform TDM on a data stream Depending on the actual implementation, some of the logical components may or may not be included TDM/FDMblodc2000∞mprisesasu^^ aib-blockrepeater2008, andasync inserter 2010. Sub-block repeater 2002 is configured to receive a sub-block of data, such as block 2012 comprising bits a(0) to a(3) for example. Sub-block repeater is then configured to repeat block 2012 to μυvide repetition, which in turn leads to more robust communication. Thus, sub-block repealer 2002 generates block 2014, which comprises 2 blocks 2012. Sub-block scrambler 2004 is then configured to receive block 2014 and to scramble it, thus generating block 2016. Cne method of scrambling can be to invert half ofblock 2014 as illustrated in block 2016. But other scrambling methods can also be inplemented depending on the embodiment

Sub-block terminator 2006 takes block2016 generated by aib-block scrambler 2004 and adds a termination block 2034 to Hie front ofblock 2016 to form block 2018. Termination block 2034 ensures 1hat each block can be processed indepαidαώy in tiie receiver. Vλthouttenτιinatiαi block 2034, some blocks may be delayed duetoniultipalh, for example, and they would therefore overlap pat of the next block of data But by including termination block 2034, the delayed block cai be prevented fiϋm overlapping aiy of file actual data in the next block Termination block 2034 can be a cyclic prefix termination 2036. A cyclic prefix temiination 2036 simply repeats the last few symbols ofblock 2018. Thus, for example, if cyclic prefix termination 2036 is three symbols long, then it would simply repeat the last three symbols ofblock 2018. Alternatively, termination block 2034 can comprise a sequence of symbols to are known to both the transmitter and receiver. The selection of what type ofblock termination 2034 to use can impact what type of equalizer is used in the receiver. Therefore, receiver complexity aid choice of equalizers must be considered when determining what type of terrriination block 2034 to use in TTMFDM block 2000. After sub-block terminator 2006, TDMFDM block 2000 can include a sub-block repeater 2008 configured to perform a second block repetition step in which block 2018 is repealed to form block 2020. In certain embodiments, sub-block repeater can be configured to perform a second block scrambling step as welL After sub-block repeater 2008, if included, TDMFDM block 2000 comprises a sync inserter 210 configured to

paiodically insert an appropriate synchronization code 2032 after a predetermined number ofblocks 2020 and/or to insert known symbols into eachblodc The purpose of sjirhonization code 2032 is discussed in section 3.

Figure 20, on the other hand, illustrates an example embodiment of a TDMFDM block 2100 configured for FDM, which comprises sub-block repeater 2102, sub-block scrambler 2104, block coda" 2106, sub-block transformer 2108, sub-blcdctemiir^or 2HO, aid sync ir^^ block 2114 and generates block2116. Sub-block scrambler tiien scrambles block 2116, generating block 21 IS. Sub-block coder 2106 takes block 2118 and codes it, generating block 2120. Coding block correlates lie data symbols together and generates symbols b. This requires joint demodulation in the receiver, which is more robust but also more complex. Sub- block transformer 2108 then performs a tiHEformatioii on block 2120, generating block 2122. Preferably, the transfoimation is an IFFT of block 2120, which allows for more efficient equalizers to be used in the receiver. Next, sub- block terminator 2110 terminates block 2122, generating block 2124 and sync inserter 2112 periodically inserts a syndironization code 2126 after a certain number ofblocks 2124 and/or insert known symbols into each block Preferably, sub-block terminator 2110 only uses cyclic prefix termination as described above. Again this allows for more efficient receiver designs. TDMZFDM block 2100 is provided to illustrate ύie logical components that can be included in a TDMZFDM block configured to perfomi FDM on a data stream. Depending on the actual implementation, some of the logical components may or may not be included. Moreover, TDMZFDM block 2000 and 2100 ae preferably programmable so that the appropriate logical components can be included as required by a particular mplementation. This allows a device that incorporates one ofblocks 2000 or 2100 to move between different systems with different reqiώanents. Further, it is preferable that TDMZFDM block 1608 in figure 15 be programmable so that it can be programmed to perfomi TDM, such as described in conjunction with block 2000, or FDM, such as described in conjunction witii block 2100, as required by a particular communication system. Afier TDMZFDM blocks 1608, in figure 15, the parallel data streams are preferably passed to interpolators 1610. Afier Interpolators 1610, the parallel data streams are passed to filters 1612, which apply the pulse shapping described in conjunction with the roll-off factor of equation (2) in section 1. Thai the parallel data streams are sent to frequency shifter 1614, which is configured to shift each parallel data stream by the frequency offset associated with the sub-chaπneltowliichtlie particular paiallel data stream is associated

Figure 21 illustrates an example embodiment of a frequency shifter 2200 in accoidance with the systems aid methods described herein. As can be seen, frequency shifter 2200 comprises multipliers 2202 configured to multiply each parallel data stream by the appropriate exponential to achieve the required frequency shift. Each exponential is of the form: exp(j2T$jiT/rM), where c is the conesponding sub-channel, e.g, c = 0 to N-I, and n is time. Preferably, frequency shifter 1614 in figure 5 is programmable so that various charmelZsub-channel configurations can be accommodated for various different systems. Alternatively, an IFFT block can replace shifter 1614 and filtering can be done after the IFFT block This type of implementation can be more efficient depending on Ihe implementation After the parallel data streams are shifted, they are summed, e.g, in summer 1512 of figure 14. The summed data stream is then transmitted using the entire bandwidth B of the communication channel being used But the transmitted data stream also comprises each of the parallel

data steams shifted in frequency such that they occupy the appropriate sub-channel Thus, each sub-channel may be assigned to one user, or each sub-channel may cany a data steam intended for different users. The assignment of subchannels is described in section 3b. Regardless of how the sub-channels are assigned, however, each user will receive the enfe baidwidth, comprising all the sub-channels, but will only decode those sub-channels assigned to the user. 6. Sample Receiver Embodiments

Figure 22 illustrates an example embodiment of a receiver 2300 to can be configured in accordance with the present invention Receiver 2300 comprises an antenna 2302 configured to receive a message transmitted by a transmitter, such as transmitter 1500. Thus, antenna 2302 is configured to receive a wide band message comprising the entire bandwidth B of a wide band channel 1hat is divided into sub- channels ofbandwidth b. As described above, the wide band message comprises aplurality of messages each encoded onto each of a corresponding sub-channel All of the sub-channels may or may not be assigned to a device that includes receiver 2300; Therefore, receiver 2300nτay or nrayrrot be required to decode all of the sub-channels. After Hie message is received by antenna 2300, it is sent to radio receiver 2304, which is configured to remove the carrier associated with the wide band communication channel and extract a baseband sigial compiising die data stream transmitted by the transmitter. The baseband signal is then sent to correlator 2306 and demodulator 2308. Conelator 2306 is configured to correlated with a synchranization code inserted in Hie data stream as described in section 3. It is also preferably configured to perform SIR and multipath estimations as described in section 3(b). Demodulator 2308 is configured to extract Hie parallel data streams from each sub-channel assigned to the device comprising receiver 2300 aid to generate a single data stream therefiom.

Figure 23 illustrates an example embodiment of a demodulator 2400 in accordance with Hie systems and methods described herein. Demodulator 2402 comprises a frequency shifter 2402, which is configured to apply a frequency onset to Hie baseband data stream so Hiat parallel data streams compiising Hie baseband data stream can be independently processed in receiver 2400. Thus, the output of frequency shifter 2402 is a plurality of parallel data streams, which are then preferably filtered by filters 2404. Filter's 2404 apply afilterto each paallel data stream that correspondstothepulsesliape applied in Hie transmitter, e.g, transmitter 1500. Alternatively, an IFFT block can replace shifter' 1614 aid filtering can be done after the IFFT block This type of implementation can be more efficient depending on Hie implementation. Next, receiver 2400 preferably includes decimators 2406 configured to decimate the data rate of the parallel bit streams. Sampling at higher rates helps to ensure accurate recreation of Hie data But Hie higher Hie data rate, the larger aid more complex equalizer 2408 becomes. Thus, Hie sampling rate, and Hierefore Hie number" of samples, can be reduced by decimators 2406 to an adequate level that allows for a smaller and less costly equalizer ¹ 2408. Equalizer' 2408 is configured to reduce Hie effects of multipath in receiver 2300. Its operation will be discussed more fully below. After equalizer 2408, the parallel data streams are sent to de-scrambler, decoder, aid de-interieaver 2410, which perform the opposite operations of scrambler, encoder, and interieaver 1506 so as to reproduce the original data generated in Hie transmitter. The parallel data streans ae then sent to parallel to serial converter 2412, which generates a single serial data stream from the parallel data streams.

Equalizer 2408 uses the murtipafli estimates provided by conelator 2306 to equalize Hie effects of multipath in

receiver 2300. In one embodiment, equalizer 240S comprises Single-In Single-Out (SBO) equalizers operating on each parallel data stream in demodulator 2400. In this case, each SISO equalizer comprising equalizer 2408 receives a single input and generates a single equalized output Alternatively, each equalizer can be a Multiple-lh Multiple-Out (MMO) or a Multiple-lh Single-Out (MEO) equalizer. Multiple inputs can be required for example, when a fiequency encoder or rate controller, such as irequency encoder 1900, is included in the transmitter. Because irequency encoder 1900 encodes information from more than one parallel data stream onto each sub-channel, each equalizes comprising equalizer 2408 need to equalize more than one aib-channeL Thus, for example, if a parallel data stream in demodulator 2400 comprises d(l) + d(8), then equalizer 2408 will need to equalize both d(l) aid dβ) together. Equalizer 2408 can then generate a single output corresponding to d(l) or d(S) (WBSO) or it can generate both d(l) and d(S) (MMO). Equalizer 2408 can also be a time domain equalizer (IDE) or a frequency domain equalizer (FDE) depending on Ihe embodiment Generally, equalizer 2408 is a TDE if the modulator in the transmitter performs TDM on the parallel data streams, and a FDE if the modulator performs EDM But equalizer 2408 can be an FDE even ifTDM is used in the transmitter. Therefore, Ihe preferred equalizer type should be taken into consideration when deciding what type ofblock termination to use in the tansmitter. Because of power requirements, it is often preferable to use EDM on the forward Mc aid TDM on the reverse link in a wireless communication system As with transmitter 1500, the various components comprising demodulator 2400 are preferably programmable, so that a single device can operate in aplurality of different systems and still maintain superior performance, which is a primary advantage of the systems aid methods described herein Accordingly, the above discussion μovides systems and methods for mplementing a channel access protocol that allows the transmitter and receiver hardware to be rφrogrammed slightly depending on the communication system. Thus, when a device moves from one system to another, it preferably reconfigures the hatdwae, i.e. transmitter and receiver, as required and switches to a protocol stack corresponding to the new system. An impoitant part of reconfiguring the receiver is reconfiguring, or programming, the equalizer because muMpath is a main problem for each type of system. The multipath, hoλvever, vaies depending on the type of system, which previously lias meant that a different equalizer is required for different types of communication systems. The enamel access protocol described in the preceding sections, however, allows for equalizers to be used that need only be reconfigured slightly for operation in various systems. a Sample Equalizer Embodiment

Figure 24 illustrates an example embodiment of a receiver 2500 illustrating one way to configure equalizers 2506 in accordance with the systems aid methods described herein. Before discussing the configuration of receiver 2500, it should be noted that one way to configure equalizers 2506 is to simply include one equalizer per channel (for the systems and methods described herein, a channel is the equivalent of a sub-channel as described above). A correlator, such as conelator 2306 (figure 22), cai then provide equalizers 2506 with an estimate of the number, amplitude, and phase of any muWpaths present, up to some maximum number. This is also known as the Channel Impulse Response (CIR). The maximum number of muttipaths is detennined based on design criteria for a particular implementatioa The more muMpafhs included in the CIR the morepathdvαstythe receiver has aid the more robust conimuricaticninliiesystaTiwillbe.Pailidiversityis

discussed alitflemoie fullybelow.

If tee is one equalizer 2506 par channel, the CIR is preferably provided directly to equalizer 2506 from tlτe couelator (not shown). If such a conelator configuration is used, then equalizers 2506 can be mn at a slow rate, but the overall equalization process is relatively fast For systems with a relatively small number of channels, such a configuration is therefore preferable. The problem, however, is that there is large variances in the number of channels used in different types of communication systems. For exariple, an outdoor system can have has many as 256 channels. This would require 256 equalizers 2506, which would make the receiver" design too complex and costly. Thus, for systems with a lot of channels, the configuration illustrated in figure 25 is preferable. In receiver 2500, multiple channels share each equalizer 2506. For example, each equalizer can be shared by 4 channels, eg, Chl-Ch4, Ch5-Ch8, etc., as illustrated in figure 25. In which case, receiver 2500 preferably comprises a memory 2502 configured to store information arriving on each channel Memory 2502 is peferably divided into sub-sections 2504, which are each configured to store information for a particular subset of channels. Information for each channel in each subset is then alternately sent to the appropriate equalizer 2506, which equalizes the infonriatiori based on the CIRprovided for that channel In this case, each eqiraMzer-mustraimirch faster tliari it would if there was simply one equalizer ¹ per ¹ channel. For example, equalizers 2506 would need to run 4 or more times as fast in order ¹ to effectively equalize 4 channels as opposed to 1. In addition, extra memory' 2502 is required to buffer the channel infonnation But overall, the complexity of receiver" 2500 is reduced, because there ae fewer equalizer's. This should also lowerthe overall costto hrrplemert receiver 2500.

Preferably, memory 2502 and the number- of channels that are sent to a particular equalizer' is programmable. Ih this way, receiver 2500 can be reconfigured for the most optimum operation for a given system. Thus, if receiver 2500 were moved from an outdoor system to an indoor system with fewer- channels, then receiver 2500 can preferably be reconfigured so that there are fewer, even as few as 1, channel per equalizer. The rate at which equalizers 2506 are run is also peferably programmable such that equalizers 2506 can be run at the optimum rate for the number of channels being equalized In addition, if each equalizer- 2506 is equalizing multiple channels, then the CIR for those multiple paths must alternately be provided to each equalizer 2506. Preferably, therefore, a memory (not shown) is also included to buffer the CIR information for each channel. The appropriate CIR information is then sent to each equalizer from the CIR memory (not shown) when the conespondrhg channel information is being equalized The ClR memory (not shown) is also peferably programmable to enaueopthτrmnopαationregardlessofwlτattyρeof system receiver 2500 is operating in

Returning to the issue of path diversity, the number of paths used by equalizers 2506 must account for the delay spead d _s in the system. For example, if die system is an outdoor system operating in the 5 Giga Hertz (GHz) range, the communication channel can comprise a bandwidth of 125 Mega Hertz (MHz), e.g, the channel can extend from 5.725 GHz to 5.85GHz. If the channel is divided into 512 sub-channels with a roll-off factor r of .125, then each subchannel will have a bandwidth of approximately 215 kϋohertz (KHz), which pυvrdes appoximately a 4.6 microsecond symbol diπ^CHiSmce the worsteasedeky spread maximum of 5. Thus, there would be afirstpathPl at zerx> microseconds, a second pathP2 at 4.OmTaOSeCOnO ¹ S, a third path

P3 at 92 microseconds, afourthpaihP4 at 13.8 lracroseconds, aid fifBipatiiP5 at lS.4niicrøseconds, which is close to the delay spread <r4 Inanotheremlx)dήτieωl,asixthpMic^

20 microseconds is Hie worst case. In feet, a delay spread d _s of 3 niicroseconds is a more typical value. In most instances, therefore, tlie delay spread d _s will actually be shorter and an extrapath is not needed. Alternatively, fewα sub-channels can be used, thus providing a larger symbol duration, instead ofusing an extrapath. But again, this would typicallynot be needed

As explained above, equalizes 2506 are preferably configurable so that they can be reconfigured for various communication systems. Thus, for example, the number of paths used must be sufficient regardless of the type of communication system. But this is also dependent on Hie number of sub-channels used If, for example, receiver 2500 went from operating in the above described outdoor system to ai indoor system, where the delay spread d _s is on the order of 1 miαoseeond,thmircdvα25()0caiprefei^^^ bandwidth of 125 MHz, Hie bandwidth of each sub-channel is approximately 4 MHz and Ihe symbol duration is appiOximately 250 nanoseconds. Therefore, there will be a first path Pl at zero nicroseconds and aibsequent paths P2 to P5 at 250ns, 500ns, 750ns, and 1 microsecond, respectively. Thus, the delay spread ds should be covered for the indoor enviiOnment Again, Hie 1 microsecond delay spread d _s is worst case so the 1 microsecond delay spread d _s provided in the above example will often be more than is actually required This is preferable, however, for indoor systems, because it can allow operation to extend outside of the inside enviiOnment, e.g, just outside the building in which the inside enviiOnment operates. For campus style sivironments, where a user is likely to be traveling tetween buildings, this canbe advantageous. 7. Sample Embodiment of a Wireless Commuπicafioudevice

Figure 25 illustrates an example embodiment of a wireless communication device in accordance with the systems and methods described herein Device 2600 is, for example, apoitable communication device configured for operation in a plurality of indoor and outdoor communication systems. Thus, device 2600 comprises an antenna 2602 for ftansmitting and receiving wireless communication signals over a wireless communication channel 2618. Duplexor 2604, or switch, can be included so Ihat transmitter 2606 and receiver 2608 can both use antenna 2602, while being isolated from each other. Duplexors, or switches used for this purpose, are well known and will not be explained herein. Transmitter 2606 is a configurable transmitter ∞nfigured to implement Hie channel access protocol described above. Thus, transmitter 2606 is capable of ttansmitting and encoding a wideband communication signal comprising aplurality of sub-channels. Moreover, fransmitter 2606 is configured such that the various sub-components that comprise transmitter 2606 can be reconfigured, or programmed, as described in section 5. Similarly, receiver 2608 is configured to implement the channel access protocol described above and is, therefore, also configured such that Ihe various sub-components comprising receiver 2608 can be reconfigured, or reprogammed as described in section 6. Transmitter 2606 and receiver 2608 em interfaced with processor 2610, which can comprise vaious processing, controller, and/or Digital Signal Processing PSP) circuits. Processor 2610 contols the operation of device 2600 including encoding signals to be transmitted by transmitter 2606 and decoding signals received by receiver 2608. Device 2610 can also include memory 2612, which can be configured to store operating instructions, e.g, firmware/software, used by processor 2610 to control the operation of device 2600. Processor 2610 is also

preferably configured to reprogram transmitter 2606 and receiver 2608 via control interfaces 2614 and 2616, respectively, as required by fee wireless communication system in which device 2600 is operating. Thus, for example, device 2600 can be configured to periodically ascertain the availability is a prefared communication system. If the system is detected, then processor 2610 can be configured to load the corresponding operating instruction from memory 2612 and reconfigure transmitter 2606 andieceiver 2608 for opαationhtheprelened system.

For example, it may preferable for device 2600 to switch to an indoor wireless LAN if it is available. So device 2600 may be operating in a wireless WAN where no -wireless LAN is available, while periodically searching for the availability of an appropriate wireless LAN. Once the wireless LAN is detected, processor 2610 will load the operating instructions, e.g, the appropriate protocol stack, for the wireless LAN environment and will reprogram transmitter' 2606 and receiver" 2608 accordingly. Bi this manner, device 2600 can move fiorn one type of communication system to another; while maintaining superior perfonnance. It should be noted that a base station configured in accordance with the systems and methods herein will operate in a similar manner as device 2600; however; because the base station does notmove from one type of system to another, there is generally no need to configure processor 2610 to reconfigure transmitter 2606 aid receiver 2608 for operation in accordance with the operating instruction for a different type of system. But processor 2610 can still be configured to reconfigure, or reprogiam the sub-components of transmitter 2606 and/or receiver 2608 as required by the operating conditions within the system as reported by communication devices in communication with the base station Moreover, such abase station can be configured in accoridance with the systems and methods describedherein to implement more than one mode of operation. Si which case, controller 2610 can be configured to reprogram transmitter 2606 aid receiver 2608 to implement the appropriate mode of operation 8. High dalarate transmitter and receiver

Referring now to FIGS.2649, additional embodiments of the present invention are illustrated. The embodiments described below may contain some of the features and functionality as described above.

The embodiments of the present invention discussed below employ ultra-wideband cαnmunication technology. Referring to FIGS. 26 and 27, impulse type ultra-wideband (UWB) communication employs discrete pulses of electromagnetic energy that are emitted at, for example, nanosecond or picosecond intervals (generally tens of picoseconds to a few nanoseconds in duration). Fa- this reason, ultra-wideband is oftαi called "impulse radio." That is, the UWB pulses may be transmitted without modulation onto a sine wave, or a sinusoidal carrier, in contrast with conventional carrier wave communication technology. This type ofUWB generally requires neilher an assigned frequency nor apower amplifier ^* .

Anexampleofaconventiorralcamerwave∞nim^ 26. IEEE 802.1 lais a wireless local area network (LAN) μOtocoI, which transmits a sinusoidal radio frequency signal at a 5 GHz center frequency, with a radio frequency spread of about 5 MHz. As defined herein, a carrier wave is an electromagnetic wave of a specified fiequaxy aid anplitLidetø is erratt The802.11 protocolis an example of a carrier wave communication technology. The carrier wave comprises a substantially continuous sinusoidal waveform having a specific rraiOW radio frequency (5 MHz) that has a duration that may range from seconds to minutes. Ih

contrast, anultra-wideband (UWB) pulse may have a 2.0 GHz center frequency, with a fequency spread of approximately 4GHz, asshowninHG.27,wMchfflustratestwo1ypicalUWBpulses. HG.27 illustrates thattheshcrtertheUWBpulsein time, the broader 1he spread of its frequency spectrum This is because bandwidth is inversely proportional to the time duration of the pulse. A 600-picosecond UWB pulse can have about a 1.8 GHz center frequency, with a frequency spread of approximately 1.6 GHz and a 300-picosecondUWB pulse can have abouta3 GHzceπterfrequaxy,withajβEquency spread of approximately 3.3 GHz. Thus, UWB pulses generally do not operate within a specific frequency, as shown in FIG.26. Eiiher of 1he pulses shown in HG.27 may be frequency shifted, for example, by using heterodyning to have essentially the same barrdwidth but center And because UWB pulses are spread across an extrernelywide fii≤quencyiange, UWB communication systems allow ccmmurications atveryhighdatarates, such as 100 megabits per second or greater. Several differentmethods of ultra-wideband (UWB) ∞rnmunications havebεenμ-oposed For wireless UWB communications in the United States, all offhe^me£bc<3smustmed: the constrain bythe Federal Communications Commission (FCQ in their Report and Order issued April 22, 2002 (ET Docket 98-153). Currently, the FCC is allowing limited UWB ccmmumcations, but as UWB systems are deployed, and additional experience with this new technology is gained, the FCC may revise its current limits and -dlowfcr expanded use ofUWB commuricarion technology. The FCC April 22 Rφort and Qderrequires that UWB pulses, or signals occupy greaterthan 20% fractional bandwidth or 500megaher1z, whichever is smaller. Fractional bandwidth is defined as 2times fee difference between the high and low 10 dB cutoff frequencies divided by Hie sum of tie high and low 10 dB cutoff frequencies. Specifically, the fractional bandwidth equation is:

where/, is the high 10 dB cutoff frequency, andj _/ ϊsthelow 10 dB cutoff frequency.

Stated difϊerently, fractional bandwidthistherpeicenta^ofasignal's center frequaxyfeatftie signal occupies. For example, a signal having a center frequency of 10 MHz, andabandwiά^of2MHz(ie. ₅ fiOm9to π MHz),hasa20% fractionalbandwidth. Trjatis ₎ oaiterfrequency,

HG.28 illustrates the ultra-wideband emission limits for indoor systems mandated by the April 22 Report and Order. TheRqx^ andQrdar∞nstrairβUWB com

GHz, with intentional emissions to not exceed 41.3 dBm/MHz. The report and order also established emission limits for hand held UWB systems, vehicular radar systems, medical imaging systems, surveillance systems, through-wall imaging systems, groundpenetøtingradarandoiher UWB systems. SwiUbeappreciatedtbattheinventimdesc^^ employedindoors, and/or outdoors, andmaybe fixed, and/oπnobile, andmay employ either a wireless orwire media for a ccmmunicationchaπneL

Generally, in the case of wireless ccmmuracations, a multiplicity ofUWB pulses, or signals may be transmitted at relatively low power density (rnilliwatis per megahertz). However, an atenative UWB communication system, located

outside the United States, may transmit at ahigherpower density. For example, UWB pulses or signals may be transmitted between 30 dBm to -50 dBm.

UWB pulses, however, transmitted through many wire media will not interfere with wireless radio frequency tiansmissions. Therefore, die power (sampled at a single frequency) of UWB pulses transmitted tfiough wire media may range from about +30 dBm to about -140 dBm The FCCs April 22 Report and Order does not apply to communications through wire media

Commiuiication standards committees associated with die International Institute of Electrical aid Electronics Engineers (IHER) ae considering a number of iilta-wideband (UWB) wireless communication methods that meet the constraints established by die FCC. One UWB communication melhod may transmit UWB pulses that occupy 500 MHz bands wilhin die 7.5 GHz FCC allocation (from 3.1 GEHz to 10.6 GHz). In one embodiment of tiiis comrυunication mefliod, LWB pulses have about a 2-nanosecond duration which conesponds to about a 500 MHz bandwidth. The center frequency of die UWB pulses can be varied to place diem -wherever desired within die 7.5 GHz allocation Ih aiotiier embodiment of diis communication method, an Inverse Fast Fourier Transform (IFFT) is peribmied on parallel data to produce 122 carriers, each approximately 4.125 MHz wide. In tiiis embodiment, also known as Orthogonal Frequency Division Multiplexing (OFDM), die resultant LWB pulse, or signal is approximately 506 MHz wide, and has approximately 242-nanosecond duration It meets die FCC rules for UWB communications because it is an aggregation of many relatively narrowband earners raflier tiiai because of die duration of each pulse.

Another UWB communication method being evaluated by die THFF, standards committees comprises transmitting discrete UWB pulses tiiat occupy greater tiian 500 MHz of frequency spectrum. For example, in one embodiment of tiiis communication method, LWB pulse durations may vay from 2 nanoseconds, which occupies about 500 MHz, to about 133 picoseconds, which occupies about 7.5 GHz of baidwidtii That is, a single UWB pulse may occupy substantially all of die entire allocation for communications (from 3.1 GHzto 10.6GHz).

Yet aπotiier UWB communication mediod being evaluated by die IEEE staidards committees comprises transmitting a sequence of pulses that may be approximately 0.7 nanoseconds or less in duration, and at a chipping rate of approximately 1.4 gjga pulses per" second. The pulses ae modulated using a Direct-Sequence modulation technique, and is called DS-UWB. Operation in two bands is contemplated, witii one band is centered nea" 4 GHz widi a 1.4 GHz wide signal, while die second baid is centered near" 8 GHz, with a 2.8 GHz wide UWB signal. Operation may occur at eidier or bodi of die UWB bands. Data rates between about 28 Megabits/second to as much as 1320 Megabit^second ae contemplated

Anotiier mediod of UWB communications comprises ftaismitting a modulated continuous carrier wave where die frequency occupied by die transmitted signal occupies more dial die required 20 percent fractional bandwidth. In this metiiod the continuous carier wave may be modulated in a time period diat creates die frequency baid occupaicy. For exaiφle _> ifa4GHz earner is mc>άϊilated us^

the resultant signal may occupy 1.3 GHz ofbandwidlh aound a center fiequency of 4 GHz. Li this example, the fractional baαdwiddαis approximately 32.5%. Thissign.ύwoddbeαmsida«iUWBιinderte

Thus, described above are four different methods of ultra-wideband (UWB) commiiriication It will be appreciated that the present invention may be employed by any of the above-described UWB methods, or otheis yet to be developed.

Referring now to FIG.29, which illustrates ablock diagram of a transmitter 5210 consistent with one embodimait oflhepresent invention htliisembodimetitdataSllOofinta^nTaybepiovidedtodatainterik^ SCMO. Anumberofdata interfaces 5040 are known in the at and can be used to practice 1he cunent invention The data interface 5040 may include an industry standard such as a Universal Serial Bus (USB) standad interlace, ai TFHF, 1394 standard mterface, a Peripheral Component Interconnect standard (PCI), a Peripheral Component Interconnect Express (PQ-Express) standad, a MHSPEC-1760 standad, aid a MILSPEC-1553 standard. Non-industry standad interfaces may also be employed and the present invention is not limited with respect to Hie type of data interlace 5040 used. Data 5110 is sent ficm data irtetace 5O4OtofoeMβdiimA∞^ fonτraplitta%offiames5100.

As illustrated inFIG.31, a data fiame5100 comprises amedium access contiOlheader5120,adatasection5110,a source ID, a destination ID, a rate field 5130, and in some embodiments may include a Cyclical Redundancy Check 5115 (CRQ appended to the end of Ihe fame 5100. Referring back to EIG.29, the data fames 5100 are then sent to a baseband processor 5020, which performs a number of fimetions (described below) aid produces baseband flame 5050, illustrated in HG.33.

A '"fane" as defined ha^ whether a date include maiy different constructions and arrangements. Generally, a "fame" usually consists of a representation of the original data to be transmitted (generally comprising a specified number' ofbits, or binary digits), together" with other bits that may be used tor enυr detection or control. A "flame" may also include routing information, such as a source address, a destirMonadάess, andotlieririformatiori A "flame" may be of afferent lengths, and contain variable amounts of data It will be appreciated that die construction of baseband flame 5050 aid data flame 5100 may vary without exceeding the scope of the present invention

For example, additional bits in a "flame" may be used for routing (possibly in the form of an address field), syncliOnization, overhead information not directly associated with the original data, a flame check sequence, aid a cyclic reduTidaicy check (CRQ, anong others. CRC is ai error detection algorithm that is known in the at of communications. One embodimαit of a CRC may be described as follows. Gr ^' vai a data section 5110 having bits of length "fc," the transmitter 5210 generates ai n-bit sequence, known as Ihe Frame Check Sequence (FCS) such that by appending the FCS to the data section 5110, the resulting data section 5110 lias a length k+n. Tlαe FCS is calculated in such away that when a receiver divides Ihe received resulting data section 5110 by apredeterrnined number there is no remainder. Ifno iernainder is found the data sectionSllOisassumedtobe error flee.

HG.33 illustrates the basebaid fiame 5050 produced by the baseband processor 5020. The baseband feme 5050 comprises a physical layer header 51S0, the medium access control header 5120 and a number of data packets 5200. Each data packet 5200 includes a code block 5190, which is used by the receiver 5220 (shown in FIG.34) for syndirønization of flie packet 5200. Additionally, the basebaid fiame 5050 may include the FCS used to decode the CRC as described above. Physical layer header 5180 may comprise a number of sjαxlirorr-zation code blocks 5190 which are used by the receiver 5220 to synchronize its timing reference to the timing reference of the transmitter 5210.

Generally, syncIuOiiizarion is used to obtain a fixed relationship anong conesponding significant instants of two or more signals. Put diflerenily, synchonization (also known as fiame syixhiϋnization, fiame alignment, or flaming) is used by a receiver to lock onto an incoming fiame so that it may receive the data contained in the fiame. Generally, the receiver synchronizes its time base, or referencetothetime base of the transmitter.

For example, a ' ^ame syrdronization pattern,' ' generally comprising a incurring pattern ofbits, is transmitted that enablestiiereαavertoaHgnitsdoclςcα-timerefe^^ Repetition of the bit pattemhelps ensure that the receiver will have an opportunity to 'lock" in on tlie timing of theiixomingsignaL

In one αnbodπυent of the present invention, the synchronization code blocks 5190 ae comprised of 256 bit Golay codes. fo another emlxdhient, one or more of the Go It vvillbe appreciated that other types of synchtOnization codes, comprised of other bit sizes, maybe employed by the present invention. One feature of the present invention is that upon reception of the synchronization sequence, the receiver may adjust its time base, its frequency base, and a setting of an automatic gain control anplifier (not shown).

Returning to FIG.29, Hie baseband fiame 5050 is then sent from the baseband processor 5020 to modulator 5420, which contains a digital circuit 5080 and local oscillator 5090. Modulator 5420 perlbnns modulation of the basebaid fiame 5050, which includes representations of individual data bits, into a transmission signal 5070. That is, the baseband μυcessor 5020 outputs a signal comprised ofhigh and low signal values, each having a time duration, or time base TQ, shown in HG. 30, AvMchrepresenttliedata comprising the baseband fiame 5050.

The locd oscillator 5090 generates aclocksigiTal 5060 at a tiriie base Ti,iUustratedinFIG.30. Ih one embodiment of the μeseπt invention local oscillator 5090 maybe a voltage controlled oscillator. As mentioned above, the signal values representing the baseband fiame 5050 are at a time base To Using the clock signal 5060, the digital circuit 5080 modulates, or chaiges the signal values representing the baseband fiame 5050. Ih the illustrated embodiment the type of modulation is phase modulation

As shown in FIG.30, the inverse of the clock signal time base Ti is the center frequency of the transmission signal 5070. That is, 1/T ₁ = center frequency. It will be appreciated that virtually aiy center frequency can be employed by the present invention For example, the local oscillator 5090 may generate a clock signal 5060 with a time base Ti of 250 picoseconds. Ih this example, the digital circuit 5080 produces a transmission signal 5070 that would be centered at 4 Giga- Heitz (GHz), which is the inverse of 250 picoseconds. The inverse of the baseband fiame 5050 signal values (time base To) controls the amount of occupied bandwidth around the center" frequency of the tiansmission signal 5070. rh die above

example, if Hie time base T ₀ of tie baseband fiame 5050 signal values is 750 picoseconds, the transmission signal would occupy 1.3 GHz of bandwidlh, around a 4 GHz cater frequency. In this case the bandwidlh (Le., amount of radio frequency spectrum) occupied would extend from approximately 3.33 GHz to approximately 4.66 GHz. Hie fractional bandwidth of this signal, calculated by the formula given above, would be approximately 3325%. Thus, this transmission signal 5070 would be considered UWB under the current FCC definition because its fiaclionalbandwidlh exceeds 20%.

In another example, the clock signal 5060 time base Ti maybe approximately 133 picoseconds and the time base To of the baseband frame 5050 signal values may be approximately 146 picoseconds. The transmission signal 5070 in this case would have a center frequency of 6.85 GHz and the signal would occupy 7.5 GHz of bandwidth around the center frequency. In this example, the transmission signal 5070 would occupy the entire available UWB spectrum from 3.1 GHz to 10.6GHz. It woiddlrave a firatiorialba:riwidfh of approximately ^ Inyetanother example, a clock signal timebaseTi of approximately 2 nanoseconds with atime base T ₀ of the baseband frame 5050 signal values of approximately 5300 picoseconds yields a transmission signal 5070 that occupies 500 MHz ofbandwidth located around a center frequency of 3.35 GHz. The fractional bandwidth of this exemplary transmission signal 5070 is only approximately 15%. While this signal does not meet tie current UWB definition in terms of fractional bandwidth, it is still considered UWB since it occupies tie required minimum of 500 MHz ofbandwidth. In yet another example, a clock signal 5060 time base Ti may be approximately 100 picoseconds and the time base To of the baseband frame 5050 signal values may be 200 picoseconds. Li this example, the transmission signal 5070 would occupy 10 GHz ofbandwidth around a center frequency of 5 GHz. ThebandwidthcrajpiedbyfctøTsm zero Hertz up to 10 GHz. This signal would occupy a fractional bandwidlh of approximately 200%. Under the current UWB definition this signal would be a LWB signal but under the current FCC regulations would not be allowed for wireless transmission as aportion of the signal would bebelow the FCC mandated 3.1 GHz frequency boundary.

One feature of the present invention is that by generating a clock signal at the desired center fiequeπey used for transmission, tie present invention does not need to employ a mixer to position the signal at the transmission frequency. As discussed above, tie present invention can generate a signal anywhere within (or outside of) tie FCC mandated UWB radio frequency band by using a high-speed clock signal at tie desired frequency. This feature reduces the overall cost and complexity of tie device. In one embodiment, file high-speed clock is a 10.6 Giga-Hertz (GHz) clock, but it will be appreciated that other clocks, such as 4 GHz, 8 GHz, 12 GHz, and others may be employedbythe present mverώon.

Several embodiments of digital circuit 5080 are illustrated in FIGS.32a, 32b, aid 32c. One feature of tie digital circuits 5080 discussed below is tiiat they directly generate tie transmission signal, without mixing, or up-converting tie signal to tie radio frequency used for transmission.

Referring to FIG.32a, tie locally generated clock 5060 aid tie basebaid frame 5050 signal values are tie inputs to ai "exclusive or" function or gate. As is known in the art, and shown in TABLE I, ai "exclusive or" (KOK) gate performs tie following function:

TABLEI

As illustrated in FIG. 30, during time periods where signal values, or baseband data 5050 has a 'low" value, the "high" values in clock 5060 will cause the transmission signal 5070 to be "high." During time periods where the signal values 5050 are "low," ihe 'low" values will result in a "low" in the transmission signal 5070. Put differently, during 'low" signal value 5050 tinieperiods To, the tiansmission signal 5070 minors the clock 5060. During timeperiods Towhere signal values 5050 have a "high" value, ihe "high" values in clock 5060 result in 'low" values in Ihe transmission signal 5070. Additionally during "high" signal value 5050 time periods, the 'low" clock 5060 values result in a 'WgIi" transmission signal 5070. In other words, during 'liigh" signal values 5050 time periods, the inverse of clock 5060 becomes the transmission signal 5070. In this manner the signal values 5050modulate toephaseoftoetansmission signal 5070.

In an alternate embodiment of digital circuit 5080, illustrated in FIG.32b, signal values 5050 and clock 5060 are inputs into aα "aid" gate. Additionally, the inverse of clock 5060 and signal values 5050 are inputs into another "and" gate. Combiner 5160 may then passively combine Ihe outputs of the two "and" gates, or Junctions. As is known in the art, and shown in TABLE D, an ' 'and' ' gate perfornis the following logical fimctiorr.

TABLE π

In a like manner, and as illustrated in HG. 38, dining time periods To where the signal values 5050 are high, the output of the "aid" gate 5150a follows the clock 5060. When Hie signal values 5050 ate low, the output of "and" gate 5150 is 'low". The inverse of signal values 5050 and the inverse of clock 5060 are inputs to "and" gate 515Ob. During time periods where toe signal values 5050 are low, the inverse of signal values are ' "high. ¹ ' During this time period die transmission signal 5070 becomes the inverse of the clock 5060. The two outputs trom "and" gates 5150a and 515Ob may then be combined by combiner 5160 to produce transmission signal 5070. Si like manner to the "exclusive or" inφlementation described above, the phase of the clock 5060 is modulated by toe signal values 5050 to become transmission signal 5070. It should be noted that toe tansmission signal 5070 generated by the embodiment in FIG. 32a has an inverse phase relationship to toe transmission signal generated by toe embodiment shown in FIG 32b. Ether circuit may be modified by one with skill in toe art to produce the other sigial

Yet another embodiment of digital circuit 50S0 is illustrated in FIG.32c. This embodiment can produce either of the transmission signals 5070 shown in HG.30 and FIG.38, by reversing the inputs of clock 5060 and its inva.se. Ih this embodinteni, a 2:1 multiplexer 5170 is used to generate a transmission signal 5070. The clock 5060 and its inverse are connected to the multiplexer 5170. The signal values 5050 from the baseband Same are connected to Hie control So. When the signal value 5050 lias a low value, die signal present at input 0, clock 5060, is passed to the output transmission signal 5070. WIiQi the signal value 5050 lias a "high" value, the signal present at input 1, inva.se clock 5060, is passed to die output In this manner, fee clock 5060 is phase modulatedby signal values 5050 toproduce transmission sigial 5070.

Many spread spectrum communications technologies are known in the ait of communications. Generally, data to be transmitted is multiplied by a chipping code, where the time period of the code is refared to as a chip, or chip duration The chipping code usually lias a shorter duration time period than the signal value used to represent the data Stated otherwise, the chip duration is usually shorter ¹ than the data symbol, or signal value duration. The resulting sigial is a signal that occupies the bandwith of the chipping signal and caries the data signal. This bandwidth can be expressed as the inverse of die chip duration The ratio of chips per- data symbol is commonly referred to as the spreading factor. The process of multiplying the data signal by the chipping code is generally referred to as spreading the signal Ih 13® manner; the process in a receiver of recovering the data signal from a spread signal may be referred to as de-spreading. Ii conventional spread spectrum communications systems, the spread signal is then multiplied by a carrier wave to place the sigial at the radio fiequency used for transmission. Ih some communication systems, orthogonal codes are used to enable a multiple access scheme, where multiple users can communicate simultaneously.

The spreading factor introduces generally unwanted overhead into a commirπications system. For example, a data symbol could be transmitted without spreading. In this case, a spreading factor of I is employed, implying the data has not been spread. When using a spreading iactor of 256 the same data symbol would be 256 times lager than the same symbol using a spreading factor of 1. For example, ifaspreading factor of 1 isusedtosend 1 bit of data, then 1 bk is transmitted, lfa spreading factor of 256 is employed, then 256 bits are used to transmit 1 bit of data So, as the spreading factor increases, the amomit of data transmitted decreases.

One advantage of spreading the signal with a chipping code is that a receiver- may use the entire chipping code to recover Ihe signal. This process is commonly refeaed to as processing gain Processing gain, expressed in dB, assists die receiver in detection of the signal, which increases communication reliability. Another advantage of spreading with a chipping code, is that when orthogonal codes are employed in different networks, the users in one network will not intercept the signals of the users in other networks.

Jh one embodiment of die present invention, the transmission signal 5070 is spread by a chipping code or code block 5190, shown in FIG. 33. Ih one embodiment, portions of the transrnissiori signal 5070 have a different spreading factor. For example, the physical layer' header 51 SO may have a spreading factor of 256 where die medium access control header 5120 may have a spreading factor of 64. h. another embodiment of die present invention, the packets 5200 may have

a spreading factor that is dynamically controlled by the medium access controller 5030 that inserts the chosen spreading iactoriniBtefield5130of the medium access controlheader5120.

In 1his fashion, the spreading factor may be dynamically adjusted to accommodate a changing communication environment For example, if the distance the transmission signal 5070 must travel increases, the spreading factor may also increase, so that a receiver can recover the signal Qr, in a communication environment that is conducive to multipath, the spreading factor may also be increased. Alternatively, when the communication environment is favorable to ∞nmiurήcaiicftTSjiheφreadin^

Referring now to FIG.35, which illustrates some functions performed by baseband processor 5020. Frames 5100 are received from the medium access controller 5030 by the baseband processor 5020. The rate field 5130 in Hie medium access control header 5120 is evaluated to determine the data rate for Hie packets 5200. Based on the rate field 5130, FEC encoder 5300 applies a FEC (forward error correction, discussed below) αxoding level to the data frame 5100. For example, in one embodiment shown in TABLE ID, the baseband processor 5020 uses the rate field 5130 to set the FEC encoding and/or the spreading factor. Itwill be arpnxiatedthatdiiϊerent spreading factors, and/or FEC encoding levels may be employed by the present invention

TABLE EI

34

Encoding for Forward Enυr Correction (FEC) is a μocess by which redundancy is added to the data to be transmitted With the additional redundancy the receiver may then attempt to detect aid coniect errors in the received data An initial step in a FEC algorithm is to encode Hie data with additional bite. There are a number of FEC encoding algorithms. OfsignificantimpoitaiTcemconTinuiiicalionsae block codes and convolutional codes. Both types of encoding algorithms transform the original data set into a coded sequence of larger size. This increased size can yield a decrease in peifcmiance of infonnation throughput for a paticular data rate but may enable a more robust communication Mc In convolutional axoding the coded sequence depends not only on the current data bite being encoded but also on one or more previous data bite. fa convolutional coding the enαxling is performed on a continuous basis. In block encoding a distinct block of data bits is encoded by a code. The FEC encoding level, otherwise known as the coding efficiency is aratio of the original data to the encoded data fa other words, a FEC encoding level of ¹ A implies a 50% overhead or redundancy has been added to the data (50% more bits). Likewise, a ³ A EEC encoding level includes a 25% overhead A EEC encoding level of 1 means that no additional bite have been added to the data Other encoding rates <ss known in the at of communications aid maybe used Those encoding levels include l/S ⁴¹ rate encoding ¹ A rate encoding, 3/8 ^h rate encoding /2 rate encoding 5/8 ¹¹ ^e encoding 7/8 ^ύl rate encoding aκi ³ /4rate encoding

Referring again to FIG. 35, after the FEC encoder 5300 lias encoded the data, the data is then passed on to the interleaver 5310. Interleaving is a process by which the order of the bits to be tansmitted is changed One purpose of interleaving bite or a block of bite is to improve a communications systems' noise immunity. For exanple, if bite from different portions of the data rrame 5100 are interleaved, or mixed into a packet 5200 and that packet is corrupted by noise, or other factors during transmission, the impact of the conupted packet is distributed across multiple aveas of die data This reduces the number of potential errors in any contiguous block of data, thereby increasing the probability that a receiver can coniect the corrupted data

Arlei- the data is been interleaved, the data is forwarded to scrambler 5320. Sαambling the data reduces Ihe probability ofhaving long strings of simila data bite. Long strings of similar data bite may alter the distribution of transmitted power, known as Ihe Power Spectral Density (PSD), within die spectrum. In maiy cases it is advantageous to have the effect of 1he data on the PSD be minimal. In those instances the effect of data should be random, or white, within Ihe spectrum. A number of sαambling algorithms ae known in the art aid may be used to practice 1his embodiment of Ihe invention

The data is then sent to thespreader 5430. Depαidingon1heuifonωationmtheratefield5130aspiϊadiiigfact oris applied to the data As discussed above, the spreading factor may change based on the contents of the rate field 5130.

The spread data is then sent to the packetizer 5340 where it is broken into discrete blocks appropriate for each packet 5200. The synchronization generator 5350 generates syiichitOTzationcodeblocl«s5190foreachpacket The header generator 5330 generates aid forms the physical layer header 5180. The physical layer header 5180 is then appended onto

the medium access control header 5120. A completed baseband frame 5050 is then forwaided to the modulator 5420. It will be appreciated that Hie data processing oider described above may be changed, aid that other processing steps may be added or attracted

An exemplary receiver 5220 is depicted in FIG. 34. In one embodiment, an RF signal is received Irani the communication media (wire or wireless) by the RF fiont end 5010. The RF fiont aid 5010 sends the received signal to an aialog-to-digital convater (ADQ 5230. The ADC 5230 maybe a 1-bit ADC, a 2-bit ADC, a 3-bit ADC, a 4-bit ADC, a 5-bit ADC, a 6-bit ADC, a 7-bit ADC or an 8-bit ADC Other bit densities for ADCs are known in the art of commiuiications aid may be used to practice Hie invention Additionally, a number of ADC architectures are known in the ait aid may be used to practice Hie invention but will not be discussed here. In one embodiment of the present invention ADC 5230 is a 1-bit sigma delta ADC. Bi this embodiment, ADC 5230 sanples the RF signal aid creates a serial data signal. The serial data signal is sent to basebaid processor 5020 which converts, or reassembles the packets 5200 from die serial data signal into data flames 5110 which εas sent to the medium access controller 5030. The medium access controller converts the data flames 5110 into data 5100, which is sent to a data interface 5040. Data interface 5040 may comprise a number of different data interfaces as described above.

RF front end 5010 may comprise a number of components including one or more antennas for communications iiiawiidessmedajOrcouplhigdiuritsforcoiniiuπicfflionusingwi remedia Thebasebandproressor5020,asilhislrøtedin FIG.36, may comprise a poly-phase filter 5240, a de-spreader 5250, a channel impulse response detector 5260, a channel matched filter 5410, a de-scrambler 5270, a de-intaleavσ 5280 aid a FEC decoder 5290.

Ih one embodiment RF fiαit end 5010 may comprise two or more receive antennas (not shown). In this embodiment the receive antennas ae sepaated by a physical distaice from each other that approximates at least one wavelength of the center frequency of the signal the receiver is designed to receive. The wavelength is calculated by dividing the speed of light by the fiequeiicy. For example, a communication signal with a 4 GHz center frequency lias a wavelength of approximately 7.5 cm By sepasting multiple receive antennas by this distaice, the receiver lias a better chance of detamining which received signal is a direct path signal aid which is a multi-path signal. Additionally, the multiple receive antennas provide additional energy collection which may be used to detect the communication signal

Two embodiments of poly-phase filter 5240 ae illustrated in FIGS.37 aid 38. One function of the poly-phase filter 5240 is to down-convert the serial data signal into two lower frequency signals. The two signals ae commonly referred to as ii-phase (T) and Quadrature (Q). This conversion is accomplished by multiplying the serial data signal by a complex sinusoid. Since the serial data signal is discrete (having been sampled) the complex sinusoid is discrete sanples of a sinusoid The real aid imaginary pats of a conplex sinusoid may be calculated aid stored in a look-up table 5400. The serial data signal is split into two parallel signals by serial to parallel converter 5360. Serial to parallel convater 5360 merely outputs alternate sanples onto each output Multipliers 5370 multiply the samples by real and imaginary coefficients that represent the complex sinusoid Ih the embodiment illustrated in FIG.37, the resultant product signals are filtered by fitters 5380. In one embodiment, fitters 5380 are low-pass finite impulse response (FIR) fitters. FIR filters a^ known in the art of signal

processing and will not be discussed herein. S will be appreciated lhat oilier types of filters may be employed by the present invention Decimatois 5390 thai decimate the filtered signals. DecimationisapiOcessbywbichanumberofsaiiplesate discaded Li tlie embodiment illustratedinFIG.3S, decimation occuispriorto filteringtiie signals.

Returning to HG.36, Ihe poly-phase filter sends the resultant signal to the de-spreader 5250. The process of de- spreading the signal involves comdating tlie signal with a synchronization code block If the received signal contains the same, or an inverse of the syncliiOni2ation code block, the de-spreader finds a strong correlation, either positive or negative. The synclitonization code block may then be removed and replaced by a value. The de-spreader 5250 then sends tlie signal to tlie channel impulse response detector 5260 aid the channel matched filter 5410.

Qie feature of the present invention is that it provides ai adaptive matched filter system fliat can rapidly adjust to changing communication conditions. A wireless communication channel is generally characterized as a multipath lading channel, which includes multipafli signals that cause intessymbol interference. A conventional matched filter includes an estimated model of the communication channel, which is used to aid the matched filter in obtaining Hie strongest possible signal. However, when Hie estimated model does not accurately reflect the actual communication channel, file signal may be poorly recovered leading to apoor signal-to-noise ratio (SNR). A degraded SNR may result in ai mαeasedbk-eπor-rate (BER), or may reduce tlie effective range of the communication systeiu

Li die present invention, a channel impulse response detector 5260 is employed to provide a real-time aialysis of the actual communication channel to the channel matched filter 5410. Dining the detection of tlie physical layer header 5180, die channel impulse response detector 5260 measures the communication channel impulse response by ' listening ¹ ' for correlations at a number of time periods. Generally, the impulse response is detected from die time period in which strong correlations are found with tlie codes contained within the physical layer header 5180 aid code blocks 5190 (in packets 5200). A number of codes are known in the art, but exemplary codes may include Golay, Walsh and perfect code sequences.

In the presence of multipafli signal components, the de-spreader 5250 may correlate on delayed or multipafli copies of the intended signal Because of αWaaitpiυpagation path lengflis,m different time period than the intended signal. Li this situation, the channel impulse response detector 5260 provides the time of anival and strength of the multipafli copies to Hie channel matched filter 5410. The channel matched filter 5410 may flien sum flie received energy within the multipafli copies to provide a stranger signal strength. Li this way, the actual communication chamel characteristics ae detemiined and used to obtain the strongest possible signal.

Referring to FIG. 36, the channel matched filter 5410 may also include an equalization capability, or function. Generally, the goal of equalization is to provide as accurate ai estimate of the original data as possible. This "estimated" data can then be forwarded to subsequent processing blocks, such as the foiwad enor correction (FEQ decoder 5290. One function of the FEC is to detect aid correct eoDis in the estimated data If errors are detected, various remedial measures are performed. Thesemeasuieswillbedisακsedbelowin<xmecticn

The channel matched filter 5410 includes a simplified decision feedback equalization (DFE) function.

The multipath copies of previous signals may anive at a time when receiver 5220 is μυcessing a current signal. In this case the received signal may be made up of the intended signal and a number of multipath copies of previous signals. For example, Hie data signal may comprise the sum of 2 or more autocorrelation functions. In one case, the data signal may comprise the sum of 64 autocorrelation functions, or alternatively, the data signal may comprise the sum of 32, 128 or oilier stuns of autocorrelation functions. Asaresult

Where: Al* and A2* ae the complex conjugates of Al and A2. Ih this example the last four terms in equation (9) are what is known as Inter" Symbol Interference (ISl) or Inter" Chip Interference (ICI). Ih some cases where the symbol is substantially longer- than the chip duration the interference may be intra-symboL The first term is the data So, in this exaple, a current data sample, or decision, actually depends on the cunent sample as well as two past samples and two future samples. Obtaining die two past samples should not drive the complexity of the equalizer- however, obtaining two future samples does increase complexity. Accordingly, in one embodiment of equalizer of the present invention, the two future samples are ignored. In this embodiment, a "hard" estimation is employed This is in contrast with most conventional equalizers, which often depend on "soft" decisions. Tlieoutput from the hard decision is used to obtain the past two samples, which aemultiplied by the associated amplitude factors Al and A2 and combined. AsshowninFIG.40,theAl and A2 and other amplitude factors are represented by gr^ The coefBcients g^ me determined from the channel estimations. In one embodiment, the amplitude factors are supplied to the channel matched filter" 5410 fetn the channel impulse response detector 5260 during processing of the physical layer header 51 SO and during Hie processing of each packet 5200. Thus, the DFE converts the Mowing;

Whichbecomes:

The output of the DFE can then be passed onward for further processing. A general implementation ofDFE is an iterative process that significantly reduces the BI or IQ Further, the DFE may include a parity check, or the like, in order" to detect errors. If there ae no errors, then there is no reason to feedback the data and perform the iteration

As shown in FIG.36, in one embodiment, the channel matched filter- 5410 then sends the signal to de-scrambler 5270. Mai embodiment whae the trHisrτtitter 5210 scrambled the data, de-scranibler" 5270 de-scrambles 4ie data The de- scrambler" sends the de-scrambled data to the de-intαieaver 5280. In an embodiment where transmitter 5210 interleaved the data, de-interleaver 280 de-interleaves the data FEC decoder 5290 detects and corrects errors in the recovered data 5110. A number of decoding algorithms ae known in the art aid maybe used to practice the invention, h one embodiment the EEC decoding algorithm is a low density parity check (LDPQ algorithm.

There are a number" of error control methods known in the art of coniniunications. Generally speaking error control comprises two methods, error detection and error correction In most eπor detection algorithms, the received data is merely checked for error. If errors ae found, Ihe teansmitter may be notified and the data may be retransmitted, ϊi error

conection algorithms, Hie receiver attempts to correct detected errors. Ih. one class of algorithms, known as Forward Error Correction (FEQ ₅ extra bits εas transmitted with the data that can be used by the receiver to detect and correct errors in Hie data Hiat was received. Depending on Hie inplementation, Hie receiver can then ask Hiat the databe re-sent if too m any errors ae detected. Accordingly, as can be seen in HG.41 , an FEC encoder 3202 adds bits to an input data steam 3204 to create aioιitoιλdatastrøn3206fliatnecessailyiαpir^ MHτeexanpleoffiguτe32,EEC encoder 3202 is a ¹ A rate FEC encoder, which means fliat tor every input bit d _m FEC encoder 3202 adds a bit that can be used to detect errors when data stream 3206 is decoded. As discussed above other rate encoders, such as full rate, or ³ A rate encoders maybe employed by Hie present invention Thus, in Hie case of a ¹ A rate FEC encoder, data rate of output 3206 is twice that of input 3204. Data stream 3206 can flien be modulated and transmitted to a receiver. In Hie receiver, an FEC decoder 3208 cai be used to remove the extra bits and detect enors in Hie original data Thus, EEC decoder 3208 should match FEC encoder 3202, Le, EEC decoder 3208 should be a 1/2 rate FEC decoder, in Hie above example.

A problem with conventional EEC encoders and decoders is that Hie data rates can be too high for conventional technology. This can be especially true, for example, in an ultra-wideband application, where Hie data rates can be extremely high. One way to overcome this problem in accordance with Hie systems aid methods described herein is illustrated in FIG. 42, which depicts apoition of a transmitter chain 3300. Ii Hie example of figure 33, a data stream 3302, with a data rate (R) is fust split into a plurality of parallel data streams 3306 in serial to parallel converter 3304, each with a lower data rate (RAi) where n is Hie number of parallel data streams 3306. The parallel data streams 3306 can then be encoded using a plurality of EEC encoders. Here two encoders 3308 aid 3310 ae illustrated, Thus, each of FEC encoders 3308 aid 3310 can, depending on Hie inplementation, encode half as much data aid operate at a Iowa" speed fliai required in a conventional system. More generally, FEC encoders 3308 aid 3310 can be configured to assist each oflier with FEC axoding and reduce Hie overall load on each EEC encoder in Hie system. This, of course, requires some coordination, or message passing, between Hie two FEC encoders.

The outputs ofFEC encoders 3308 aid 3310 can then, for exaiple, be passed through parallel to serial converters 3312 and 3314 aid combined via combiner 3316 into a single data stream with EEC encoding The single data stream can Hien be optional filtered aid/orpulse shaped before beingmodulated aitransmitted,e.g,viaoptionalblock3318 _!

In another exaiple αnbodinient, of an EEC encoder configured in accordance with Hie systems and methods described herein, a code word is generated fiom an input data word by adding parity bits to the data wad as illustrated in FIG.43. Li Hiis exaiple embodiment, EEC encoder 3402, referred to as a Low Density Parity Check (LDPQ encoder, takes data woid 3404 aid generates output code woid 3406. As can be seen. Hie data word and code word are illustrated in matrixfomi Thus, forexanp^fliedatawordisaiiiatrixcoiiprisiiigp _m +irowsaid 1 column

LDPC is ai error correction algorithm where Hie data to be sent is encoded by a generator matrix and decoded bya parity matrix. Derivation of the two matrices is seen below in equation 12. The EEC encoder 3402, a "K" length block of data <? Kxi is multiplied by Hie generator matrix GN _X K _. which produces a "N" length block C na where N > K The

additional length is attributed to the overhead described above. The parity matrix may represent a connection of two types of nodes in the decoder. The locations of l's in the matrix represent Ihe connection of the two types of nodes.

The decoding of the block on receipt is usually an iterative process by which Ihe first type of node may calculate information related to the probability of the bit under consideration being a 1 or a 0. Ih some cases this probability may be expressed as a "log likelihood ratio'' ormathematically:

where In is the natural log, the numerator is the probability that the bit c, is a zero and the denominator is the probability it is a 1. This infonnation is passed to the other type of nodes specified by Hie parity check matrix, who perform a similar calculation based on the information received fiαn each of the first type of node. The second type of node then sends its calculation to each of Ihe first type of nodes it is connected to. This process continues until it is stopped or reaches some figure of merit in its result Since each node is connected to a number of nodes of the oilier type, each iteration improves the probability calculation at each node.

In one embodiment of an LDPC 3402, the code word can be generated using a generator matrix as illustrated by the following equation" where: GN _X K ϊS the Generator Matrix; N= M + k; R=IsN; andifR ⁼¹ /2, ihenM=lc

The generator matrix can, in turn, be generated fiυm an identify matrix and aparity check matrix as illustrated in the following equation; where: I=the identity matrix; andP=1he parity matrix.

Alternatively, aparity matrix H can be used to generate the code word C according to the following:

Thepairymatrix/fcantlienbedefiiieda

Accordingly, and dropping the subscripts for simplicity:

The goal now is to solve for P , since d isknαwn,Le.,itistheinputdata To facilitate finding P in one embodiment, Hf is configurad as adual diagonal matrix withMrows andMcolumns. Dual diagonal matrices ate well known and will not be described here; however and exemplary one is illustrated by Ihe following:

Further, H° can, depending on the embodiment, be foimed from a matrix of matrices. In one embodiment, this matrix of matrices is itself block cyclic. For example, in one embodiment, 4 matrices A, B, C and D can be used as in the following:

Here, each of the matrices 4 B ₁ C, and-D will have l</4 rows and W4 columns. Thus, an encoder and decoder configured in accordance with the systems aid methods described herein can be optimized for a dual diagonal ϊf and a block cyclic H ^D , as explained below. Many methods can be used to generate matrices A, B ₁ C ₁ and D consistent with the systems aadmethcidsdesraTDedheiein Qneexanplemethod,howevα;wiUte This example method will assume, for the time being, that k = 16 and therefore k/4 = 4. Then an identity matrix I cm be used, such as the following:

Each of matrices A, B ₁ C, and D can then be generated from this identity matrix I. For exanple, a permutation vector,intlτisexampleoflength4 ₃ caiαthenbeιisedtogeiiaωteyl Qfcoui^otliei-methodsforgeιieiatingnτatrices4-5 ₎ C, and D can be used consistently with the systems and methods described herein. Thus, the matrix A can, e.g, have the following fbmi, once an appropriate permutation vector is used to modify identity matrix 7:

Basically, as can be seen, apennutation vector can be used to shift 1he positions of the l's inidentitymatrix/. Ih one

embodiment, a single pemiutation matrix can be require! Qice the first matrix A is generated using the single permutation vector, then Iheother matrices B, Q axlZ ⁾ canbegenαatedbyshiilmgmatrixA For example, in one embodiment, each subsequent matrices B, C, sxήD is generated by shiflύigtlie previous matrix, starting with4 by 90°. Thus, i? would be as follows:

But as can be seen, in the example embodiment for generating matrices A, B, C, and D described above;, each row lias only a single 1. Ih one embodiment, Galois Field algebra (GF(2)) can be used to define the following equations for use in solving for P :

Thus, evenresults are equal to 0, while oddiesults are equal to 1. Nowieturningto the equation at issue:

This can be rewritten as: Butusingteequations(22),-l = l,1iαei^fore: ϋi one embodiment tiie following equation can beused Aauxάaφ

Eqιιation(27)caiibeiiiplemerλederrectrvelyif u can be gpiaated efficiently. Moneen±odimenζbasedonthe exanples above, iflc= 6, then z7 canbe detenrώied as follows:

This will result in the following equations:

ηieequationsof(29)definetiiefollowing^naal equation:

This equation then suggests a configuration for an IDPC encoder 3402, such as that illustrated in EIG.44. As can be seen, the ύ values are fed into Exclusive-OR (XOR) 3502, the output of which is fed tough a delay 3504 and back to Ihe other input of XOR 3502. A remaining issue, however, is the generation of the u tarns. Ih other wads, die equation (ϊf * P )= u as implemented by block 3506 should also be done in Ihe most efficient manner possible. In the example above, H° was pattitionedin 4s, therefore d shouldalsobepffl1itionedby4asillus1iatedin1liefollowiiig

The above equation can be implemented efficiently, for example, using a circuit such as the example ckuit illustrated in ElG.45. The circuit of EIG.45 is generalized for the situation where k = 128; however, it will be appreciated to the example embodiiωente described herein are not limited to any particular lengths or configurations. As can be Sean, the circuit ofFIG.45 uses a baik of cyclic shift registers 3606 to implement d . The outputs of shift registas 3606 can Ihen be passed to a plurality XORs 3602 as shown Thus, XORs 3602 collect the appropriate outputs from shift registers 3606 in order to generate Ihe ϋ terms. But since it is known, in the examples above, that the output of each cyclic shift register will only have one 1 , due to tie feet tot A, B, C, and D have only one 1 in each row, the outputs of cyclic shift registers 3606 can be rearranged and fixed so that, e.g, the first outputs of each go to the first XOR 3602, the second outputs go to 1he second XOR 3602, etc. Accordingly, efficient fixed connections 3608 can be used to reduce the complexity ofLDPC 3402. The 27

tarns can then be registered and fed to XOR 3502 as illustrated. Accordingly, if everything is segmented by 4's as illustrated in tile above examples, then the cyclic shift registers 3606 can be shifted k/4 times. Qi each clock cycle, k/4 of the solution would be generated, such that it takes k/4 cycles to get Hie entire solution This can result in a highly parallel encoder, such as that illustrated in figure 36, for high-speed operation The result can also be a low cost encoder, because Hie hardware can be reduced to l/4th that required by conventional circuits through the reuse of the components. The IDPC encoder ofFIG.44 can, therefore, be used to generate code word C, which can be modulated and tansrnitted But the receiver will receive C corrupted by noise as illustrated in the following;

The job of Hie decoder- is then to extract d from tie signal represented by equation 32. Ei one embodiment, this can be accomplislied by making soft decisions as to Hie value of x and combining it with hard decisions related to the sign x suchthat f/ c^iflienbeacmratelydelerrnined. ηiesoftdecisioiiscaibebasedcαiamiMevelpossibility. Forexample, if 4 bits are used in 2's complement, then you can have up to 16 levels. Si one embodiment, the levels can, for example, be from -8 to 7. Alternatively, using offset 2's complement, the levels can be from -7.5 to 7.5. An advantage of the later- is that the levels are not biased, e.g, toward the negative. An advantage of the former-, however, is that it includes the level 0. Qf course, any level scheme cai be used as long as it allows for accurate determinations of d .

The levels can be used to determine the probabilities of the value of x and ultimately d . For example, if the level determinedfor x is7or7.5,thmthedeαxlercair)econfiguredtoseete 1. tfthe level is -8 or -7.5, then this can be seen as a high probability that the value is -1. Parity check equations can then be generated from the following:

This will produce a set of parity equations in which, based on the examples above, there will be 6 terms, except in the last one, because there is exactly one 1 in each row of A, B, C, and A The first of these parity equations would then, e.g, look like the following, based on the above examples:

Then, ifSo=+l, then the operation canbe viewed aspassing. If on the other hand, itis-1, then.it can be viewed as a failure. A parity node processor 3702 can be used to implement equation 34, as illustrated by the example embodiment

depicted in HG.46. Message passing algoritiims can be used to allow each suchnode 3702 to make final estimations. EG. 48 is a diagram illustrating and example enibodiment in which a plurality of parity node processors 3702 are configured in accordance with the systems aid methods described herein. Thus, each node 3702 receives infcimation as to what the values Xo ₃ Xi, ...xuate believed to be. Agivmnode3702canthenpκx^flτisiτfora the node believes the output of the other nodes should be and feed this information back in such a manner that the subsequent input to the other nodes is modified. It should be noted, therefore, that in such ai αnbodiment, a node does not produce information to be fedback to its own input related to what it believes its own output should be. This is illustrated in FIG.47 for a single node processor at time = 0. As can be seen, information for each bit is provided to node 3702, which processes the information and produces information related to what it determines each bit should be. These inputs and outputs can be referred to as edges (E). Each output edge is fedback to the relevant input bit The node processors 3702 will, therefore, comprise storage to store the information being fed to it and processed as required. As a result, both storage and routing overheads can become excessive. For example, when information related to bit xø is fed to node SQ ₁ the information fiom each other node related to xo is also added into the information provided to so. This is illustrated by the following: xo +

Again, as mentioned above, in this embodiment, the edge prOducebynodeSoisnottedbacktobitxo.

FIG.49 is a diagram illustrating an example decoder 4000 that can be configured to reduce storage and routing overhead in accordance with one embodiment of the systems and methods described herein The basic premise behind decoder 4000 is that all die edges produced form parity node processor 4002 can be added and then the last edge for each node, produced by that node, can be subtracted out Thus, on die right hand side ofFIG.49, a given row can be updated for all edges and thai shifted in shift registers 4004. The appropriate edge can then be subtracted out for each row using die data provided fora registers 4014, as opposed to doing each row, storing die result and updating it with infonnation fiom odier nodes. It should be noted diat die output of shift registers 4004 can be reananged aid fixed to reduce routing overhead It should also be noted that das process provides ai approximation of the correct data; however, the results converge and ultimately provide the same answer. Qi die left hand side of decoder 4000, each shift register 4008 gets information fiorn only two nodes 3702, e.g, viaregisters 4010 and 4012.

One feature of the present invention is that it may be used to increase the baidwidth of wireless networks or networks diat employ wed media The present invention can be used to transmit ultra-widebaτd signals across any type of wired media Fa" example, die wired media can include optical fiber- ribbon, fiber optic cable, single mode fiber optic cable, multi-mode fiber optic cable, plenum wire, PVC wire, aid coaxial cable, h addition, the wired media can include twisted- pairwiring, whether shielded or unshielded. Twisted-pairwiιenmycorisistof'parrs"ofoolor<odedwires. Common sizes of twisted-pair wire ais 2 pair, 3 pair; 4 pair; 25 pair, 50 pair and 100 pair. Twisted-pair wire is commonly used for telephone aid computer' networks. It comes in ratings ranging from category 1 to category 7. Twisted-pair wiring also is available unshielded. That is, die wiring does not have a foil or odier type of wrapping around the group of conductors widώi die

jacket This type of wiring is most commonly used for wiring for voice and data networks. The foregoing list of wired mediaismeanttobeexemplaryjandnotexclusive.

As described above, the present invention can provide additional baidwidth to enable the transmission of large amounts of data over an existingwiredm television provider, or a computer network located in a business or iiniversity. The additional bandwidth can allow consumers to receive the high speed Internet access, interactive video andother features that are bandwidth intensive.

The present invention may be employed in any type of network, be it wireless, wire, or a mix of wire media and wireless components. That is, a network may use both wire media, such as coaxial cable, and wireless devices, such as satellites, or cellular" antennas. As definedherein, anetworkis agroup ofpoints or nodes connected by commmiicationpaths. The communication paths may use wires or they maybe wireless. Anetwork as defined herein can interconnect with other networks and contain sub-networks. A network as defined herein can be charactaized in terms of a spatial distance, for example, such as a local areanetwork (LAN), apersonal aea network (PAN) ₃ ametrøpolitan areanetwork (MAN), a wide area network (WAN), and a wireless personal area network (WPAN), among others. A network as defined herein can also be charBCterized by the type of data transmission technology used by the network, such as, for example, a Transmission Control Protocol/Internet Protocol (TCP/IP) network, a Systems Network Architecture network, among others. A network as defined herein can also be characterized by whether- it caries voice, data, or both lands of signals. A network as defined herein may also be charactaized byusers of the network, &ich as, for example, users ofa public switched telφtone network (PSTN) or other type of public network, and private networks (such as within a single room or home), anong others. A network as defined herein can also be characterized by ύie usual nature of its connections, for example, a dial-up network, a switched network, a dedicated network, and a non-switched network, among others. A network as defined herein can also be characterized by the types of physical links that it employs, for" example, optical fiber, coaxial cable, a mix of both, unshielded twisted pair, and shielded twisted pair, among others. The present invention may be employed in any type of wireless network, such as a wireless PAN, LAN, MAN, or WAN. In addition, the present invention may be employed in wire media, as the present invention damalically increases the bandwidth of conventional networks that employ wire media, yet it can be inexpensively deployed without exterisivemodfficationtotiie existing wir^medianetworlc

Qie feature of the present invention is that it has a data rate and quality of service high enough to support multiple video streams. Forexa:nple _> oneαribodimentof thepresentm^^ rate of 1.3 gigabits per second. This high data rate is particularly useful in hand held security devices. Such systems can provide dramatically improved national security. For example, current airport security systems involve lage, stationary equipment that scans luggage and passengers. However, an individual may pass through a security checkpoint without being scanned or checked for identification. At most cαnmercial airports it may be exceedingly difficult to locate the individual using current methods and equipment In most cases the security personnel are relying on a verbal description of the individual, which may be inaccurate. Under current regulatory guidelines the terminal must be closed, emptied of passengers ardnianually searched

With the data rates provided by the present invention, security camera access points throughout the airport may transmit one or more channels of streaming video directly to video viewers canied by secuiity personnel, thereby allowing the search to be conducted in a more efficient manner. The data rates of conventional wireless communication systems cannot supportmultiple video steams, and therefore cairøtpiOvide the features and functionality ofthe present inventioa

Thus, it is seen that systems and methods of providing a high speed transmitter and receiver ae provided. One skilled in the at will appreciate that the present invention can be practiced by other than the above-described embodiments, which are presented in this description for purposes of illustration aid not of limitation The specification and drawings are not intended to limit flie exclusionaiy scope of this patent document It is noted that various equivalents for the paticula- embodiments discussed in this description may practice the invention as welL That is, while the present invention lias been described in conjunction with specific embodiments, it is evident that many alternatives, modifications, permutations and variations will become apparent to those of oidinary skill in Hie at in light of the foregoing description. Accoidingly, it is intended that Hie present invention embrace all such alternatives, modifications and variations as fall within the scope of the appended claims. The fact that aproduct, process or method exhibits differences from one or more of the above-described exemplary embodiments does not mean that the product or process is outside the scope (literal scope aid/or other legally- re∞gnized scope) of the following claims.