APPARATUS

Field of the Invention

The present invention relates generally to the field of amplifiers and particularly to dual mode power amplifiers.

Background, of the Invention

The rapid expansion of the number of cellular radio¬ telephones coupled with the desire to provide additional ser¬ vices has prompted the use of an improved transmission technique, time division multiple access (TDMA). TDMA in- creases system capacity over the current analog system through the use of digital modulation and speech coding tech¬ niques. A TDMA transmission is comprised of many time slots.

A linear modulation technique, π/4 differential quadra- ture phase shift keying (π/4 DQPSK), is used to transmit the digital information over the channel. The use of linear modu¬ lation in the U.S. Digital Cellular system provides spectral ef¬ ficiency allowing the use of 48.6 kbps channel data rates, π/4 DQPSK transmits the data information by encoding consecu- tive pairs of bits, commonly known as symbols, into one of four phase angles (±π/4, ±3 π/4) based upon gray encoding. These angles are then differentially encoded to produce an 8 point constellation.

Transmitters designed for use in the U.S. Digital Cellular system are required to operate in both the analog and digital modes. The digital mode uses the π/4 shift DQPSK modulation, and can be implemented using a linear transmit¬ ter. The analog mode uses conventional frequency modula¬ tion and allows the use of higher efficiency non-linear trans- mitters.

Conventional linear amplifiers are inherently less effi¬ cient than their constant envelope counterparts due to the types of signals they must amplify. A constant envelope am¬ plifier is required to put out a signal at only one power level over time. It can therefore be optimized for peak efficiency at that power level. This optimization entails placing a load impedance on the device such that, at the designed power out, the AC collector voltage magnitude is close to or even exceeds the DC supply voltage. In this condition, the amplifier is close to or actually in saturation and has optimum efficiency. The linear amplifier must amplify signals at power levels that vary over time, with whatever amplitude modula¬ tion that has been impressed upon the input signal. No satu¬ ration is allowed in the linear amplifier, or there will be se- vere distortion of the envelope. This distortion causes loss of amplitude information and spreading of the transmit spec¬ trum into adjacent channels. The amplifier circuit must op¬ erate such that at peak power out, the amplifier is not in satu¬ ration. While it is possible to optimize for good efficiency at peak power out, the efficiency falls off rapidly as power out falls.

This creates a problem for the U.S. Digital Cellular ra¬ dio which is intended to operate in both linear and constant envelope modes. When compared to current analog radios with constant envelope amplifiers, efficiency will be much lower in the digital radio. In the linear mode, efficiency is op¬ timized for peak power out, but the signal spends only a short time there. Average efficiency will be lower than peak power efficiency because of this. For digital cellular this is not a se- vere limitation since a TDMA system is used and the trans¬ mitter is only on 1/3 of the time (only every third time slot is used by the radio). Even if the average efficiency is poor in this mode, transmit current is not significantly worse (and may be better) than a conventional analog radio. The problem arises when this same radio is used for an analog call with a constant envelope signal. Now the trans-

itter is 'on for 100% of the conversation time and, since it op¬ erates at an average power 3.2 dB below the optimum power out, efficiency is poor. Data on test circuits shows drops in the 12 percentage point range. This translates into a substantial increase in transmit current. The increased current re¬ quirements will substantially decrease the time that a battery powered radiotelephone will be useful. There is a resulting need for an amplifier circuit that operates efficiently in both linear and constant envelope modes.

Summary of the Invention

The present invention encompasses a dual mode power amplifier apparatus that is comprised of at least one amplify- ing means. The at least one amplifying means has a collector supply input, a signal input, and an output. Each amplifying means generates a signal at the output that has amplified characteristics of ah input signal coupled to the signal input. The dual mode power amplifier apparatus is urther comprised of voltage conversion means having a mode select input, a power input, and an output. The voltage conversion means generates a first voltage at the output in response to a first control signal and a second voltage at the output in re¬ sponse to a second control signal. The first and second control signals are coupled to the mode select input of the amplifying means. The voltage conversion means output is coupled to the collector supply input of the amplifying means.

Brief Description of the Drawings

FIG. 1 shows the preferred embodiment of the dual mode power amplifier apparatus of the present invention.

FIG. 2 shows another embodiment of the dual mode power amplifier apparatus of the present invention. FIG. 3 shows a flowchart of the method of the present invention.

FIG. 4 shows yet another embodiment of the dual mode power amplifier apparatus of the present invention.

FIG. 5 shows a block diagram of a typical radiotele¬ phone in accordance with the present invention.

Detailed Description of the Preferred Embodiment

The dual mode power amplifier apparatus operates lin¬ early in the linear mode for use in a digital cellular system as well as the constant envelope mode for use in the present ana¬ log cellular system. This efficient operation is accomplished without RF switching.

The preferred embodiment of the present invention is il¬ lustrated in FIG. 1. The apparatus is comprised of a final power amplifier (101) and two power amplifier drivers (102 -

103) connected to the RF input signal (502). The power ampli¬ fiers (101 - 103) are connected in series with the final power amplifier (101) outputting the amplified RF signal. The power amplifiers (101 - 103) used in the preferred embodiment are MHW927A manufactured by Motorola, Inc.

A voltage convertor (104) is connected to the collector supply of each power amplifier stage. The voltage convertor (104) must be an efficient supply that, in the preferred embod¬ iment, is a switching power supply. The voltage convertor (104) is connected to a voltage V _cc that is 12.5 V in the pre¬ ferred embodiment. The voltage convertor (104) is capable of generating two voltages from V _cc . Which voltage is generated is chosen by the analog/ digital mode select line (105) con¬ nected to the convertor (104). A logical high on this line (105) selects the analog mode and a logical low selects the digital mode. In the preferred embodiment, a logical high is +5.0 V and a logical low is 0 V. The analog mode is also referred to in the art as the constant envelope mode and the digital mode is referred to as the linear mode. When the digital mode is selected, the voltage convertor

(104) supplies a voltage Vi that is close to V^; 12.5 V in the

preferred embodiment. This is the design voltage of the power amplifiers (101 - 103) and allows the amplifiers (101 - 103) to provide its designed value of linearity at its rated power out. Providing 12.5 V to the power amplifiers (101 - 103) is accom- plished by turning off the voltage convertor (104) and turning on an internal pass device that allows the full V _cc to reach the amplifiers with minimal loss. The pass device is transistor, so V _cc minus the voltage drop across the transistor equals the voltage to the power amplifiers. If the analog mode is selected, the voltage convertor

(104) is on and generates a predetermined voltage N ₂ to the power amplifiers (101 - 103) that is significantly less than V _cc . In the preferred embodiment, this voltage is 8.65 V. With this reduced supply voltage, the AC collector voltage magnitude at the reduced power out will again be close to the DC supply voltage as it was at the peak linear power out with the higher supply voltage of V _cc . Efficiency in the analog mode, there¬ fore, is still optimum.

In an ideal situation, the relationship between the Ni and N ₂ is the same as the relationship between the peak linear mode power and the constant envelope mode power. For U.S. Digital Cellular, the difference is 3.2 dB, thus making N ₂ 3.2 dB less than Ni. For optimum performance in a real situa¬ tion, however, the difference between Vi and N ₂ varies from the ideal due to saturation and other non-ideal effects.

To illustrate the use of the dual mode power amplifier apparatus of the present invention, it is first assumed that the hybrid digital/analog radiotelephone is operating in the U.S. Digital Cellular system. The mode selection signal (105) is brought low to select the digital mode. This turns off the switching power supply (104), thus allowing the 12.5 V V _cc to be applied to the collector supply of the power amplifiers (101 - 103). The higher collector voltage allows the power amplifiers (101 - 103) to operate linearly at their designed value of linear- ity at their rated output power.

When the radiotelephone is operating in the analog cel¬ lular system, the mode selection signal (105) is high to select the analog mode of the amplifier apparatus. The switching power supply (104) is on, generating a voltage of 8.65 V that is applied to the collectors of the power amplifiers (101 - 103). Since the transmitted signal is a constant envelope signal at a reduced power out, the reduced collector voltage allows the power amplifiers (101 - 103) to operate efficiently at an AC col¬ lector voltage closer to the supply voltage. The above described method of the present invention is illustrated in the flowchart of FIG. 3.

Another embodiment of the present invention is illus¬ trated in FIG. 2. This embodiment is comprised of a four stage power amplifier module, such as a Motorola MHW927A module. The bias for the last two stages (203) and both bias and supply for the first two stages (204) are tied together. The collectors of the last two stages are also tied together. These collectors are connected to a voltage convertor (204) that gen¬ erates two voltages. This voltage convertor (204) the same kind of voltage convertor (104) as discussed in the preferred embod¬ iment. The signal to be amplified is coupled to the RF input (502) and the RF output (202) is the amplified signal. A bias select circuit (401), illustrated in greater detail in FIG. 4, gen¬ erates the bias voltage Vbb- The V _bb selection is performed by the analog/digital se¬ lect line (516). By bringing this line low to select the digital mode, the transistor (412) is off making the only voltage at the bias of the final stage (420) of the power amplifier the 0.70 V drop across the diode (411). The power amplifier is now in the linear mode. When the analog/digital select line (516) is high, the transistor (412) is in saturation and the voltage at the final stage (420) of the power amplifier is the 0.20 V saturation volt¬ age of the transistor (412). A resistor (410) in the collector leg of the transistor (412) is chosen very small, 0.1 Ω in the pre- ferred embodiment, for current limiting. The power amplifier drivers (421 - 422) always have Vbb as the bias voltage.

The embodiment of FIG. 2 will be used to illustrate the efficiency provided by the dual mode power amplifier. The bias connection is held at 9.5 V. The collector voltage of the last two stages was varied between the rated supply voltage of 12.5 V and a value 3.2 dB less, 8.65 V. The results of this oper¬ ation are illustrated in the following table:

The system efficiency was computed assuming .2 Ω re- sistance in a FET pass device for the digital mode and an 85% efficient voltage convertor for the analog mode. The efficiency calculation for the power amplifiers is determined by the equation:

^ηpβ - v _b *fc+v«*ι« where P ₀ is the power out.

The system efficiency is computed by adding the dissi¬ pation of the voltage convertor to the denominator of the ^P* expression. The above table shows that even with the some¬ what higher convertor losses, the concept still improves effi¬ ciency by 4% over the digital mode.

FIG. 5 illustrates a block diagram of a typical radio transmitter of the present invention. The actual circuitry em¬ bodying the functional blocks of the diagram may be mounted on one or more circuit boards and housed within a conven¬ tional radiotelephone housing. The radio transmitter re¬ quires only the high efficiency dual mode power amplifier ap¬ paratus (104) of the present invention by providing means to operate the amplifier in either a linear mode or a non-linear

mode. Efficient amplification of a frequency modulated sig¬ nal, and amplification of a π/4 DQPSK modulated signal is thereby possible, while, at the same time, minimizing circuit size and cost. The microphone (54) converts voice signals into an elec¬ trical, information signal (56 and 58). The information signal supplied on line (56) is utilized when, similar to conventional cellular, radiotelephone communications, a frequency modu¬ lated information signal is to be generated by the radiotele- phone. The information signal supplied on line (58) is utilized when a discrete, encoded signal modulated to form a compos¬ ite modulated information signal is to be generated by the ra¬ diotelephone.

The information signal generated on line (56) is sup- plied to a voltage controlled oscillator (60) where the informa¬ tion signal is combined with an oscillating signal of a certain frequency. A frequency modulated information signal (62) is generated by the voltage controlled oscillator (60) to a modula¬ tor (64). When the radiotelephone is to transmit a frequency modulated information signal, the modulator (64) does not al¬ ter the frequency modulated information signal (62), but rather "passes-through" the frequency modulated information signal. The oscillator (60) and modulator (64) may together comprise a hybrid modulation apparatus (68). The information signal (58) is supplied to a vocoder (72) where the analog information signal is digitized and encoded according to an encoding scheme, and generates a discrete, encoded signal (76) that is supplied to the modulator (64). The modulator (64) modulates the discrete, encoded signal (76) to form a composite, modulated information signal of a pre-de- termined frequency.

The modulated information signal modulated accord¬ ing to either a frequency modulation technique or a composite modulation technique, is supplied to a mixer (80). The mixer (80) mixes this signal with an offset transmission-frequency carrier wave generated by a synthesizer (90) and supplied to

the mixer (80). The mixer (80) mixes the modulated informa¬ tion signal with the carrier wave (92). The mixer (80) then generates a modulated information signal (96) upon a carrier wave of a carrier frequency determined by the oscillating fre- quency of synthesizers (90 and 60).

The modulated information signal (96) is coupled to a filter (540) that forms a passband of frequencies centered about, or close to, the carrier frequency of the transmitted wave. This filter (540) generates a filtered signal (502), con- taining the modulated information signal, that is coupled to the power amplifier (500) of the present invention. The power amplifier (500) amplifies the modulated information signal to power levels adequate for transmission through an antenna (506). Prior to transmission of the amplified signal, and as il- lustrated, the amplified signal may be filtered by another filter

(507) that may, for example, form a portion of a duplexer. This filter (507) is positioned in-line between the power ampli¬ fier (500) and the antenna (506).

A processor (508) provides control signals (510, 512, 514, and 515) to control operation of the oscillator (60), the vocoder

(72), the modulator (64), and the synthesizer (90), respectively, to control modulation of the information signal generated by the microphone (54).

The processor (508) controls whether the information signal generated by the microphone (54) is modulated by the oscillator (60) to form a frequency modulated information sig¬ nal, or alternately, is encoded by the vocoder (72) according to ^~ a discrete encoding scheme and modulated by the modulator (64) to form a composite modulated information signal. As the modulated signals supplied to the amplifier

(500), in the preferred embodiment, are frequency modulated or composite modulated information signals, the processor

(508) supplies analog/digital select signal (516) to the amplifier (500) to cause operation of the amplifier (504) in either the lin- ear mode or the non-linear mode. The linear mode is enabled when a π/4 DQPSK modulated signal is to be transmitted and

the non-linear mode is enabled when an FM signal is to be transmitted.

FIG. 5 further illustrates the radiotelephone receive circuit for a signal transmitted to the antenna (506). The sig- nal transmitted to the antenna (506) is supplied to the filter (517) which passes signals of desired frequencies to the mixer (518). The mixer (518) receives an oscillating signal (519) from the synthesizer (90) and generates a mixed signal that is sup¬ plied to a demodulator (520). The demodulator (520) supplies a demodulated, electrical information signal to a speaker (521). The processor (508) may supply a signal (520A) to the demodu¬ lator (120) to control its operation. The speaker (521) converts the electrical information signals into audible signals.