FIELD

[0001] Aspects of the present invention generally relate to voltage regulation and, more particularly, to a parallel combination of a low dropout voltage regulator with a switching-mode power supply to regulate voltage.

BACKGROUND

[0002] Many functions of an advanced electronic device, such as a mobile phone, are implemented in a system-on-a-chip (SoC) integrated circuit. The SoC consumes current that changes with the number and kind of operations it performs. Descriptions of an element in terms of current or power are interchangeable after scaling by a respective voltage. The operations performed can change rapidly, for example, a few nanoseconds. The change in current consumption can be large, for example, a few amps. This results in a large current time derivative (dl/dt) that can interfere with operation of the SoC.

[0003] A power distribution network supplies power, for example, as a voltage supply, to the SoC. The SoC may be packaged in an integrated-circuit package that may be mounted on an interconnection substrate, such as a printed circuit board, for connection with other components including, for example, a power supply and battery. The power distribution network includes connections through the printed circuit board and integrated-circuit package. The connections of the power distribution network can have substantial parasitic inductance. This inductance combined with the large current time derivatives can cause large spike-like dips in the supply voltage, also referred to as droop, in the voltage supplied to the SoC. The droop can be so large as to interfere with proper operation of the device.

[0004] The voltage level supplied to the SoC is generally increased (which may be referred to as guardbanding) by the amount of voltage droop so the "drooped" voltage is sufficient for proper operation of the SoC. Guardbanding the voltage level increases power consumption and is undesirable, for example, due to increased temperature and decreased battery duration. Some prior systems have attempted to reduce the voltage droop, for example, by reducing inductance in the power distribution network or adding decoupling capacitors on or close to the SoC. For example, external landside capacitors (LSCs) and embedded-passive- substrate (EPS) capacitors may be added during routing of the PDN. Added decoupling capacitors may only slightly reduce the voltage droop. Additionally, they can be size and cost prohibitive.

SUMMARY

[0005] In one aspect, a hybrid parallel power supply is provided that includes: a first power supply connected to a power rail and configured to supply current to a load device via the power rail, the first power supply being further configured to regulate the power rail to a first target voltage level; and a voltage regulator connected to the power rail and configured to supply current to the load device via the power rail, the voltage regulator being further configured to regulate the power rail to a second target voltage level.

[0006] In one aspect, a method is provided for of supplying power to a load device. The method includes: supplying current from a power supply to the load device via a power rail at a first target voltage level; and supplying current from a voltage regulator to the load device via the power rail at a second target voltage level.

[0007] In one aspect, a hybrid parallel power supply is provided that includes: a first means for supplying power connected to a power rail and configured to supply current at a first target voltage level to a load device via the power rail; and a means for regulating voltage connected to the power rail and configured to supply current at a second target voltage level to the load device via the power rail.

[0008] Other features and advantages of the present invention should be apparent from the following description which illustrates, by way of example, aspects of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

[0009] The details of the present invention, both as to its structure and operation, may be gleaned in part by study of the accompanying drawings, in which like reference numerals refer to like parts, and in which:

FIG. 1 is a schematic diagram of a model of a power distribution network;

FIG. 2 is a graph of a droop characteristic of a power distribution network in the time domain; FIG. 3 is a graph of impedance of a power distribution network in the frequency domain;

FIG. 4 is a graph of frequency responses of a switching-mode power supply and a low dropout voltage regulator;

FIG. 5 is a functional block diagram of an electronic system with a hybrid parallel power supply according to a presently disclosed embodiment;

FIG. 6 is a functional block diagram of an electronic system with a hybrid parallel power supply according to a presently disclosed embodiment;

FIG. 7 is a functional block diagram of an electronic system with a hybrid parallel power supply according to a presently disclosed embodiment;

FIG. 8 is a functional block diagram of an electronic system with a hybrid parallel power supply according to a presently disclosed embodiment;

FIG. 9 is a time-domain graph of a droop characteristic of a hybrid parallel power supply according to a presently disclosed embodiment; and

FIG. 10 is a flowchart of a process for supplying power to an electronic device according to a presently disclose embodiment.

DETAILED DESCRIPTION

[0010] The detailed description set forth below, in connection with the accompanying drawings, is intended as a description of various configurations and is not intended to represent the only configurations in which the concepts described herein may be practiced. The detailed description includes specific details for the purpose of providing a thorough understanding of the various concepts. However, it will be apparent to those skilled in the art that these concepts may be practiced without these specific details. In some instances, well-known structures and components are shown in simplified form in order to avoid obscuring such concepts.

[0011] FIG. 1 is a schematic diagram of a model of a power distribution network (PDN). The PDN may be segregated into interconnect domains corresponding to a die domain 105, a package domain 110, a power-supply domain 112, and a circuit-board domain 115. Each of the PDN interconnect domains includes respective combinations of resistances, capacitances, and inductances that form a characteristic impedance for that domain. The resistances, capacitances, and inductances may be referred to as parasitic elements. These parasitic elements may be aggregated into equivalent component values of resistance, capacitance, and inductance that are present between components mounted within the PDN interconnect domains. For example, the circuit- board domain 115 may contribute series inductance due to the traces conducting current from a battery 165 to a power supply 170 in the power-supply domain 112. The components within the PDN interconnect domains may have equivalent values of resistance, capacitance, and inductance that interact electrically with the parasitic elements of the PDN interconnect domains.

[0012] In the PDN model of FIG. 1, the circuit-board domain 115 includes a circuit- board-inter-planar capacitance 125 that may aggregates per unit length of interconnect traces across the interconnect substrate. The interconnect traces also contribute a circuit- board-equivalent- series resistance 120 and a circuit-board-equivalent-series inductance (L-pcb) 130 that aggregate per unit length of the trace.

[0013] The battery-equivalent impedance includes an equivalent-battery-output resistance 175 in series between the battery 165 and the parasitic elements of the circuit- board domain 115. The power supply 170 in the power-supply domain 112 receives power from the battery 165 through the equivalent-battery-output resistance 175, the circuit-board-equivalent- series resistance 120, and the circuit-board-equivalent- series inductance 130. The power supply 170 includes an equivalent-power-supply-output resistance 180. Other impedances of the power supply may be lumped into an equivalent-power-supply output impedance 185 including series inductance, resistance, and capacitance. A power- supply-domain inductance 190 models the inductance encountered in routing traces between the power supply 170 and the package domain 110.

[0014] In a fashion similar to that of the circuit-board domain 115, the package domain 110 includes a package-inter-planar capacitance 135, a package-equivalent- series inductance 140, and a package-equivalent- series resistance 145. Each of these parasitic elements aggregates per unit length that the PDN runs through the package domain 110. The die domain 105 is modeled with an equivalent-load impedance 150 and a switched-equivalent-load impedance 155. The equivalent-load impedance 150 represents portions of the SoC (which from a power distribution network viewpoint may be referred to as the load) that are not switching and/or portions of SoC circuitry that are in a continuous or standby operation. The switched-equivalent-load impedance 155 represents portions of SoC circuitry that are activated or deactivated according to particular operations of the SoC. A switch 160 symbolically represents the switched activity of the switched-equivalent-load impedance 155.

[0015] FIG. 2 is a graph of the droop characteristic of a power distribution network in the time domain. The droop characteristic will be described with reference to the PDN of FIG. 1 but similar effects occur in other systems. The graph shows a supply voltage 215 at the SoC in the die domain 105. The y-axis of the graph is voltage and the x-axis of the graph is time. Initially, the SoC draws a low level of current from the battery 165 via the power supply 170. At time 250, the SoC switches to drawing a high level of current. The low level of current is modeled by the switch 160 in the load model being open and the high level of current is modeled by the switch 160 being closed. Initially, the supply voltage 215 (at the load) is at a nominal voltage level. At time 250, the current drawn by the load increases rapidly. This increase in current is satisfied by additional current from the battery 165 flowing through the circuit-board domain 115, the package domain 110, power-supply domain 112, and the die domain 105. The inductances in this path result in a spike-like dip 205 in the supply voltage 215. The spike-like dip 205 may result in the supply voltage 215 dropping below a minimum level for proper operation of the SoC. For example, in some systems, a droop characteristic 210 may be as much as an 18-30% drop in the supply voltage 215 for a 2.3 amp load-current transient occurring in a 4 ns timeframe.

[0016] FIG. 3 is a graph of impedance of a power distribution network in the frequency domain. The graph will be described with reference to the power distribution network of FIG. 1 ; however, other networks result in similar impedances. The graph shows several impedance peaks that correspond to the respective impedances formed from parasitic elements in the respective PDN interconnect domains. A first impedance peak 305 corresponds to the current limiting impedances in the die domain 105. The first impedance peak 305 may be primarily associated with the frequency dependent impedances seen by the portion of the PDN path on-die. The first impedance peak 305 may have a center frequency of about 100 MHz and may be the largest impedance peak. A second impedance peak 310 corresponds to the current limiting impedances created by the parasitic elements formed as the PDN is routed through the package domain 110. The second impedance peak 310 may have a center frequency of about 2 MHz. A third impedance peak 315 corresponds to the current limiting impedances caused by the parasitic elements in the circuit-board domain 115. The third impedance peak 315 may have a center frequency of about 100-200 kHz. When variations in SoC currents occur at or near the frequencies of the impedance peaks, the effect of voltage droop is increased.

[0017] The system of FIG. 1 includes the power supply 170 between the battery 165 and the load device to provide voltage regulation and avoid droop characteristics. The power supply 170 may be a switching-mode power supply (SMPS) and provide power to the load device with the degree of supply voltage regulation that is available according to the regulating abilities of the SMPS. The SMPS may be implemented within the circuit-board domain 115 on a separate die from the load device and provide power through the circuit board interconnection layers to the SoC die (load device in the die domain 105). The SMPS receives power from the battery 165. The battery 165 connects to the SMPS through the circuit board. The SMPS is coupled to the die through the parasitic elements introduced by the interconnection path through the circuit-board domain 115 and the package domain 110. These interconnection path parasitic elements correspond to the package-inter-planar capacitance 135, the package- equivalent-series inductance 140, the package-equivalent- series resistance 145, and the power- supply-domain inductance 190 described above.

[0018] FIG. 4 is a graph of frequency responses of a switching-mode power supply (SMPS) and a low dropout voltage regulator (LDO). The graph includes a SMPS frequency response 405 and an LDO frequency response 425 plotted versus frequency. The SMPS frequency response 405 and the LDO frequency response 425 are plotted in terms of output power in FIG. 4. Other measures, for example, current, may also be used. The frequency responses are generally flat at low frequencies and then decline at higher frequencies. The SMPS frequency response 405 may be described as having a SMPS corner frequency 410 where the power is attenuated by 3 dB (or a factor of two) and a corresponding SMPS bandwidth 415. Similarly, the LDO frequency response 425 may be described as having a LDO corner frequency 430 and a corresponding LDO bandwidth 435.

[0019] An exemplary SMPS may have a bandwidth of about 2 MHz. Generally, the SMPS bandwidth 415 may occur at about one-fifth of the SMPS clock frequency. This bandwidth is not sufficient for the SMPS to supply rapidly changing power requirements that can occur in an SoC. The SMPS is, however, suited for supplying relatively large amounts of current with a high efficiency. For example, the SMPS may be able to supply 3.0 A with an efficiency of about 85%.

[0020] An exemplary LDO may have a bandwidth of about 600 MHz. This bandwidth is much better suited than the SMPS bandwidth for supplying rapidly changing power requirements that can occur in an SoC. The LDO is, however, less efficient than the SMPS. For example, an LDO that receives a 1.2 V input supply voltage and produces a 0.9 V output will have an efficiency less than 75%.

[0021] FIG. 5 is a functional block diagram of an electronic system including a hybrid parallel power supply according to a presently disclosed embodiment. The system uses a hybrid parallel power supply to provide power to a load device 560. The hybrid parallel power supply connects a power supply (e.g., a switching-mode power supply) in parallel with a voltage regulator (e.g., a low dropout voltage regulator). The hybrid parallel power supply combines advantageous features of both switching-mode power supplies and low dropout voltage regulators. The hybrid parallel power supply is power efficient and can supply a stable voltage to load device, such as a smartphone SoC, whose current demand rapidly changes.

[0022] The hybrid parallel power supply includes a first power supply ("SMPS1") 505 connected in parallel with a voltage regulator ("LDO") 510. The first power supply 505 supplies power to a power rail 520. The first power supply 505 works to regulate the power rail 520 to a first target voltage level, for example, 0.9 V. The first target voltage level may be configurable, for example, using a control register. The first power supply 505 receives power from a first supply terminal 545 (e.g., connected to the battery 165 in the system of FIG. 1 via connections on a circuit board 515). The first power supply 505 is able to supply power to the power rail 520 with high efficiency. The first power supply 505 may be a switching-mode power supply (SMPS).

[0023] The voltage regulator 510 also supplies power to the power rail 520. The voltage regulator 510 works to regulate the power rail 520 to a second target voltage level. The second target voltage level may be configurable, for example, using a control register. The voltage regulator 510 has an input connected to a second supply terminal 540 from which the voltage regulator 510 receives power. The voltage regulator 510 is able to provide current with a high-frequency-response characteristic to the power rail 520. The voltage regulator 510 may be a low dropout voltage regulator (LDO). An LDO can operate with low headroom (difference between the input voltage and output voltage), for example, regulating a 1.2 V input to produce a 0.9 V output.

[0024] The voltage regulator 510, in an embodiment, operates with a Class-B current mode output. Such circuits can have high bandwidths, for example, 600 MHz. The Class-B current mode output of the voltage regulator 510, in contrast with a push-pull output, sources current to the power rail 520 (e.g., to the load device 560) when the voltage on the power rail 520 is below the second target voltage level but does not sink current from the power rail 520 when the voltage on the power rail 520 is above the second target voltage level. This facilitates connecting the voltage regulator 510 and the first power supply 505 in parallel to the power rail 520.

[0025] The first power supply 505 and the voltage regulator 510 operate to regulate the voltage level of the power rail 520. This voltage regulation may be understood as the monitoring of the voltage level of the power rail 520 and when the voltage level differs from the respective target voltage level, changing operation of circuits driving the power rail 520 so that the voltage level moves toward the target voltage level. The voltage level of the power rail 520 may vary due, for example, to inexact target voltage levels in the first power supply 505 and the voltage regulator 510, output impedances of the first power supply 505 and the voltage regulator 510, and response times of the first power supply 505 and the voltage regulator 510 to changes in current demand of the load device 560.

[0026] The second target voltage level, to which the voltage regulator 510 regulates the power rail 520, may be different from the first target voltage level, to which the first power supply 505 regulates the power rail 520. The second target voltage level may be, for example, an offset voltage less than the first target voltage level. For example, in a hybrid parallel power supply where the first target voltage level is 0.9 V, the second target voltage level may be 088 V. In this arrangement, the first power supply 505 may supply most of the current dissipated by the load device 560 with the voltage regulator 510 rapidly supplying current in response to any dips in the voltage on the power rail 520.

[0027] The hybrid parallel power supply of FIG. 5 includes a second power supply ("SMPS2") 530 to supply power to the voltage regulator 510 via second supply terminal 540. The second power supply 530 works to drive the first supply terminal 545 to a third target voltage level. The third target voltage level may be chosen, for example, to allow efficient operation of the voltage regulator 510. The third target voltage level may also be a level used by other components in the system. The third target voltage level may be, for example, 1.2 V when the nominal level on the power rail 520 is 0.9 V. The second power supply 530 is able to supply power to the voltage regulator 510 with high efficiency. The second power supply 530 receives power from the first supply terminal 545. The second power supply 530 may be a switching-mode power supply.

[0028] The voltage regulator 510 and the load device 560 may be fabricated on a first die ("DIE 1 ") 525. Since the voltage regulator 510 and the load device 560 are located together, parasitic impedances between the voltage regulator 510 and the load device 560 are small. Portions of the power rail 520 may also be fabricated on the first die 525. Board and package parasitic impedances do not impair the connection between the voltage regulator 510 and the load device 560.

[0029] In the embodiment of FIG. 5, the first die 525 includes an on-die capacitor 550 connected to the second supply terminal 540, which supplies power to the voltage regulator 510. The on-die capacitor 550 can supply current to the voltage regulator 510 with a high-frequency-response characteristic. The on-die capacitor 550 supports the voltage regulator 510 supplying current to the load device 560 with a high-frequency- response characteristic and aids in reducing a droop characteristic that would otherwise occur on the power rail 520 when the current demand of the load device 560 rapidly changes. The on-die capacitor 550 stores more charge than a capacitor on the power rail 520 due to the higher voltage of second supply terminal 540 compared to the power rail 520. As a result, the parallel combination of the voltage regulator 510 and the first power supply 505 may operate without (or with little) further capacitance, such as a bulk capacitor, an external capacitor, a landside capacitor, or an embedded-passive- substrate (EPS) capacitor. According to certain exemplary embodiments, the on-die capacitor 550 may be about 220 nF.

[0030] The on-die capacitor 550 may alternatively be fabricated external to the first die 525, for example, on a circuit board or integrated circuit package, or a combination of on-die and external capacitors. Provisioning of capacitance on die, on package, or on circuit board may be determined according to the relative costs and performance of the respective implementations. These costs may include the costs of capacitors, interconnection, and package pins. Performance of the power distribution network generally improves with capacitance closer to the load device.

[0031] The first power supply 505 and the second power supply 530 may be fabricated on a second die ("DIE 2") 535. Combining the voltage regulator 510 and the load device 560 on the first die 525 and combining the first power supply 505 and the second power supply 530 on the second die may allow the various components to be manufactured using fabrication processes that are selected for the particular requirements of the components. For example, the voltage regulator 510 and the load device 560 may be manufactured using a high-density complementary metal-oxide- semiconductor (CMOS) process that allows many functions to be provided by the load device 560 and the first power supply 505 and the second power supply 530 may be manufactured using a high-power process that allows high efficiency power supplies. The first die 525 and the second die 535 may be mounted (directly or using integrated circuit packages) on an interconnection substrate, such as the circuit board 515. In an embodiment, the second die 535 may be a power-management integrated circuit (PMIC).

[0032] A system using a hybrid parallel power supply may be more power efficient than systems using an SMPS, an LDO, or a series SMPS-LDO combination. The efficiency of the hybrid parallel power supply for an example implementation is about 76%. The hybrid parallel power supply can also lower system power by allowing a smaller voltage guardband. This is particularly valuable in systems (such as a CMOS SoC) where the power is proportional to the voltage squared. An SoC with the voltage regulator 510 and the load device 560 may also have a reduced number of pins due, for example, to direct connection of the first power supply 505 and the voltage regulator 510 to the load device 560 and reduced use of decoupling capacitors on the power rail 520.

[0033] The parallel combination of the LDO and SMPS combines the characteristics of the individual circuits to efficiently supply power while reducing voltage droop caused by rapid load current changes. The SMPS can be viewed as generally involved in supplying the steady-state current needs of the load device. In this way, the relatively large current demand of the load device is provided with the high efficiency of the SMPS. The LDO can be viewed as generally involved in supplying current to the load device in response in changes in the load current that could otherwise cause large voltage droops. That is, the LDO rapidly reacts to drops in the supply voltage and supplies current to the load device until the SMPS can react. The high bandwidth of the LDO enables the hybrid parallel power supply to provide sufficient current in a timely manner such that the droop characteristics are greatly reduced. Additionally, a parallel combination of the LDO and SMPS may allow use of a simplified SMPS, for example, an SMPS with fewer phases.

[0034] FIG. 6 is a functional block diagram of an electronic system with a hybrid parallel power supply according to a presently disclosed embodiment. The hybrid parallel power supply of FIG. 6 generally corresponds to the hybrid parallel power supply in the system of FIG. 5. According, the description of the system of FIG. 6 may omit details common to the system of FIG. 5. The system of FIG. 6 includes the first power supply 505 and the voltage regulator 510 (receiving power from the second power supply 530) supplying power to the load device 560 via power rail 520. The load device 560 (which may be an SoC including a processor and other circuits) is modeled with an equivalent load impedance 675a and a switched equivalent load impedance 675b.

[0035] The voltage regulator 510 includes an operational amplifier 610, a current mirror 615, a reference converter ("DAC") 620, and a bandgap source 625. The reference converter 620 and the bandgap source 625 combine to produce a reference voltage that sets the second target voltage level at which the voltage regulator 510 supplies power to the power rail 520. The bandgap source 625 produces a reference output voltage that is nearly constant (e.g., less than 1% variation with process, supply voltage, and temperature). The bandgap source 625, in an embodiment, produces the reference output voltage at a sub-bandgap level (e.g., 0.64 V). The reference converter 620 scales the reference output voltage from the bandgap source 625 to produce the reference voltage. For example, the reference converter 620 may scale a 0.64 V reference output voltage from the bandgap source 625 by 11/8 to produce a 0.88 V reference voltage. The reference converter 620 may be a digital-to-analog converter (DAC). The reference converter 620 may receive a digital input to configure the reference voltage. The digital input may be used to adjust the second target voltage level.

[0036] The operational amplifier 610 and the current mirror 615 operate as an LDO. The current mirror 615 is coupled to the output of the operational amplifier 610 and can provide current from the second power supply 530 to the power rail 520. The current mirror 615 includes a sense device 630 and a drive device 635. In the embodiment of FIG. 6, the sense device 630 and the drive device 635 are p-channel transistors. The source of the sense device 630 and the source of the drive device 635 connect to the output of the second power supply 530. The gate and the drain of the sense device 630 and the gate of the drive device 635 connect to the output of the operational amplifier. The drain of the drive device 635 connects to the power rail 520. The sense device 630 sources a current to the operational amplifier 610 and produces a corresponding gate voltage for the drive device 635. The current mirror 615 allows for scaling of currents between the drive device 635 and the sense device 630. The drive device 635 sources current to the power rail 520 that is scaled (e.g., by scaling of device sizes between the drive device 635 and the sense device 630) from the current sunk by the operational amplifier 610. For example, the size (e.g., transistor width) of the drive device 635 may be 200-800 times the size of the sense device 630. The drive device 635 may be, for example, fabricated from multiple devices arranged in parallel to produce the scaled drive strength.

[0037] The operational amplifier 610 has its non-inverting input ("+") connected to the reference voltage from the reference converter 620 and its inverting input ("-") connected to the power rail 520. The operational amplifier 610 may be a Class-B (push- pull) operational transconductance amplifier (OTA). The output of the operational transconductance amplifier sources or sinks current based on the voltage difference between the inverting and non-inverting inputs. The feedback loop from the output of the operational amplifier 610 through the current mirror 615 to the power rail 520 back to the inverting input of the operational amplifier 610 causes the voltage regulator 510 to regulate its output to the level (second target voltage level) of the reference voltage.

[0038] The voltage regulator 510 may be considered to operate as a comparator that turns on to source current to the power rail 520 when the voltage on the non-inverting input is greater than the voltage on the inverting input and turns off when the voltage on the non-inverting input is less than the voltage on the inverting input. Viewed thusly, the operational amplifier 610 compares the level of the reference voltage to the level of the power rail 520 with the level of the reference voltage serving as triggering level to turn the current mirror 615 on. An operational transconductance amplifier can have high bandwidth so that the voltage regulator 510 can provide current to the power rail 520 with a high-frequency-response characteristic. Additionally, a Class-B OTA may be fabricated using standard logic transistors and without special devices or device fabrication techniques. The voltage regulator 510 may also have low quiescent current and thus contribute to overall power reduction for the system. The LDO may also be implemented within a small amount of die area

[0039] The system of FIG. 6 includes the on-die capacitor 550 and a second capacitor 650 located in a package 605 housing the SoC and connected to the input of the voltage regulator 510. The values of the second capacitor 650 and the on-die capacitor 550 may be selected based on, for example, cost, performance, and size. In an example embodiment, the capacitance of the second capacitor 650 may be about 200 times greater than the capacitance of the on-die capacitor 550. For example, the capacitance of the second capacitor 650 may be 200 nF and the capacitance of the on-die capacitor 550 may be 1 nF. In some embodiments, a further capacitor may be located on the circuit board 515 and connected to the input of the voltage regulator 510.

[0040] In some embodiments, a rail capacitor 680 located in the first die 525 is connected to the power rail 520. The rail capacitor 680 may work in combination with the on-die capacitor 550 to provide current to the load device 560 during a droop event. Other combinations of capacitors on the circuit board 515, the package 605, the first die 525, or other locations may also be used.

[0041] FIG. 6 also shows interconnection-parasitic elements 665a,b that represent the reactive elements encountered in making various electrical connections between the first power supply 505 and the power rail 520 and the second power supply 530 and the voltage regulator 510. The interconnection-parasitic elements 665a,b are shown in the domain of the circuit board 515 between the first die 525 and the second die 535. The interconnection-parasitic elements 665a,b may be combinations of resistors, capacitors, and inductors that represent an equivalent impedance encountered by electrical connections spanning between the first die 525 and the second die 535.

[0042] FIG. 7 is a functional block diagram of an electronic system with a hybrid parallel power supply according to a presently disclosed embodiment. The hybrid parallel power supply of FIG. 7 is similar to the hybrid parallel power supply in the system of FIG. 6. Accordingly, the description of the system of FIG. 7 omits details common to the system of FIG. 6. [0043] The hybrid parallel power supply of FIG. 7 includes a PMIC control module 705 to supply a control signal to the first power supply 505. The PMIC control module 705 may be implemented in various ways. When the first power supply 505 is arranged to receive a digital control signal, the PMIC control module 705 supplies the control signal in digital form. For example, the PMIC control module 705 may signal the first target voltage level to the first power supply 505 using a serial protocol. The PMIC control module 705 may signal the first target voltage level open loop. Alternatively, the PMIC control module 705 may signal the first target voltage level closed loop by, for example, comparing the relative levels of the power rail 520 and the reference output voltage from the bandgap source 625.

[0044] Alternatively, when the first power supply 505 is arranged to receive an analog control signal, the PMIC control module 705 supplies the control signal in analog form. The PMIC control module 705 may signal the first target voltage level to the first power supply 505 open loop by supplying the control signal at the first target voltage level or a scaled version of the first target voltage level. Alternatively, the PMIC control module 705 may signal the first target voltage level closed loop by, for example, comparing the relative levels of the power rail 520 and reference output voltage from the bandgap source 625. The first power supply 505 can utilize the closed-loop control signal to produce a corresponding change in the voltage generated by the first power supply 505 on the power rail 520. The PMIC control module 705, in an example closed-loop analog embodiment, includes a second operational amplifier having an output that provides the control signal to the first power supply 505. The non-inverting input ("+") of the second operational amplifier connects to a supply reference voltage. The inverting input ("-") of the second operational amplifier connects to the power rail 520. The supply reference voltage sets the first target voltage level at which the first power supply 505 supplies power to the power rail 520. The supply reference voltage is produced by a second reference converter that scales the reference output voltage from the bandgap source 625 to produce the supply reference voltage. For example, the second reference converter may scale a 0.64 V reference output voltage from the bandgap source 625 by 45/32 to produce a 0.9 V supply reference voltage. The second reference converter may be a direct current (DC) converter, also known as a DC-to-DC converter. The second reference converter may receive a digital input to adjust the level of the supply reference voltage. [0045] The second operational amplifier may be considered to operate as a comparator in a fashion similar to that described above in relation to the operational amplifier 610. The second operational amplifier compares the voltage on the power rail 520 (applied to the inverting input) to the voltage level of the supply reference voltage (applied to the non-inverting input). The result of the comparison is signaled on the feedback signal to the first power supply 505. A feedback loop from the second operational amplifier to the first power supply 505 back to the second operational amplifier via the power rail 520 works to drive the power rail 520 to the supply reference voltage (first target voltage level).

[0046] The voltage regulator 510, in the embodiment of FIG. 7, includes a stabilization capacitor 750 in the current mirror 615. The stabilization capacitor 750 is coupled between drain and gate of the driver device 634. The stabilization capacitor 750 may be configured to provide an impedance to stabilize the voltage regulator 510. According to certain exemplary embodiments, the stabilization capacitor 750 may be about one-tenth the gate capacitance of the drive device 635. Other compensation techniques may also be used.

[0047] FIG. 8 is a functional block diagram of an electronic system with a hybrid parallel power supply according to a presently disclosed embodiment. The hybrid parallel power supply of FIG. 8 is similar to the hybrid parallel power supply in the system of FIG. 6. Accordingly, the description of the system of FIG. 8 omits details common to the system of FIG. 6.

[0048] The voltage regulator 510, in the hybrid parallel power supply of FIG. 8, includes a reference level module 825 that produces a reference voltage that sets the second target voltage level at which the voltage regulator 510 supplies power to the power rail 520. The reference level module 825 supplies the reference voltage by scaling and low-pass filtering the power rail 520. Setting the second target voltage level at a scaled level relative to the average level on power rail 520 allows the first power supply 505 supply most of the current to the load device 560. The voltage regulator 510 then supplies current to the load device 560 when the power rail 520 drops below the reference voltage. The reference level module 825 may, for example, use a scale factor (of the second target voltage level relative to the level of the power rail 520) of 15/16. The amount of filtering may be chosen, for example, based on the bandwidth of the first power supply 505. [0049] The reference level module 825, in the embodiment illustrated in FIG. 8, includes a first resistor 826 coupled in series with a second resistor 827 between the power rail 520 and the ground reference. The midpoint of the first resistor 826 and the second resistor 827 connects to a filter capacitor 828. The relative values of the first resistor 826 and the second resistor 827 control the amount of scaling of the supply level and the capacitance of the filter capacitor 828 in combination with the values of the first resistor 826 and the second resistor 827 control the amount of filtering. One or both of the first resistor 826 and the second resistor 827 may be variable so that the amount of scaling can be controlled. Additionally, a resistor-DAC may be used in the reference level module 825.

[0050] FIG. 9 is a time-domain graph of a droop characteristic in a system using a hybrid parallel power supply according to a presently disclosed embodiment. The graph illustrates operation of the hybrid parallel power supplies of FIGS. 5-8. Similar to the graph of FIG. 2, the graph of FIG. 9 illustrates an example of a rapid change in current at time 950. The graph of FIG. 9 plots load current 905 for the load device 560 and supply voltage 910 for the power rail 520. Prior to time 950, the load current 905 is 1.25 A. At time 950, the load current 905 rapidly increases to 3.55 A.

[0051] In addition to the step in current demand of the load device at time 950, small variations in the load current 905 occur, for example, as different calculations are performed in a processor in the load device 560. These small variations in the load current 905 cause ripples (small variations, e.g., 4 mV) in the supply voltage 910. The ripples in the supply voltage may be at frequencies that are higher than the bandwidth of the first power supply 505. The first power supply 505 will then supply current the power rail 520 based on a low-pass filtered average of the supply voltage. The ripples in the supply voltage may, however, be at frequencies that are within the bandwidth of the voltage regulator 510. The voltage regulator 510 will then supply current the power rail 520 that reduces the magnitude of the ripples.

[0052] Prior to time 950, the supply voltage 910 is at a nominal voltage level, for example, 0.9 V. At time 950, the rapid increase in load current causes a droop characteristic 925 in the supply voltage 910. The high frequency response of the voltage regulator 510 allows it to quickly increase current supplied to the load device 560 to reduce the magnitude of the droop characteristic 925. An example system has a resultant droop characteristic of < 7% for a 2.3 A/5 ns step change in load current. The apparatus and systems described above relate to a droop characteristic defined in terms of a spikelike lowering of the supply voltage at a load device on a die. A similar effect, but with a positive- going spike, may be realized in certain circuit situations where the current demand of a load device rapidly decreases. Techniques and circuits similar to those described above may be applied, although in a complementary sense, to overcome the fluctuations in the power supply level experienced on a power rail during switching related to load devices.

[0053] FIG. 10 is a flowchart of a process for supplying power to an electronic device according to a presently disclose embodiment. The process will be described with reference to the system of FIG. 5; however, various embodiments of the process may be applied to any suitable apparatus.

[0054] In block 1005, the process supplies current from a power supply to a load device via a power rail at a first target voltage level. In block 1005, an efficient power supply, such as a switching mode power supply, is used. For example, the first power supply 505 can regulate the power rail 520 to a first target voltage level to supply current to the load device 560.

[0055] In block 1015, the process supplies current from a voltage regulator to the load device via the power rail at a second target voltage level. In block 1015, a voltage regulator, such as a low dropout voltage regulator, with high-frequency-response characteristic is used. For example, the voltage regulator 510 can regulate the power rail 520 to a second supply voltage level to supply current to the load device 560. The first target voltage level and the second target voltage level may be different and may be configurable. The process may generate the first target voltage level and the second target voltage level by generating a reference output voltage based on a bandgap source and then generating the second target voltage level based on a first digital input and the reference output voltage and generating the first target voltage level based on a second digital input and the reference output voltage.

[0056] The process of FIG. 10 may be modified, for example, by adding or altering blocks. Additionally, blocks may be performed concurrently.

[0057] Although features the invention are described above for particular embodiments, many variations are possible. For example, hybrid parallel power supplies may be formed using other fabrication processes including processes different types of transistors. Additionally, hybrid parallel power supplies may use different types of voltage regulators and different types of power supplies. Further, hybrid parallel power supplies may have different numbers of power supplies and voltage regulators. In another variation, the voltage regulator 510 can be shut off and removed from providing voltage regulation. In yet another variation, a low-power retention voltage regulator is included on the SoC for use during standby modes. Additionally, features of the various embodiments may be combined in combinations that differ from those described above.

[0058] Those of skill in the art will appreciate that the various illustrative blocks and modules described in connection with the embodiments disclosed herein can be implemented in various forms. Some blocks and modules have been described above generally in terms of their functionality. How such functionality is implemented depends upon the design constraints imposed on an overall system. Skilled persons can implement the described functionality in varying ways for each particular application, but such implementation decisions should not be interpreted as causing a departure from the scope of the invention. In addition, the grouping of functions within a module, block, or step is for ease of description. Specific functions or steps can be moved from one module or block or distributed across to modules or blocks without departing from the invention.

[0059] The above description of the disclosed embodiments is provided to enable any person skilled in the art to make or use the invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles described herein can be applied to other embodiments without departing from the spirit or scope of the invention. Thus, it is to be understood that the description and drawings presented herein represent a presently preferred embodiment of the invention and are therefore representative of the subject matter which is broadly contemplated by the present invention. It is further understood that the scope of the present invention fully encompasses other embodiments that may become obvious to those skilled in the art and that the scope of the present invention is accordingly limited by nothing other than the appended claims.