This application claims the right to priority based on British Patent Application No. 0417686.3, which is hereby incorporated by reference herein in its entirety as if fully set forth herein.

This invention relates to a sensing apparatus and to a method for sensing values of a parameter.

Various forms of inductive sensor have been used to generate signals indicative of the position of two relatively movable members. Typically, one member carries an excitation coil and two or more sensor coils while the other member carries a resonant circuit. The magnetic coupling between the resonant circuit and each of the sensor coils varies with position so that, by applying an oscillating signal at the resonant frequency of the resonant circuit to the excitation coil, a signal is induced in each of the sensor coils which isolates at the resonant frequency but whose amplitude varies as a function of the relative position of the- two members.

One form of inductive sensor is described in international patent application No. WO 03/038379 in which the excitation signal comprises a carrier signal that is amplitude modulated by a lower frequency signal, and the signal induced in the sensor coil is demodulated in a signal processing unit in order to determine the position of the elements. International patent application No. WO 2004/036148 describes an improved form of sensor in which the signal processing unit generates a second signal at an intermediate frequency that differs from that of the excitation signal only by a small amount, and mixes the second signal with the signal received from the sensor coil to generate a low frequency signal whose phase contains information relating to the position of the object. As well as detecting the position of an object, the sensor may be employed to detect external parameters such as temperature and humidity by means of co-located resonant circuits having different resonant frequencies. By obtaining a position measurement for each resonant circuit the difference in the position measurements can form a measure of such external environmental parameter. However, such as sensor has the disadvantage that measurements must be made at about the resonant frequency of each resonant circuit in order to obtain a single value of the parameter. Also, resonant circuits having relatively high quality factors (Q factors) are employed with the result that the intensity of the received signal will decrease relatively rapidly as the resonant frequency of the resonant circuit varies due to variations in the parameter to be determined. If the signal intensity for either of the resonant circuits falls below the noise level of the system, the value of the parameter cannot be determined. According to the present invention, there is provided a sensor for sensing an external parameter, the sensor comprising:

(i) an excitation coil; (ii) a signal generator operable to generate an excitation signal and arranged to apply the generated excitation signal to the excitation coil; (iii)a sensor coil that can be electromagnetically coupled to the excitation coil via an intermediate device such that, in response to the excitation signal being applied to the excitation coil by the signal generator, there is generated in the sensor coil a periodic electric signal; and (iv) a signal processor operable to process the periodic electric signal generated in the sensor coil to determine a value representative of the parameter being sensed;

wherein the intermediate device is sensitive to the parameter to be sensed so that the periodic electric signal generated in the sensor coil is affected by the parameter; the sensor includes means for generating a further signal at substantially the same frequency as the periodic electric signal generated in the sensor coil, which will normally, but not necessarily, be the same frequency as the excitation signal; and the signal processor is operable to determine the value of the external parameter from the periodic signal generated in the sensor coil and the further signal.

By the term "external parameter" is meant any parameter that does not depend on the position of parts of the sensor, for example the position of the intermediate device or so-called "target". Thus for example the sensor may be employed to detect any environmental parameter that will affect the signal generated by the intermediate device, for example temperature, humidity, electromagnetic radiation intensity, the presence or concentration of chemical or biological species and the like. Electromagnetic radiation that is sensed may have any wavelength, and may for example be infrared, visible or ultraviolet, microwave or x-radiation or even γ radiation.

The sensor according to the invention works on the basis that if the intermediate device is sensitive to changes in the external parameter to be measured, any variations in the external parameter will affect the current flowing in the intermediate device, e.g. by changing the resonant frequency and/or by changing the Q-factor of the device in the case of a resonant circuit, which will result in a change in the apparent position of the intermediate device that can be measured.

The sensor according to the invention has the advantage that a single signal at a given frequency is detected by the . sensor coil rather than two separate signals that must be processed separately. Furthermore, because the signals that are processed have substantially the same frequency and so form a single signal, the resulting signal can be processed to determine the value of the parameter even if the signal picked up by the sensor coil from the intermediate device has fallen below the noise level of the system or environment. For example, in the case where the signal received by the sensor coil and the further signal, (which may also be received by the sensor coil from a further intermediate device) have different phases, for example differing by 90°, the resulting detected signal will still have a measurable phase shift even if one of the signals has an amplitude that is below the noise level of the environment. Even if one of the signals has an amplitude that is reduced by 90%, the combined signal will be detectable and will still have a phase shift of 10° or thereabout.

The sensor according to the present invention also has the advantage that the phase change that is employed in order to detect the change in external parameter is not the phase at the frequency of the carrier signal, with the result that errors caused by so-called breakthrough signals are significantly reduced, and often substantially eliminated. Breakthrough signals are signals that are picked up by the sensor coil directly from the excitation coil, i.e. without having been received via the intermediate device, and so have a phase difference of 90° to the signals that are received via the intermediate device (since they are not sent via a resonant circuit) . Because the breakthrough signals are in phase with the excitation signals and out of phase with the signals from the intermediate device by 90°, any interference from the breakthrough signals can prevent accurate and deterministic measurement of the parameter to be sensed. That is to say, although changes in the received signal can be detected, it can be difficult to say what caused such changes. In the sensor according to the invention, a synchronous detector, which will normally be in synch with the received signal from the intermediate device rather than the excitation signal, is normally employed that has the effect of rejecting any breakthrough signals. In the sensor as described in WO 2004/036148, on the other hand, what is sensed is the phase of the signals at the carrier frequency, with the result that the sensor can be extremely sensitive to breakthrough signals, and deterministic sensing can be difficult.

Usually the excitation signal and the further signal will have' frequencies that differ by not more than 50% and especially by not more than 10% at at least one value of the external parameter, and preferably over the entire range of measurement of the parameter, so that the signal intensity will not fall to too low a value and only a single signal is received that is processed. There is indeed no advantage given by employing signals having different frequencies, and the excitation signal and the further signal will therefore normally have substantially exactly the same frequency.

The intermediate device may comprise any other number of devices for example formed from high permeability materials,- e.g. ferrites or electrically conductive materials. Alternatively electrical devices may be used for example active or passive band pass devices (which for the sake of simplicity will be written referred to hereinafter as resonators) . The resonator will preferably have a resonant frequency substantially equal to the frequency of the excitation signal. In this way, unwanted higher harmonics are effectively filtered out by the electromagnetic coupling between the excitation coil and the resonator. The resonator may have a relatively high quality factor, at least for certain values of the parameter to be sensed, for example at least 5 and especially at least 10, the upper limit on the quality factor essentially being set by the ability to ensure a stable resonant frequency with respect to variations in the environment and with respect to tolerances in the values of electrical parameters of the components .

It is possible for the further signal to be generated by the signal generator for the excitation coil and to be sent directly to the signal processor, or to be generated by the signal processor itself since the further signal need not be affected by the parameter to be determined. In most cases, however, the intermediate device will comprise a plurality of resonators, each of which couples the sensor coil to the excitation coil so that both the periodic electric signal and the further signal are transmitted via the resonant circuits. Both or all the resonant circuits may be affected by changes in the external parameter if desired, for example by shifting the resonant frequency in different directions or by altering the Q-factor of the circuit, or one of the resonant circuits may be affected by changes in the parameter while the other may be relatively insensitive to variations in the parameter.

The sensor according to the invention may, for example, be a pulse-echo type sensor, or a continuous excitation plus balanced sense circuit type sensor. According to a preferred aspect of the invention, the sensor is a continuous excitation type sensor which may or may not be ratiometric. Ratiometric sensors have the advantage that they are significantly less sensitive to variations in the system, for example fluctuations in signal levels due to power supply ripple, or changes in the separation (and coupling) between the intermediate device and the sensor. The preferred embodiments are ratiometric inductive sensors, for example those that have two (or more) excitation coils, and has a single sensor coil as described in EP-A-1442273, and those that have a single excitation coil and two (or more) sensor coils as described in EP-A-0760087. Thus, for example, the or each excitation coil may comprise a pair of sinusoidal loops that are arranged in space quadrature so that the same current flowing in each loop will produce magnetic fields in quadrature relationship. One loop of the or each excitation coil may be arranged to produce a magnetic field whose magnitude perpendicular to the plane of the coil varies as the sine of the distance from a reference point, while the other loop of the or each excitation coil may be arranged to produce a magnetic field whose magnitude perpendicular to the plane of the coil varies as the cosine of the distance from the reference point.

The terms "sine" and "cosine" as used herein are not intended to be interpreted as strictly trigonometrical functions, but are intended to include all complementary functions that are in quadrature relationship to each other. For example, the waveforms employed with the excitation coils and maybe digital signals, for example square wave signals, that are produced directly from a digital signal generator (after amplification when necessary) .

According to another aspect, the invention provides a method of sensing an external parameter, which comprises : applying an excitation signal to an excitation coil; and processing a signal generated in a sensor coil which is electromagnetically coupled to the excitation coil via an intermediate device in response to the application of the excitation signal to the excitation coil, in order to determine a value representative of the parameter being measured; wherein the intermediate device is sensitive to the parameter to be sensed so that the periodic electric signal generated in the sensor coil is affected by the parameter; a further signal is generated at substantially the same frequency as the excitation signal; and the value of the external parameter is determined from the periodic signal generated in the sensor coil and the further signal.

Various forms of sensor according to the present invention will now be described by way of example with reference to the accompanying drawings, in which:

Figure 1 schematically shows a perspective view of an inductive position sensor that illustrates the principle of the sensor according to the invention;

Figure 2 schematically shows the main components of the position sensor illustrated in Figure 1;

Figure 3A shows the lay-out of the sine coil which forms part of the position sensor illustrated in figure 1;

Figure 3B shows the lay-out of the cosine coil which forms part of the position sensor shown in figure 1; Figure 3C shows the lay-out of the sensor coil which forms part of the position sensor illustrated in figure 1;

Figure 4 schematically shows how the phase of the signal induced in a resonant circuit which forms part of the sensor element varies with the frequency of driving signal;

Figure 5 schematically shows how the amplitude of the signal induced in the resonant circuit which forms part of the sensor element varies with the frequency of the driving signal;

Figure 6 is a block diagram showing the processing circuitry of a linear variable phase transducer type sensor;

Figure 7 is a block diagram showing the processing circuitry of figure 6 in more detail;

Figure 8 is a block diagram showing the processing circuitry of a linear variable displacement transducer type sensor;

Figure 9 is a schematic showing one form of intermediate device or a target that may be employed in a temperature sensor according to the invention;

Figure 10 is a schematic showing a target that may be employed in a light sensor according to the invention; Figure 11 is a schematic showing a target that may be employed in a chemical sensor according to the invention;

Figure 12 is a schematic of a target that may be employed in a sensor that can detect whether or not the target has been touched;

Figure 13 is a schematic of a substrate forming a target having a pair of resonant circuits;

Figure 14 is a schematic showing a modification of the target shown in figure 12 which may be used for detecting the presence of a finger;

Figure 15 is a schematic of the target that may be employed in a rotary sensor;

Figure 16 is a graph showing the weighted average position of a pair of resonant circuits forming a target as a function of the relative level of induced current in the resonant circuits; and

Figure 17 is a graph showing the sensitivity of such a target as a function of the relative level of induced current in the resonant circuits.

Figure 1 schematically shows a position sensor for detecting the position of a sensor element 1 which is slidably mounted to a support 3 to allow linear movement along a measurement direction (the direction X in Figure 1) . A printed circuit board (PCB) 5 extends along the measurement direction adjacent to the support 3 and has printed thereon conductive tracks which form a sine coil 7, a cosine coil 9 and a sense coil 11, each of which are connected to a control unit 13. A display 15 is also connected to the control unit 13 for displaying a number representative of the position of the sensor element 1 along the support 3.

As shown in Figure 1, the PCB 5 is generally rectangular in shape with the lengthwise axis aligned with the measurement direction and the widthwise axis aligned perpendicular to the measurement direction. The sine coil 7, cosine coil 9 and sense coil 11 are connected to the control unit via a lengthwise edge of the PCB 5, which corresponds to the position value of x equals zero, with the position value increasing along the length of the PCB 5 from the lengthwise edge corresponding to x equals zero.

An overview of the operation of the position sensor illustrated in • Figure 1 will now be given with reference to Figure 2. The control unit 13 includes a quadrature signal generator 21 which generates an in- phase signal I (t) and a quadrature signal Q(t) at respective different outputs. The in-phase signal I (t) is generated by amplitude modulating an oscillating carrier signal having a carrier frequency fo, which in this embodiment is 2MHz, .using a first modulation signal which oscillates at a modulation frequency fi, which in this- embodiment is 3.9kHz. The in-phase signal I (t) is therefore of the form:

I(t) = A sm.2nfλtcos2τif0t n j

Similarly, the quadrature signal Q(t) is generated by amplitude modulating the oscillating carrier signal having carrier frequency fo using a second modulation signal which oscillates at the modulation frequency fi, with the second modulation signal being π/2 radians (90°) out of phase with the first modulation signal. The quadrature signal Q(t) is therefore of the form:

Q(i) = A cos 27fxt cos27fQt , 2 ,

The in-phase signal I (t) is applied to the sine coil 7 and the quadrature signal Q(t) is applied to the cosine coil 9.

The sine coil 7 is formed in a pattern which causes current flowing through the sine coil 7 to produce a first magnetic field B1 whose field strength component resolved perpendicular to the PCB 5 varies sinusoidally along the measurement direction in accordance with the function:

(3) where L is the period of the sine coil in the x direction.

Similarly, the cosine coil 9 is formed in a pattern which causes current flowing through the cosine coil 9 to produce a second magnetic field B2 whose field strength component resolved perpendicular to the PCB 5 also varies sinusoidally along the measurement direction, but with a phase difference of π/2 radians (90°) from the phase of the first magnetic field B1, giving:

In this way, the total magnetic field Bτ generated at any position along the measurement direction will be formed by a first component from the first magnetic field Bi and a second component from the second magnetic field B2, with the magnitudes of the first and second components resolved perpendicular to the PCB 5 varying along the measurement direction.

■ By applying the in-phase signal I (t) and- the quadrature signal Q(t) to the sine coil 7 and the cosine coil 9 respectively, the generated total magnetic field component Bτ resolved perpendicular to the PCB 5 oscillates at the carrier frequency f0 in accordance with an amplitude envelope function which varies at the modulation frequency fi, with the phase of the amplitude envelope function varying along the measurement direction. Thus:

In effect, the phase of the amplitude envelope function rotates along the measurement direction.

In this embodiment, the sensor element 1 includes a resonant circuit having a resonant frequency substantially equal to the carrier frequency fo. The total magnetic field component Bτ therefore induces an electric signal in the resonant circuit which oscillates at the carrier frequency fo and has an amplitude which is modulated at the modulation frequency fi with a phase which is dependent upon the position of the sensor element 1 along the measurement direction. The electric signal induced in the resonant circuit in turn generates a magnetic field which induces a sensed electric signal S (t) in the sense coil 11, with the sensed electric signal S (t) oscillating at the carrier frequency fo- The amplitude of the sensed signal S (t) is also modulated at the modulation frequency fi with a phase which is dependent upon the position of the sensor element 1 along the measurement direction. The sensed signal S(t) is input to a phase detector 23 which demodulates the sensed signal S (t) , to remove the component at the carrier frequency f0, and detects the phase of the remaining amplitude envelope function relative to the excitation waveform. The phase detector 23 then outputs a phase signal P(t) representative of the detected phase to a position calculator 25, which converts the detected phase into a corresponding position value and outputs a drive signal to the display 15 to display the corresponding position value.

By using a carrier frequency fo which is greater than the modulation frequency fχf the inductive coupling is performed at frequencies away from low-frequency noise sources such as the electric mains at 50/60 Hz, while the signal processing can still be performed at a relatively low frequency which is better suited to digital processing. Further, increasing the carrier frequency fo facilitates making the sensor element 1 small, which is a significant advantage in many applications . Increasing the carrier frequency f0 also produces higher signal strengths.

The separate components of the position sensor shown in Figure 1 will now be discussed in more detail.

As shown in Figure 3A, the sine' coil 7 is formed by a conductive track which generally extends around the periphery of the PCB 5 apart from a cross-over point halfway along the PCB 5 in the measurement direction, at which the conductive track on each widthwise edge of the PCB 5 crosses to the corresponding opposing widthwise edge of the PCB 5. In this way, effectively a first current loop 21a and a second current loop 21b are formed. When a signal is applied to the sine coil 7, current flows around the first current loop 21a and the second current loop 21b in opposite directions, and therefore the current flowing around the first current loop 21a generates a magnetic field which has an opposite polarity to the magnetic field generated by current flowing around the second current loop 21b. This results in the sinusoidal variation of the field strength of the component of the' first magnetic field Bi resolved perpendicular to the PCB 5 given by equation 3 above.

In particular, the lay-out of the sine coil 7 is such that the field strength of the component of the first magnetic field Bi resolved perpendicular to the PCB 5 which is generated by current flowing through the sine coil 7 varies along the measurement direction from approximately zero at the point where x equals 0, to a maximum value at x equals L/4 (the position A as shown in Figure 3A) , then back to zero at x equals L/2 (the position C as shown in Figure 3A) , then to a maximum value (having opposite polarity to the maximum value at position A) at x equals 3L/4, and then back to zero at x equals L. Thus' the sine coil 7 generates a magnetic field component perpendicular to the PCB 5 which varies according to one period of the sine function.

As shown in Figure 3B, the cosine coil 9 is formed by a conductive track which generally extends around the periphery of the PCB 5 apart from two cross-over points, located one-quarter and three-quarters of the way along the PCB 5 in the measurement direction respectively. In this way, three loops 23a, 23b and 23c are formed of which the outer loops 23a and 23c are half the size of the inner loop 23b. When a signal is applied to the cosine coil 9, current flows in one direction around the outer loops 23a and 23c and in the opposite direction around the inner loop 23b. In this way, the magnetic field generated by the current flowing around the inner loop 23b has an opposite polarity to the magnetic field generated by the current flowing around the outer loops 23a and 23c. This results in the sinusoidal variation of the field strength of the component of the second magnetic field B2 resolved perpendicular to the PCB 5 given by equation 4 above.

In particular, the lay-out of the cosine coil 9 is such that the field strength of the component of the second magnetic field B2 resolved perpendicular to the PCB 5 which is generated by current flowing through the cosine coil 9 varies along the measurement direction from a maximum value at x equals 0, to zero at x equals L/4 (the position A as shown in Figure 3B) , then back to a maximum value (having opposite polarity to the maximum value at x equals 0) at x equals L/2 (the position C as shown in Figure 3B) , and then back to zero at x equals 3L/4, and then back to a maximum value (having the same polarity as the maximum value at x equals 0) at x equals L.. Thus, the cosine coil 7 generates a magnetic field component perpendicular to the PCB 5 which varies according to one period of the cosine function as given by eguation 4 above.

As shown in Figure 3C, the sense coil 11 is formed by a conductive track which generally extends around the periphery of the PCB 5 forming a single loop.

The layout of the sine coil 7 is such that the electric current induced in the sense coil 11 by current flowing around the first current loop 21a is substantially cancelled out by the electric current induced in the sense coil 11 by current flowing around the second current loop 21b. Similarly, for the cosine coil 9 the current induced in the sense coil 11 by the outer loops 23a, 23c is cancelled out by the current induced in the sense coil 11 by the inner loop 23b. Using such balanced coils has the further advantage that the electromagnetic emissions from the sine coil 7 and the cosine coil 9 diminish with distance at a faster rate than for a single planar winding. This allows larger drive signals to be used while still satisfying regulatory requirements for electromagnetic emissions. This is particularly important because the regulatory requirements for electromagnetic emissions are becoming stricter and stricter.

As described above, when an oscillating drive signal is applied to one or both of the sine coil 7 and the cosine coil 9, an oscillating signal at the same frequency is induced in the resonant circuit of the sensor element 1. However, a phase lag occurs between the drive signal and the induced signal (although this is not the phase lag that is used to determine- the value of the external parameter) , the amount of the phase lag being dependent upon the relationship between the frequency of the drive signal and the resonant frequency of the resonant circuit. As shown in Figure 5A, the phase lag varies most quickly around the resonant frequency of the resonant circuit, with the phase lag at the resonant frequency being π/2 radians (90°) . The higher the quality factor of the resonant circuit, the more quickly the phase varies around the resonant frequency. However, as shown in Figure 5B, the lower the quality factor for the resonant circuit, the less the amplitude of the electric signal induced in the resonant circuit. It is therefore necessary to strike a compromise between signal strength and rate of change of phase with frequency when selecting the value of the quality factor for the resonant circuit.

The overall design of one form of LVPT sensor ■ according to the invention is shown in Figure 6. In such a sensor, a carrier signal of approximately 2MHz frequency is generated by unit 61 in microprocessor 41 and is modulated by two modulation signals produced by units 62 and 63 that are in quadrature relationship to one another. The modulated signals 64 and 66 are then sent to the excitation coils 68 and 70 having a sine and cosine geometry as shown in figures 3A and 3B, from which they are transmitted and picked up by the target 72 in the form of a resonant circuit, normally a passive RLC resonant circuit. The inductor of the target 72 will transmit the signal to the sensor coil of the form shown in figure 3C where the signal voltage will be the sum of the transmitted signals with a modulation phase shift θ governed by the position of the . target with respect to the excitation coils, and will have the general form as shown in box 74. The received signal is synchronously detected with a phase shifted carrier signal 61 and filtered in order to leave the received modulating signal 80 that contains the phase information. This is compared in box 82 with the two modulating signals generated by the microprocessor in order to obtain information regarding the position 84 of the target from the phase shift θ.

The processing circuitry used to generate the in-phase signal I(t), the quadrature signal Q(t) and the anti¬ phase signal I(t), and to process the sensed signal S(t) to determine a position value, will now be described with reference to Figure 7. • As shown in Figure 7, the processing circuitry consists of a microprocessor 41, digital components 61, analogue driving circuitry 81 and analogue signal processing components 91.

The microprocessor 41 includes a first square wave oscillator 43 which generates' a square wave signal at twice the carrier frequency fo (i.e. at 4 MHz) . This square wave signal is output from the microprocessor 41 to a quadrature divider unit 63 which divides the square wave signal by 2 and forms an in-phase digital carrier signal +1 at the carrier frequency, an anti- phase digital carrier signal -I at the carrier frequency and a quadrature digital carrier signal +Q, also at the carrier frequency. As described hereafter, the quadrature digital carrier signal +Q is modulated to form the drive signals applied to the sine coil 7 and the cosine coil 9, while the in-phase and anti-phase digital carrier signals ±1 are used to perform synchronous detection in order to demodulate the sensed signal S (t) .

The microprocessor 41 also includes a second square wave oscillator 45 which outputs a modulation synchronisation signal M0D_SYNC at the modulation frequency fi to provide a reference timing. The modulation synchronisation signal MOD_SYNC is input to a Pulse Width Modulation (PWM) type pattern generator 47 which generates digital data streams at 2MHz representative of the modulation signals at the modulation frequency fi, i.e. 3.9 kHz. In particular, the PWM ' type pattern generator 47 generates two modulation signals which are in phase quadrature with one another, namely a cosine signal COS and either a plus sine or a minus sine signal ±SIN in dependence upon whether the in-phase signal I(t) or the anti¬ phase signal ϊ(t) is to be generated. The cosine signal COS is output by the microprocessor 41 and applied to a first digital mixer 65, in this embodiment a NOR gate, which mixes the cosine signal with the quadrature digital carrier signal, +Q, to generate a digital representation of the quadrature signal Q(t) . The sine signal ±SIN is output by the microprocessor and applied to a second digital mixer 67, in this embodiment a NOR gate, together with the quadrature digital carrier signal +Q to generate a digital representation of either the in-phase signal I (t) or the anti-phase signal I(t) . The digital signals output from the first and second digital mixers 65, 67 are input to first and second coil driver circuits 83, 85 respectively and the amplified signals output by the coil drivers 83, 85 are then applied to the cosine coil 9 and sine coil 7 respectively.

The digital generation of the drive signals applied to the sine coil 7 and the cosine coil 9 introduces high frequency harmonic noise. However, the coil drivers 65, . 67 remove some of this high frequency harmonic noise, as does the frequency response characteristics of the cosine and sine coils 7, 9. Furthermore/ the resonant circuit within the sensor element 1 will not respond to signals which are greatly above the resonant frequency and therefore the resonant circuit will also filter out a portion of the unwanted high frequency harmonic noise. As discussed above, the signals applied to the sine coil 7 and the cosine coil 9 induce an electric signal in the resonant circuit of the sensor element 1 which in turn induces the sensed signal S(t) in the sense coil 11. The sensed signal S (t) is passed through the analogue signal processing components 91. In particular, the sensed signal S(t) is initially passed through a high .pass filter amplifier 93 which both amplifies the received signal, and removes low frequency noise (e.g. from a 50 Hertz mains electricity supply device) and any DC offset. The amplified signal output from the high pass filter 93 is then input to a crossover analogue switch 95 which performs synchronous detection at the carrier frequency of 2 MHz, using the in-phase and anti-phase square wave carrier signals ±1 generated by the quadrature divider 21. The in-phase and anti-phase digital carrier signals which are 90° out of phase to the quadrature digital carrier signal +Q used to generate the drive signals applied to the sine coil 7 and the cosine coil 9, which are used for the synchronous detection, because, as discussed above, the resonant circuit of the sensor element 1 introduces a substantially 90° phase shift to the carrier signal.

The signal output from the crossover analogue switch 95 substantially corresponds to a fully rectified version of the signal input to the crossover analogue switch 95 (i.e. with the negative voltage troughs in the signal folded over the zero voltage line to form voltage peaks lying between the original voltage peaks) . This rectified signal is then passed through a low pass filter amplifier 97 which essentially produces a time-averaged or smoothed signal having a DC component and a component at the modulation frequency fx. The DC component appears as a result of the rectification performed by the synchronous detection process.

The signal output from the low pass filter amplifier 97 is then input to a band-pass filter amplifier 99, centred at the modulation frequency fi, which removes the DC component. The signal output from the bandpass filter amplifier 99 is input to a comparator 101 which converts the input signal to a square wave signal whose timing is compared with the timing of the modulation synchronisation signal MOD_SYNC to determine the position of the sensor element 1.

The sensor according to the invention may also be implemented as a linear variable displacement transducer (LVDT) , for example of the general form as shown in Figure 8. In such a form of transducer, a carrier signal is generated by unit 80 and sent to an excitation coil driver 82 and is transmitted by an excitation coil 84 in the form of a single loop of the configuration as shown in Figure 3C. The signal is picked up by a target 86 in the form of a resonant circuit and transmitted to a sensor coil in the form of a pair of loops, one having a sine configuration as shown in Figure 3A and the other having a cosine configuration as shown in Figure 3B. The received signals as shown in boxes 88 and 90 each have amplitudes proportional to either sineθ or cosθ. The signals received by the loops of the sensor coil are multiplexed and synchronised with the original signal at box 92. After filtering at box 94, a d.c. signal 96 is left that has information relating to the position of the target.

Various forms of sensor in accordance with the invention are described with reference to Figures 9 to 16. Since the electronics relating to the signal generation and processing have been described above, only the targets need be described. In each of these sensors the target is in the form of a pair of resonant circuits where one of the resonant circuits may be used for reference, although it is possible to use only a single resonant circuit in the target and to bleed in the transmit signal from the signal generator as a reference. In addition, it is possible to employ more than two resonant circuits in the target if additional information is desired, such as the position of the target or information relating to other external parameters.

Figure 9 shows a simple target circuit for a temperature sensor in accordance with the invention which employs two resonant circuits having inductors that are physically separated along the measurement direction. The circuit includes a capacitor that is tuned so that both halves of the RLC circuit resonate at the transmitter frequency of the sensor, which implies that the two inductors have the same inductance. In addition, each resonant circuit has a temperature sensitive resistor, one having a positive temperature coefficient (PTC) and the other having a negative temperature coefficient (NTC) . When the voltage is induced across the inductors by the excitation coil, different currents will flow around each of the resonant circuits depending on the relative values of the two resistors at the temperature in question, so that the quality factor of each side of the target will vary with temperature in a sense opposite to the other side, and the amplitude and spatial variation of the electromagnetic field that is generated by the target in response to the excitation coil will vary as the temperature changes.

For an LVDT type of sensor, the transmitted field is generally uniform across the target and the same voltage is induced across both inductors, but as the temperature increases, the current in the left-hand loop will become less than that in the right-hand loop. A higher field will be generated in the inductor L2 than- in Ll, with the result that the total field sensed by the receiver coil will be biased towards the position of L2 so that the targets will appear to have moved to the right of the physical centre of the target. At a low temperatures, the receive coil will detect a field that appears to be generated to the left of centre of the target. Thus, as the temperature changes the apparent position of the target moves in a deterministic manner, and the target may therefore be employed as a remote temperature sensor.

The two resonant circuits, or rather the inductors of the two resonant circuits, may have a range of separations. If the separation between the inductors is increased, so is the resolution of the sensor, but once the separation increases beyond 90° (i.e. one quarter of the sine or cosine loop, the two parts of the target start to interfere in a destructive manner so that the signal level falls.

When ■ the target is interrogated by an LVPT type of sensor, the transmitted field varies across the target so that the currents Il and 12 in the resonant circuits are different and they typically have a different phase. The phase variation can be either at the carrier frequency or at a different modulation frequency. The phase at the received signal is a function of the relative magnitude of the two currents and their phase differences. The effect of the resistors in this case is to bias the- phase of the total currents induced in the sensed coils so that the apparent position of the target changes as a function of the temperature of the target.

It is not necessary for each of the resistors to be temperature sensitive. For example one resistor may be temperature sensitive, either PTC or NTC, while the other resistor is temperature stable, although employing two temperature sensitive resistors will increase the temperature resolution. Alternatively, if desired, other temperature sensitive components such as themistors or thyristors may be used. Also, the reactive components of the resonant circuits may be temperature sensitive instead of the resistors. For example capacitors (e.g. Y5V type) may be employed, or temperature sensitive ferrites . In this case the impedance (i.e. the capacitance or the inductance) of the additional component should be less than that of the tuning capacitor Cl or the inductor Ll or L2 since the additional serious impedance will move the resonant frequency of the system. This movement of the resonant frequency will cause the change in currents Il and 12, but it is necessary for at least one of the two circuits to be close to resonance at the transmitter frequency.

In addition, it is not necessary for the two resonant circuits to share a common capacitor, and indeed in many systems the resonant circuits will be separate from one another.

■ Figure 10 shows a target that, may be employed- in a light sensor. In this case, instead of temperature sensitive components, a pair of back-to-back photo- diodes prevents currents flowing in the left-hand circuit when the target is in the dark. When the target is illuminated, current flows in both sides of the circuit, and the inductive sensor detects an apparent change in the position of the target. The photo-diodes may be specialised in order to detect at narrow optical bandwidths if a frequency selective sensor is desired. The voltage generated must be sufficiently high to overcome the diode drop voltage when the photo-diode is illuminated, and this may be achieved by increasing the number of turns on the target inductor.

The use of non-linear devices such as diodes, transistors and the like in the target, whether employed in temperature sensors, light sensors or other forms of sensor, has the additional advantage that the target can generate signals at frequencies different from the excitation signals, for example at double the frequency (or higher harmonics) which can be sensed by the sensor coil. Such systems enable sensors to be made having significantly improved signal to noise ratio, and are described in co-pending International patent application No. • entitled "sensing Apparatus and Method" in the name of Sensopad Limited and David Alun James, also claiming priority from British application No. 0417686.3, and filed on even date herewith, the disclosure of which is incorporated herein by reference. Thus, the target employed in a light detector could include a pair of back-to-back diodes that are not sensitive to light in one half, and a pair of back-to-back photo-diodes in the other half.

As with the temperature sensing target shown in figure 9, the two resonant circuits and need not share a common capacitor but can be physically and electrically separated. In addition or alternatively, other light sensitive components such as photoresistors or phototransistors may be employed in the resonant circuits.

Figure 11 shows a target that may be employed in a sensor for chemical or biological species. This target employs an ISFET, IMFET or other chemical or biological sensing electrical component that looks like a FET or bipolar transistor. As more chemical is present on the surface of the ion selective membrane of the ISFET, the current Il increases from zero when there are no ions present, with the result that the apparent position of the target moves and the ions can be detected.

The target shown in figure 12 can be regarded as being in two parts, the left-hand part comprising tuned circuit Ll, Cl and C3, and the right-hand part comprising tuned circuit L2, C2 and C3. When the target is in the vicinity of an electromagnetic field, voltages are inductively coupled across the inductors. If these voltages have components around the resonant frequency of either of the tuned circuits of the target, the current flowing in that side, either Il or 12 will be relatively highly due to the low impedance of the targets at that frequency. These high currents will be strongly coupled back to the sensor coil. Cl is chosen so that its capacitance will change due to changes in the environment, . so that the amplitude of the signal coupled from Ll to the sensor coil will also change, whereas the amplitude of the signal coupled from L2 to the sensor coil will not change. The net sensor signal will therefore change accordingly, allowing a determination of the change of capacitance to be made.

As described above, the two resonant circuits of the target may have a common capacitor. This eliminates any changes due to the capacitor itself which is often also sensitive to the environment. However, it is not necessary for the resonant circuits to be connected since they will be inductively coupled, although not directly, by means of the excitation winding(s) and the sense winding(s) . One form of such a target is shown in figure 13 in which two resonant circuits that are isolated from one another are formed on a single substrate. In this case, one of the capacitors may be sensitive to an environmental parameter, for example temperature, humidity or the presence of a finger, while the other capacitor is insensitive.

Another form of finger-sensitive • target is shown in figure 15. In this case, a resistor Rl is connected across Cl. When a finger touches the target, the resistor Rl will be shorted, thereby changing its resistance. This resistance change will cause the frequency response of the left-hand circuit to change, its resonant frequency will shift and the circuit will also have a different quality factor to the right-hand circuit. In this way, the signal sensed from the left-hand circuit will change, thereby unbalancing the system and enabling it to detect the presence of a finger. The components Rl, whether it is regarded as a resistor or not, may, for example, comprise two exposed pads with an air-gap. When no finger is present the resistance across the pads is large, typically greater than 1MΩ, but when the pads are bridged by a finger the resistance drops to values in the range of lkΩ. Indeed, it is possible for such a design to exhibit some change of capacitance across the pads when the finger approaches, so that a two¬ fold effect is obtained.

Such a form ■ of target may be employed with advantage especially in a multiple excitation type system that measures the phase angle of the sensed signal in order to determine the position of the target. The sensor may be arranged to determine the presence or absence of an object for example a finger, and then, once the object has been detected, to determine some other parameter, for example the position of the target. Such a system would enable, for example, a manually operable device such as a knob on an array, for example on a console, to detect when it was being manipulated, and then to determine what position the knob was set to. In the case of a complicated console having a large number of items of equipment, for example 100 or more sensors each controlled by a knob, it can take a considerable length of time to poll each sensor of the console to determine what position it is in or other quantitative value associated with it. However, the detection of whether or not the knob is being grasped, which has only two outcomes, is significantly faster. Thus, it is possible to employ such a target in a system in which the target forms part of a knob or other device to be touched, that is actuated to perform some function such as a quantitative determination only when it is grasped.

In one embodiment employing targets having two resonant circuits, one of the circuits, for example the right-hand circuit as shown in figures 12 to 14, may be tuned to the excitation frequency. If the other circuit is also tuned to this frequency when the capacitor Cl is at its nominal value (for example 0% humidity or with no finger present) the signal on the sensor coil will be generated equally by the signals from each of circuit with the result that the sensor will output the average position of the two circuits. When the environmental factor changes, for example when the target is grasped, the resonant response of the left-hand circuit of the target will be de-tuned from this frequency so that gradually the signal on the sensor coil will become increasingly dominated by the signal from the right-hand part of the target alone.

As with the temperature sensor described above, the sensor can be made more sensitive by arranging that not only is one of the capacitor is sensitive to the environment, but that both capacitors are sensitive in opposite senses to the other. For example, at zero humidity the left-hand circuit may be resonant at the carrier frequency while of the right-hand part is not. As the humidity increases, the left-hand resonant circuit may be de-tuned and the right-hand part may become increasingly tuned to the carrier frequency, so that the apparent position of the target (formed by the pair of resonant circuits) changes.

This can be illustrated mathematically as follows: in a multiple excitation type of system as shown in figure 6, if each separate part of the target is located at a different position this is indicated by the phase shift of the sensed signal. If the left- hand circuit is physically positioned such that it has an amplitude modulated phase shift of 0° and the right- hand circuit is physically positioned such that it has an AM phase shift of φ. The signal level from the left-hand circuit is A and from the right-hand circuit is B, so that the total signal on the sense coil is the sum of these signals:

A cos (coot) cos (ωit) + B cos (coot) cos (coit-φ) (6)

■ This signal is then passed through the synchronous detector and low-pass filter to leave only the AM components:

A cos(coit) + B cos (coit-φ) (7)

This can be re-written in the form: R cos (ωit-θ) in which R = V(A2+2AB cos (φ) + B2) (8) tan(θ) = B sin(φ)/[ (A+B) cos (φ) ] . (9)

Thus, in the nominal situation, when the two signals have the same amplitude (A = B) , it can be shown that θ = φ/2, i.e. that the sensed position is the average position of the two parts of the target. As the right-hand part is de-tuned, B→ 0, and so θ → 0, i.e. the sensed position becomes the position of the left- hand part of the target. Correspondingly, as the left-hand parties de-tuned, A → 0 so that θ → φ, i.e. the sensed position becomes the position of the right- hand part of the target. Figures 16 and 17 have been generated using such a mathematical expression.

In a multiple excitation type system that measures the phase angle of the sensed signal in order to determine the position of the target the maximum angular separation between the two targets is 180°, since in a system of the type as shown in figures 6 or 8 if the angular separation is φ, the weighted average position of the targets can range from 0° to φ. If φ is greater than 180° then the angle takes on the value of 180°- φ. However, at 180° the weighted average position switches from 0° to 180° without passing through any intermediate positions because the signals from the two circuits are out of phase with one another and all that is seen at the sensor is a reduction in signal level without a phase change, until the second target dominates the first, • at which point of the phase reverses.

The weighted average position of the two circuits is shown in Figure 16 as a function of the relative currents in each circuit and for a number of values of the angular separation φ of the resonant circuits, from which it can be seen that the greater the angular separation, the greater the angular change in the target for any given change in the environmental factor, but that as the angular separation φ increases, the response of the sensor becomes increasingly non-linear. Also, as can be seen from Figure 17, as the angular separation φ increases, the amplitude of the received signal decreases. In other words, as the angular separation φ of the circuits increases, the system becomes more accurate (if non¬ linear) but less sensitive. For many applications, the optimum angular separation between the circuits is in the range of from 90° to 150°, preferably from 120° to 150°, and especially about 135° since at angles significantly greater than this the signal level can drop to an extent that the signal-to-noise level becomes an issue. On the other hand, for systems that merely want to react rapidly to a change in the environmental factor, for example the determination of whether the knob is grasped, the optimal spacing between the two circuits is likely to be greater, for example at angular spacings of 170° to 190° and especially about 180° as described below. For embodiments based on the multiple excitation systems, it is possible to arrange the target so that voltages induced across the two inductors Ll and L2 of the targets are equal and opposite. In a rotary system, this may be achieved by having the two inductors spaced 180° apart. If both sides of the circuit are equally tuned to the carrier frequency (i.e. Cl = C2),. the currents Il and 12 will also be equal and opposite so that no net signal will be introduced in the sensor coil. However, when the value of capacitor Cl is changed due to the proximity of the finger, that side of the target is de-tuned away from the carrier frequency. The current Il is thus significantly reduced so that there will be a net signal induced on the sensor winding attributable almost entirely to 12 alone. Accordingly, when there is no finger touching the knob there is no sensed signal, but as soon as the knob is touched, a signal is induced in the sensor coil and this signal indicates angular position of the knob.

Mathematically, this is equivalent to replacing expression as (6) above by one in which φ =• 180°, but since cos (x-180°) = -cos (x) , the sensed signal after the synchronous detector and the low-pass filter is:

In other words, if A > B the sensed position is 0°, and if B > A, the sensed position is 180°, and there are no intermediate sensed positions. When A = B, the amplitude of the sensed signal is small and a threshold detector circuit could be employed so that such small signals are rejected

For embodiments based on the multiple sense type of system, the voltages induced across the two inductors Ll and L2 are generally substantially equal. However, in these embodiments the effect of separating the two inductors by 180° is that if both sides of the circuit are equally tuned to the carrier frequency, the voltages induced in the sensor coils are equal and opposite and so there is no net sensed signal. As one side of the target is de-tuned by the presence of a finger, the signal attributable to the other side of the target dominates and is induced in the sensed windings. Thus, although the signals in the target for the two types of sensor are different, the target designs and the ultimate sensed changes in the external parameter are the same.

The 180° separation of the resonant circuits in the target is particularly advantageous in the form of sensor described above which may be employed in a system in which a large number of sensors, for example in the form of different knobs are arranged in a console, so that any one of the sensors may be activated by touching a knob. In such a system, it is possible to multiplex the sensor coils of each of the knobs as an input and for those electronics rapidly to sense which of the inputs is above a threshold. Only one of the inputs will typically be above the threshold since a user will typically use only one hand to rotate a single knob at any one time. Once the knob being touched by the user has been determined, the slower rotary position measurements can be made on the sensor associated with that knob.

Although this embodiment has been described with respect to rotary sensors using multiple excitation coils, it is also applicable to other forms of sensor, for example linear (2D or 3D) curvilinear, or other sensors using multiple sensor coils, in which case a separation of 180° corresponds to one half of the length of the sense coil.