[0001] This relates generally to level shifters, and more particularly to a level shifter for a wide low- voltage supply range.

BACKGROUND

[0002] As supply voltage continues to drop for advanced microcontrollers, a need exists for low voltage circuits, such as translator products, that will allow these devices to reliably interface with legacy systems. These low voltage translators, in turn, must be capable of supporting a wide voltage range for maximum application flexibility. Existing products support a voltage range from 0.8 V to 3.6 V. However, even lower operating voltages are planned for the future. SUMMARY

[0003] Described embodiments provide a level shifter that contains a boost circuit to overcome delays caused by the adjustments for the wide voltage supply range and provide a faster switching response. The boost circuit includes small NMOS transistors that work as pull-up helpers in parallel with the PMOS transistors in the level shifter. The NMOS helper transistors have their gates driven with the input voltage signals and their effectiveness scales with the input and output voltage levels.

[0004] In one aspect, an embodiment of a level shifter includes a level-shifting circuit coupled to receive first and second input control signals and to provide an output control signal, the level-shifting circuit comprising a first P-type metal oxide silicon (PMOS) transistor and a second PMOS transistor, a first N-type metal oxide silicon (NMOS) transistor and a second NMOS transistor; a third NMOS transistor coupled between an upper rail in a first voltage domain and the drain of the first PMOS transistor, the gate of the third NMOS transistor being controlled by the first input control signal; and a fourth NMOS transistor coupled between the upper rail and the drain of the second PMOS transistor, the gate of the fourth NMOS transistor being controlled by the second input control signal.

[0005] In another aspect, an embodiment of a voltage translator is coupled to translate an input signal received in a first voltage domain (VCCA) to an output signal provided in a second voltage domain (VCCB), wherein each of the first and second voltage domains can span a wide range of low voltages. The voltage translator includes an input buffer coupled to receive the input signal and to provide a first input control signal and a second input control signal, the input buffer operating in the first voltage domain; a level shifter coupled to receive the first and second input control signals and to provide an output control signal; a gate control circuit coupled to receive the first and second input control signals and the output control signal and to provide a first gate control signal and a second gate control signal; and an output buffer coupled to receive the first gate control signal and the second gate control signal and to provide the output signal, wherein the level shifter, the gate control circuit and the output buffer each operate in the second voltage domain, the level shifter comprising: a level-shifting circuit comprising a first P-type metal oxide silicon (PMOS) transistor and a second PMOS transistor, a first N-type metal oxide silicon (NMOS) transistor and a second NMOS transistor; a third NMOS transistor coupled between an upper rail and the drain of the first PMOS transistor, the gate of the third NMOS transistor being controlled by the first input control signal; and a fourth NMOS transistor coupled between the upper rail and the drain of the second PMOS transistor, the gate of the fourth NMOS transistor being controlled by the second input control signal.

BRIEF DESCRIPTION OF THE DRAWINGS

[0006] FIG. 1 depicts an example of an output buffer according to an embodiment.

[0007] FIG. 2 depicts an implementation of an input buffer according to an embodiment.

[0008] FIG. 3 depicts an implementation of a gate control circuit according to an embodiment.

[0009] FIG. 3A depicts an implementation of a gate control circuit according to an embodiment.

[0010] FIG. 4 depicts an implementation of a level shifter according to an embodiment.

[0011] FIG. 5 A illustrates the effects of the added helper transistors on output control signals S _3T and S _2T according to an embodiment.

[0012] FIG. 5B illustrates the increased amplitude of VOU _T as one of the effects of the added helper transistors according to an embodiment.

[0013] FIG. 6 depicts a schematic of a voltage translator according to an embodiment.

[0014] FIG. 7 depicts a schematic of a conventional voltage translator.

[0015] FIG. 8A depicts a set of signals when the voltage translator of FIG. 7 is operated with standard V _T transistors and an input signal operating at 0.8 V is to be translated to an output signal operating at 3.6 V. [0016] FIG. 8B depicts a set of signals when the voltage translator of FIG. 7 is operated with low V _T transistors and an input signal operating at 0.6 V is to be translated to an output signal operating at 3.6 V.

[0017] FIG. 9 depicts an output buffer that could be used to handle the wide voltage supply range.

[0018] FIG. 10 depicts an embodiment of a conventional NAND/NOR pre-driver.

DETAILED DESCRIPTION OF EXAMPLE EMB ODEVIENT S

[0019] In the drawings, like references indicate similar elements. In this description, the term "couple" or "couples" means either an indirect or direct electrical connection unless qualified as in "communicably coupled" which may include wireless connections. Thus, if a first device couples to a second device, that connection may be through a direct electrical connection, or through an indirect electrical connection via other devices and connections.

[0020] The described embodiments evolved from a need to extend the lower voltage range of an existing voltage translator while maintaining support for the existing voltage range of the current device. FIG. 7 is a diagram of a conventional voltage translator 700. Voltage translator 700 receives a signal V _IN , which is operable in a first voltage domain, and translates signal V _IN to an output signal VOU _T , which is operable in a second voltage domain. For the purposes of this description, the first voltage domain has an upper rail designated by VCCA and the second voltage domain has an upper rail designated by VCC _B - In the drawings, the lower rail for both the first and second voltage domains is shown as ground; having each of the lower rails equal to ground is not a requirement.

[0021] Voltage translator 700 contains four main elements: input buffer 702, level shifter 704, a pre-driver 706, which is shown in the drawing as a NAND-NOR pre-driver, and output buffer 708. Input buffer 702 operates within the first voltage domain, which uses upper rail VCCA- Level shifter 704 operates in the second voltage domain, which uses VCC _B , but receives control signals V _INI , V _IN2 , which are generated in the first voltage domain. Pre-driver circuit 706 and output buffer 708 each operate in the second voltage domain. Voltage translator 700 supports a voltage range of 1.1-3.6 V and allows each upper rail VCCA, VCC _B to assume any allowed value within this range. In at least one embodiment, voltage translator 700 is bi-directional, i.e., while the circuits shown translate signals from the voltage domain that uses VCCA to the voltage domain that uses VCC _B , a second copy of this circuit operates to translate signals from the voltage domain that uses VCCB to the voltage domain that uses VCCA- One or more pins on the circuit allow the selection of the desired direction of operation. Because of this bi-directionality, both of the output ports must be able to be placed into a high-impedance mode. In order to support lower voltages, both those currently in use and those planned for the future, changes to the circuit of voltage translator 700 are necessary to support a desired voltage range of 0.65-3.6 V. Individual elements of modules 702-708 are described herein in detail in conjunction with the modifications made to each module.

Parallel VT Architecture:

[0022] A major problem that arises from the support of a wide voltage range is finding complementary metal oxide silicon (CMOS) devices that allow for optimal circuit design architectures. For example, a chip according to the embodiment of FIG. 7 was implemented using standard VT transistors having threshold voltages equal to about 700 mV and operated at voltages that range from 1.1 V to 3.6 V. A test of this chip operating at 0.8 V is shown in FIG. 8A, which depicts the signals Vi _N , VOUT, I _N T2, I _N T3, Vim and Vi _N2 - The input buffer is switching from 0 V to 0.8 V, but signals VINT ₂ , VINT ₃ , produced in the level shifter, do not switch properly. This lack of switching in the level shifter is due to the fact that the VT of the standard VT transistors is very close to the upper voltage rail, so that the transistors did not have the headroom to properly turn on. As a result, output signal VOUT is not being pulled either high or low.

[0023] The circuit of FIG. 7 was then simulated using low VT transistors having a threshold voltage of about 300 mV throughout the circuit. FIG. 8B again depicts the signals VIN, VOUT, VINT2, VINT3, VINI and VIN ₂ . In this simulation, VOUT operated properly and provided a good response to the changes in signal level of VIN- However, other problems arise from the use of low VT transistors, as low VT transistors have greater leakage problems. This issue can be exacerbated when a need also exists to support a wide range of voltages, as in this description.

[0024] Table 1 and Table 2 below depict two implementations of the circuit of voltage translator 700 with low voltage transistors having two different widths. Table 1 depicts the PMOS, low threshold voltage (PCH LVT) transistors, which are trying to pull the output voltage VOUT high. VOH is the output voltage high level and IOH is the output drive requirement. Vcc designates the voltage domain of the output voltage and Spec indicates that the output voltage must remain above the specified value on a "HIGH" value in order for the output to fall with specifications. Actual voltage values achieved during testing are provided for the PCH LVT transistors, first with transistors having a width of 650 microns and second with transistors having a width of 1200 microns. Three values are given for each transistor width and indicate process and temperature: N/27C indicates nominal models at 27 degrees C; W/40C indicates weak models at 40 degrees C and W/125C indicates weak models at 125C. The PMOS transistors having a width of 650 microns were sized initially to allow the circuit to operate at 0.8 V. However, at this width, the circuit was not able to support operation in the other voltage domains; the entries highlighted in bold below each fell below the value allowed by the specifications.

[0025] Similarly, Table 2 depicts the N-channel, low threshold voltage (NCH LVT) transistors, which are trying to pull the output voltage VOU _T low. Here, Spec indicates that the output voltage must remain below the specified value on a "LOW" value in order for the output to fall with specifications. The NMOS transistors are also shown with two widths: 200 microns and 400 microns. The NMOS transistors having a width of 200 microns were also sized initially to allow the circuit to operate at 0.8 V. Again the circuit was not able to support operation in the other voltage domains, as exemplified by the entries highlighted in bold, which fall below the value allowed by the specifications.

Table 1

Table 2

[0026] The widths of both the NMOS transistors and the PMOS transistors were then increased until operation in all of the allowed voltage domains fell within the specifications, as demonstrated by the voltage values shown. The smallest widths at which operation across the entire voltage range could be reached was 1200 microns for the PMOS transistors and 400 microns for the NMOS transistors. Although the voltage specifications could be met with these values, all of the transistors were oversized in order to meet the wide range of voltages. Such oversized transistors not only take up a large amount of real estate on a chip, but also produce output leakage that is far too high to be either desirable or competitive. Therefore, simply replacing all of the transistors in voltage translator 700 with low V _T transistors was not a viable solution.

[0027] Another possible solution to extend the voltage range is to stack several low V _T transistors in series in the output buffer. The problem in the circuit arises from the fact that while low V _T transistors are required by this circuit, the low V _T transistors must still be able to handle 3.6 V. Stacking the low V _T transistors allows each of these transistors to have a lower breakdown voltage (V _D s), since neither transistor is exposed to the entire voltage range. Subjecting the transistors to lower voltages allows for the use of smaller transistors, which in turn have less leakage. Using this configuration, FIG. 9 depicts output buffer 900, which includes two PMOS transistors M _Pi and M _P2 stacked in series between upper rail VCC _B and the output node VOU _T , with resistor R ₅ coupled in series between transistors M _Pi , M _P2 and output node VOU _T - TWO NMOS transistors M _N1 and M _N2 are stacked in series between the lower rail and the output node VOU _T , with resistor R ₆ coupled in series between transistors M _N i, M _N2 and output node VOU _T - The two PMOS transistors M _Pi , M _P2 are each controlled by gate control signal Vp and the two MOS transistors M _N i, M _N2 are each controlled by gate control signal V _N . Gate control signals V _P and V _N are both provided by the gate drive control circuit. Applicant has determined that while the configuration of output buffer 900 is operable, this configuration would require a more complex circuit design for the gate drive control circuit and would also add more risk to high voltage and high temperature reliability.

[0028] In description hereinbelow, the transistors are numbered according to the following notation. For a transistor Μ _χγζ , X has a value of either N or P and indicates whether the transistor is NMOS or PMOS; Y has a value of either S or L and indicates whether the transistors has a standard threshold voltage or a low threshold voltage; and Z has a numerical value that distinguishes the transistor from similar transistors. The described embodiments were formed using proprietary processes that set a standard V _T at 700 mV and a low threshold voltage at 300 mV. However, the described embodiments are not limited by this proprietary process and other values of standard and low threshold voltages can also be used.

[0029] FIG. 1 depicts an example of an output buffer for a voltage translator according to an embodiment. Output buffer 100 operates in the second voltage domain, which in the embodiment shown uses upper rail VCC _B - PMOS transistors, M _PL i and M _P si are coupled in parallel with each other between the upper rail VCC _B and a signal line that provides VOU _T - NMOS transistors, M _NL i and M _NS i are coupled in parallel with each other between the lower rail and the signal line that provides VOU _T - PMOS transistor M _PL i and NMOS transistor M _NL i are each low V _T transistors and are sized to handle voltages under 1 V, while PMOS transistor M _P si and NMOS transistor M _N si are each standard V _T transistors and are sized to handle voltage equal to or greater than 1 V. This differential sizing between standard V _T transistors and low V _T transistors generally extends across example embodiments. However, sizing for other voltage ranges can be used with example embodiments. Each of PMOS transistors, M _PL i, M _P si are controlled by gate control signal V _P and each of NMOS transistors, MN _LI , M _N S _I are controlled by gate control signal V _N ; both of the control signals are received from the pre-driver circuit. Additionally, resistor Ri is coupled between PMOS transistor M _PL i and output signal VOU _T ; resistor R ₂ is coupled between transistor M _P si and output signal VOU _T ; resistor R ₃ is coupled between transistor M _NLI and output signal VOU _T ; and resistor R4 is coupled between transistor MNSI and output signal VQUT- [0030] As described hereinbelow for the pre-driver circuit, gate control signals Vp and V _N can never be ON at the same time. In operation, when gate control signal V _P is low, PMOS transistors M _PL i and M _P si are turned ON and operate together to pull output voltage VOU _T high. When gate control signal Vp is high, PMOS transistors M _PL i and M _P si are OFF and allow output voltage VOU _T to be pulled low. As gate control signal Vp drops, low V _T PMOS transistor M _PL i will turn ON first and provide a quick response. Standard V _T PMOS transistor M _PS i turns ON only when gate control signal Vp is greater than 1 V, but can handle the larger currents necessary at the higher voltages. Similarly, when gate control signal V _N is high, NMOS transistors M _NLI and MNS _I turn ON and operate together to pull output voltage VOU _T low. Low V _T transistor MN _LI will turn ON first and provide a quick response. Standard V _T transistor M _NS i turns ON only when the input voltages are greater than or equal to 1 V, but can handle the larger currents necessary at the higher voltages.

[0031] Tables 3 and 4 below provide similar information to that given in Tables 1 and 2, but show the operational voltages for an embodiment in which gates of low V _T PMOS transistors have widths of 400 microns and lengths of 0.4 microns; standard V _T transistors have gate widths of 800 microns. The gates of the low V _T NMOS transistors are 150 microns wide and 1.7 microns long, while the gates of the standard V _T NMOS transistors are 200 microns wide. As shown in these tables, all levels of operation are within specification.

Table 3

Table 4

[0032] The use of a parallel V _T architecture in applications that can receive a wide range of voltages is not limited to the example shown in FIG. 1. FIG. 2 depicts an input buffer 200 for the same voltage translator according to an embodiment. Input buffer 200 operates in the first voltage domain and includes two inverters 202, 204, which receive an input signal V _IN and provide input control signals Si and S ₂ .

[0033] Inverter 202 includes PMOS low V _T transistor M _PL2 coupled in series with NMOS low V _T transistor M _NL2 between the upper rail VCCA and the lower rail. PMOS standard V _T transistor M _P S ₂ is coupled in series with NMOS standard V _T transistor M _N S ₂ between the upper rail and the lower rail. Each of transistors M _PL2 , M _PS2 , M _M . ₂ , and M _NS2 , are coupled to receive input signal V _IN on a respective gate. The midpoint between low V _T transistors M _PL2 and M _NL2 is coupled to the midpoint between standard V _T transistors M _P S ₂ and M _N S ₂ to provide input control signal Si.

[0034] Inverter 204 includes PMOS low V _T transistor M _PL3 coupled in series with NMOS low V _T transistor M _NL3 between the upper rail and the lower rail. PMOS standard V _T transistor M _PS3 is coupled in series with NMOS standard V _T transistor M _N S ₃ between the upper rail and the lower rail. Each of transistors M _PL3 , M _P s ₃ , MN _L3 and M _N S3 are coupled to receive input control signal Si on a respective gate. The midpoint between low V _T transistors M _PL3 and M _NL3 is coupled to the midpoint between standard V _T transistors M _P s ₃ and M _N S ₃ to provide input control signal S ₂ .

[0035] Using the described combination of low V _T transistors coupled in parallel with standard V _T transistors allows input buffer 200 and output buffer 100 to operate effectively across the entire range of voltages of 0.65 V to 3.6 V. The low V _T devices are sized for drive strength (i.e., current) requirements below 1 V operation while the standard V _T components are sized for the higher voltage drive strength requirements. The combination of low V _T transistors and standard V _T transistors coupled in parallel minimizes the static leakage current while still supporting the full range of device operation. As described herein for the level shifter and pre-driver circuits, many of the transistors in these modules can be implemented with the described low V _T and standard V _T transistors coupled in parallel to allow operation across the larger range of voltages while optimizing the operation across the extended range.

[0036] The described configuration is advantageous in that this configuration allows a designer to have another degree of freedom in the circuit architecture, depending on the product requirements across the full operating voltage range. The transistor widths and lengths for both low V _T and standard V _T components can be selected separately and PMOS devices and MOS devices can each be optimized. Circuits that work over a wider voltage range than existing devices are now possible.

Output Driver Gate Control Circuit

[0037] When designing an output buffer, it is critical to optimize the gate control circuit. This optimization is especially necessary when the output buffer is operated with tristate logic, i.e., the output buffer can be placed in a high impedance state where neither the PMOS pull-ups nor the NMOS pull-downs are enabled. High impedance is required, for example, in embodiments in which current flow can be bi-directional. As described hereinabove, the circuit of FIG. 7 can be implemented with two copies of voltage translator 700, one copy translating from the first domain to the second domain, the second copy translating from the second domain to the first domain. Only one of the two copies can be active at a time, yet the two copies share pins on the chip. Whenever an output buffer is not in use, that output buffer must be placed in a high impedance mode. Improper design of the gate control circuit can allow excess shoot-through current and corresponding ground noise if both output devices are enabled for a short period of time. One method of resolving this issue would be for the gate driver to turn ON the output devices slower to minimize the shoot-through current, but this would result in degraded propagation delay through the data path.

[0038] One existing solution, shown in FIG. 7, uses a NAND-NOR pre-driver 706 for outputs that can be placed in a high impedance state. FIG. 10 is an enlarged reproduction of pre-driver 706. Pre-driver 706 includes two separate circuits: NAND circuit 1002 provides gate control signal Vp and NOR circuit 1004 provides gate control signal V _N . [0039] NA D circuit 1002 has two PMOS transistors M _P3 , M _P4 coupled in parallel between upper rail VCC _B and gate control signal V _P and two NMOS transistors M _N3 , M _N4 coupled in series between gate control signal Vp and the lower rail. Transistors M _P3 and M _N4 are each controlled by a first enable signal ENi and transistors M _P and M _N3 are each controlled by signal V _INT3 , which is received from the level shifter circuit.

[0040] NOR circuit 1004 has two PMOS transistors M _P5 , M _P6 coupled in series between upper rail VCC _B and gate control signal VN and two NMOS transistors M _N5 , M _N6 coupled in parallel between gate control signal V _N and the lower rail. Transistors M _P5 and M _N5 are each controlled by a second enable signal EN ₂ and transistors M _P6 and M _N6 are each controlled by signal V _INT3 from the level shifter circuit. In pre-driver 706, signal V _INT3 provides a level-shifted version of the input signal to input buffer 702 and controls the value of gate control signals V _P and V _N to drive the transistors in output buffer 708. Enable signals ENi and EN ₂ operate to ensure that when the output buffer is placed in high impedance mode, V _P is pulled high to turn OFF PMOS transistors M _PL i and M _PS i in output buffer 100 and V _N is pulled low to turn OFF NMOS transistors M _NLI , M _N S _I - Enable signals ENi, EN ₂ also ensure that the transistors in output buffer 100 are turned OFF during power-on procedures.

[0041] In adapting pre-driver circuit 706 to operate with an extended range of voltages, the majority of the transistors were each replaced by a low-V _T transistor coupled in parallel with a standard V _T transistor, as described herein in the section about parallel V _T architecture. However, due to the additional leakage that low V _T transistors have in relationship to standard V _T transistors (e.g., three orders of magnitude more), it was also considered desirable to eliminate transistors wherever possible to keep the leakage low and the area necessary for the circuit as small as possible.

[0042] FIG. 3 depicts a gate control circuit 300 according to an embodiment. Gate control circuit 300 is specifically designed to drive an output buffer that operates with tristate logic that can be placed in a high-impedance state. Gate control circuit 300 includes four sections: a gate isolation switch 302, pull-up circuit 304, pull-down circuit 306 and an enable/disable control circuit 308. Gate isolation switch 302 provides isolation of gate control signal V _P from gate control signal V _N when necessary but allows gate control signals V _P and V _N to share the pull-up circuit 304 and pull-down circuit 306 when the output buffer is enabled. This is in contrast to circuits 1002, 1004 of FIG. 10, where pull-up transistors and pull-down transistors are coupled to control gate control signal Vp and additional pull-up transistors and pull-down transistors are coupled to control gate control signal V _N . Enable/disable control circuit 308 provides enable signals ENi, EN ₂ , which ensure that gate control signals Vp and V _N can be placed in high impedance when necessary. Pull-up circuit 304 and pull-down circuit 306 use the parallel V _T architecture described hereinabove and provide additional helpers and enable signals as described hereinbelow.

[0043] Enable/disable control circuit 308 includes three inverters, coupled in parallel between upper rail VCC _B and the lower rail. Enable/disable control circuit 308 receives an input signal 310 and provides enable signals ENi and EN ₂ . A first inverter includes PMOS transistor M _P24 and MOS transistor M _N24 ; this first inverter receives input signal 310 and provides an inverted signal 312. A second inverter includes PMOS transistor M _P25 and NMOS transistor M _N25, receives input signal 312 and provides the enable signal ENi. A third inverter includes PMOS transistor M _P26 and NMOS transistor M _N26, receives enable signal ENi and provides enable signal EN ₂ . In one embodiment, the enable/disable control circuit 308 is controlled by the settings applied to pins on a chip containing the described gate control circuit 300. In one embodiment, the value of input signal 310 is controlled by the direction of voltage translation and can also be set by an enable pin.

[0044] Gate isolation switch 302 is at the heart of gate control circuit 300 and includes two PMOS transistors M _PL9 , M _PS9 and two NMOS transistors M _NL9 , M _NS9 coupled in parallel between gate control signal Vp and gate control signal V _N . In accordance with the parallel V _T architecture, transistors M _PL9 and M _NL9 are low V _T transistors, which are selected to operate below IV, while transistors M _P S ₉ and M _N S ₉ are standard V _T transistors, which are selected to operate above IV. The two NMOS transistors M _NL9 , M _NS9 are controlled by a first enable signal ENi and the two PMOS transistors M _PL9 , M _P S ₉ are controlled by a second enable signal EN ₂ . Gate isolation switch 302 connects the gate control signal Vp and gate control signal V _N when the output is enabled and disconnects the output signals when the output buffer is disabled, i.e. in the high impedance state. While gate isolation switch 302 is shown as containing both standard V _T transistors and low V _T transistors, this combination is not necessary in gate isolation switch 302. In another embodiment (not specifically shown) that does not span the wide range of the described embodiment, gate isolation switch 302 includes only a single NMOS transistor controlled by the first enable signal and a single PMOS transistor controlled by the second enable signal. The use of gate isolation switch 302 in place of NAND/NOR gate drivers can reduce the total low V _T transistor width while maintaining consistent drive turn-on.

[0045] The output gate pull-up circuit 304 includes five transistors coupled in parallel between the upper rail, VCC _B , and gate control signal Vp. PMOS transistors M _PL8 and M _P s ₈ are the main pull-up transistors and are controlled by output control signal S _3T , which is received from the level shifter circuit and will drive the gate control signal V _P in response to the input signal received by the voltage translator. However, during development of the overall voltage translator, it was necessary to skew the sizes of the PMOS transistors in relationship to the NMOS transistors throughout the voltage translator. Because of this skewed relationship, helper NMOS transistors M _NL7 and M _NS7 are provided and are each controlled by input control signal Si, which is received from the input buffer 200. A description of the exact manner in which these helper NMOS transistors, MN _L 7, M _N S7, operate to assist the main PMOS transistors, M _PL8 , M _P s ₈ is deferred to the section herein that describes the voltage shifter. Using both standard V _T transistors and low V _T transistors provides for optimal propagation delays across the entire voltage range, while the use of the NMOS helper transistors provides a supply boost where necessary. The final transistor in pull-up circuit 304 is PMOS transistor M _PL i2, which is controlled by enable signal ENi. PMOS transistor M _PL i2 can be used during power-up of the circuit to pull gate control signal V _P high and turn the PMOS output transistors, M _PL i, M _PL2 , OFF. This transistor can be implemented either as shown or using parallel low V _T and standard V _T transistors.

[0046] In a similar fashion, output gate pull-down circuit 306 includes five transistors coupled in parallel between gate control signal V _N and the lower rail. NMOS transistors M _NLH and M _NS ii are the main pull-down transistors and are also controlled by output control signal S _3T from the level shifter circuit. Transistors M _NLH , M _N S _{I I} drive the gate control signal VN in response to the input signal received by the voltage translator. Helper NMOS transistors M _NLIO and M _N S _IO are provided and are each controlled by input control signal S ₂ , which is also received from the input buffer 200. The use of one pair of NMOS transistors driven according to the first voltage domain and one pair of NMOS transistors driven according to the second voltage domain provides for optimal propagation delays across the entire voltage range. Further description of the operation of helper NMOS transistors, M _NLIO , M _N S _IO is again deferred herein to the section that describes the voltage shifter. The final transistor in output gate pull-down circuit 306 is NMOS transistor M _NLI 2, which is controlled by enable signal EN ₂ . NMOS transistor M _NLI 2 can be used during power-up of the circuit to pull gate control signal V _N low and turn the NMOS output transistors, MN _LI , MN _L 2, OFF. As with transistor M _PL i2, transistor M _PL i2 can be implemented either as shown or using parallel low V _T and standard V _T transistors.

[0047] When it is desirable to place the output buffer into a high impedance mode, input signal 310 can be used to set enable signal EN ₁ at the lower rail and to set enable signal EN ₂ at the upper rail. This setting turns ON PMOS transistor M _PL i2 to pull Vp high and turn OFF the PMOS transistors in output buffer 100; this setting also turns ON NMOS transistor M _NLI 2 to pull V _N low and turn OFF the NMOS transistors in output buffer 100. At the same time, transistors M _PL9 , M _NL9 , Mp _S9 , M _NS9 of gate isolation switch 302 are all turned OFF. During normal operation, enable signal ENi is set at the upper rail and enable signal EN2 is set at the lower rail to turn OFF both PMOS transistor M _PL i2 and NMOS transistor M _NLI 2, allowing the other transistors in pull-up circuit 304 and pull-down circuit 306 to control the values of Vp and V _N . This setting also turns ON the switches in gate isolation switch 302. Although not specifically shown in FIG. 3, enable/disable control circuit 308 can also be implemented using the parallel architecture described earlier.

[0048] In contrast to the pre-driver circuit 706, which has been widely used in the past, the described pre-driver circuit only uses a pull-up device for the PMOS gate driver and a pull-down device for the NMOS gate driver along with the connecting transmission gate. The gate control circuit 300 effectively eliminates transistors M _N4 and M _P5 from the design, while combining the associated enable signals and additionally using the advantages of parallel V _T architecture and the helper transistors that assist across the wide voltage range. Using gate isolation switch 302, gate control circuit 300 provides a simpler control circuit from a timing perspective.

[0049] It should be noted that while the described gate isolation switch 302 was originally designed to operate with a wide range of voltages that called for the use of parallel V _T architecture, gate isolation switch 302 can also be implemented in circuits that do not use the parallel architecture. FIG. 3A depicts a gate control circuit 300A according to an embodiment. In this simplified embodiment, gate isolation switch 302A includes a PMOS transistor M _P27 coupled in parallel with NMOS transistor M _N27 between gate control signal Vp and gate control signal V _N . The gate of NMOS transistor M _N27 is controlled by enable signal ENi and the gate of PMOS transistor M _P27 is controlled by enable signal EN ₂ . Pull-up circuit 304A can be configured with a desired configuration of transistors coupled to both the upper rail and to gate control signal V _P . Similarly, pull-down circuit 306A can also be configured with a desired configuration of transistors coupled to both the lower rail and to gate control signal V _N . This simplified version of gate isolation circuit 302A can connect pull-up circuit 304A and gate control signal Vp to pull-down circuit 306 A and gate control signal V _N when gate control circuit 300A is active, yet effectively close the connection when the output buffer controlled by gate control circuit 300A is placed in high impedance mode.

[0050] The described embodiments are advantageous because the gate isolation switch 302 inherently keeps the output in a high impedance state during power-up and provides a natural break-before-make feature due to the transmission gate. Accordingly, when gate control signal Vp goes from a high value to a low value, the charge on gate control signal Vp must discharge through gate isolation switch 302. Before that can occur, gate control signal V _N will first drop, turning OFF NMOS output transistors M _NLI , M _N S _I - Only then can gate control signal Vp discharge through pull-down circuit 306. This connection simplifies the gate control circuit, minimizes the overall die area and minimizes the static leakage from the gate control circuit. In one embodiment, the use of gate isolation switch 302 provided the following advantages over the conventional NAND/NOR pre-driver configuration: gate isolation switch 302 is fifty percent smaller, has a 1.3% reduction in total static leakage current, and the propagation delay is 34.5% lower compared to the NAND/NOR circuit.

Level Shifter Boost Circuit

[0051] A challenging problem that arises as a result of expanding the voltage range of the voltage translator is designing level shifter 400 with sufficient transient response. As a general design rule, the width of the PMOS transistors is twice the width of the NMOS transistors, i.e., a 2: 1 ratio. However, when the level shifter is operating to translate a signal from 0.65V to 3.6V, i.e., the maximum upwards level shift, the NMOS transistors are receiving an ON signal that is barely able to turn the NMOS transistors ON, while the PMOS transistors are receiving a much stronger signal. In order to work properly with this large voltage difference, the PMOS transistor widths must therefore be chosen to be significantly smaller than the NMOS transistor widths. In one embodiment, the resulting ratio between the PMOS and NMOS transistor widths is 1 :3, i.e., the PMOS transistor are much smaller than usual. While this extreme skewing of transistor sizes is necessary when the input signal is low and the output signal is high, this skewing causes poor transient response for low-to-high switching when the input signal is higher and closer to the voltage level of the output. The poor transient response in turn makes fast switching difficult to achieve.

[0052] One possible solution to the extreme skewing of sizes of the PMOS/NMOS transistors can be to use passive resistors coupled in parallel with the PMOS transistors to pull-up the output signal. However, these devices will contribute additional leakage current to the design when the output is driven low and will take up a significant amount of area since the resistors would necessarily have large resistance values.

[0053] FIG. 4 depicts a level shifter 400 that has been implemented according to an embodiment. Of the design shown, level shifting circuit 402 of level shifter 400 is the original level shifter as shown in FIG. 7, although in level shifting circuit 402, all of the previous transistors are replaced by low V _T transistors. Level shifting circuit 402 is made up of two PMOS transistors M _PL i5, M _PL i6 and two NMOS transistors M _NLI S, M _NLI 6- PMOS transistor M _PLI 5 is coupled in series with NMOS transistor M _NL i5 between upper rail, VCC _B , and the lower rail and PMOS transistor M _PL i6 is coupled in series with NMOS transistor M _NLI 6 between VCC _B and the lower rail. The gate of PMOS transistors M _PL i5 is coupled to the drain of PMOS transistor M _PL i6 and the gate of PMOS transistors M _PL i ₆ is coupled to the drain of PMOS transistor M _PL i ₅ . Finally, the gate of NMOS transistor M _NL i5 is controlled by input control signal S ₂ and the gate of NMOS transistor M _NL i6 is controlled by input control signal Si; both of input control signals Si and S ₂ are created in the first voltage domain. This means that NMOS transistors M _NLI S, M _NLI 6 are controlled by signals created in the first voltage domain, while PMOS transistors M _PL i ₅ , M _PL i ₆ are controlled by signals in the second voltage domain, creating the problem noted above. Notably, only low V _T transistors are used for switching capability over the full voltage range from 0.65V to 3.6V on either supply. The width of low V _T transistors is minimized to keep the leakage current as low as possible.

[0054] Rather than providing passive resistors coupled in parallel with the PMOS transistors to pull-up the output signal, level shifter 400 discloses the use of several NMOS transistors coupled in parallel with the PMOS transistors as helper transistors. In the embodiment shown, pull-up circuit 404A includes two NMOS transistors, M _NLI 7, M _N sn, which are each coupled in parallel with PMOS transistor M _PL i ₅ between upper rail VCC _B and output control signal S _3T . The gates of NMOS transistors M _NL i7, M _NS n are controlled by input control signal Si. A second pull-up circuit 404B includes two additional MOS transistors MN _LI S, M _N si8, which are each coupled in parallel with PMOS transistor M _PL i6 between upper rail VCC _B and output control signal S _2T and have their gates controlled by input control signal S ₂ . The size of these helper NMOS transistors M _N Li7, M _N SI7, MNLIS, M _N si8, is small compared to the size of NMOS transistors M _N LIS, M _N LI6- In one embodiment, the helper NMOS transistors M _NLI 7, M _N sn, MN _LI S, M _N si8 _, have respective widths that are one-fifth to one-fourth the width of NMOS transistors M _NLIS , _NL i6- Since the NMOS transistors are driven by the input control signals Si, S ₂ , while the PMOS transistors are driven by output control signals S _2T , S _3T , the effectiveness of the pull-up circuits 404 scales with the input and output voltage levels. Accordingly, when the voltage of upper rail VCCA is low, pull-up circuits 404 will only be weakly turned on. However, since the response of NMOS transistors M _NLIS , M _NLI 6 is weak in this same situation, a strong response from pull-up circuit 404 is not desired. When the voltage on upper rail VCCA is set to be higher and the effect of the skewed sizes of the PMOS transistors M _PL i7, M _PL i8 is very evident, the effect of pull-up circuits 404 is stronger and continues to scale upward as upper rail VCCA is set to higher values. The pull-up circuits 404 help improve the switching time and data rate. A final element of level shifter 400 is a disable switch 406, which in the described embodiment contains a single low threshold voltage NMOS transistor M _N L2i- Disable switch 406 is provided to reduce the dynamic current of the full bitcell by blocking current path of non-switching half bit level shifter. Accordingly, when the output buffer to which the level shifter is coupled is placed in high-impedance mode, the disable switch 406 will turn OFF to keep the level shifter from switching based on the input state. Although not implemented in parallel architecture in the embodiment shown, disable switch 406 could also be implemented in parallel in other embodiments. The width of the low V _T transistor is minimized to keep the leakage current as low as possible.

[0055] FIG. 5A illustrates the effects of the added helper transistors on the output control signals S _3T and S _2T in an implementation of the described level shifter 400. Each of signals S _3T , S _2T is shown both with and without the helper NMOS transistors. Use of the NMOS helpers allows output control signal S _2T to rise more quickly, thus turning off PMOS transistor M _PL i6 more quickly and allowing output control signal S _3T to be pulled down quicker. Level shifter 400 is able to flip faster, so the downstream transistors controlled by output control signal S _3T also switch faster. In testing using an input upper rail VCCA equal to 0.8V and an output upper rail VCC _B equal to 3.3V using weak process models and 130°C, the use of the pull-up circuits 404 improved propagation delay, T _PD , from 15 ns to 14 ns. Using the same process conditions for an input upper rail VCCA equal to 1.65 V and an output upper rail VCC _B equal to 3.0 V, T _PD improved from 3.48 ns to 3.19 ns. This improvement is critical for achieving a data rate of 500 Mbps.

[0056] FIG. 5B is a view of a larger portion of the graph shown in FIG. 5A, but without output control signal S _2T , in order to illustrate the increased amplitude of VOU _T that results from the use of the added helper transistors. As shown in FIG. 5B, using the pull-up circuits 404, output control signal S _3T swings higher. As an effect of the higher swing of output control signal S _3T , VOU _T also swings higher. As described hereinabove, when upper rail VCC _B is 3.0 V, the output high voltage should not drop below 2.3 V. In a test for operation at a data rate of 500 Mbps with VCCA=1.65 V, VCC _B =3.0 V, Weak, 130° C, with no pull-up circuits 404, the high level of VOU _T was 1.49 V, which is less than the required high voltage of 2.3 V at this level. In contrast, when pull-up circuits 404 were added to the circuit, the high level of VOU _T was 2.54 V, which is greater than the required high voltage of 2.3 V. The addition of helper transistors thus allows the level shifter to provide the necessary voltage levels for proper operation. This capability extends across the range of support input and output voltages.

[0057] The described level shifter with NMOS helper transistors is advantageous since it provides a boost to the level shifter output that adjusts with the input and output voltage levels. When the input signal is at a low voltage level compared to the output voltage level, then the NMOS pull-ups provide very little help to the PMOS transistors, which is desirable since the PMOS-to-NMOS ratio is already skewed low. However, when the input signal voltage rail becomes larger, the NMOS pull-ups provide more drive current to pull-up the output signal, which greatly improves the transient response of the level shifter. With the low skewed PMOS-to-NMOS ratio, this extra boost allows the level shifter to provide a good response time across a wide voltage supply range for both input and output levels. In addition, the active pull-ups do not contribute additional leakage current to the design like the use of passive resistors described hereinabove.

[0058] FIG. 6 depicts a voltage translator 600 according to the described embodiments, which is generally a reproduction of the separate circuits described hereinabove, but is provided to give an overview of the various circuits described herein. In this description, references to NMOS and PMOS transistors are used in a generic sense, so these transistors are referred to as metal oxide silicon devices, even though most gates are actually made of polysilicon and other dielectrics besides oxide can be used. Variations to NMOS and PMOS transistors, whether currently known or unknown are covered by these terms.

[0059] Modifications are possible in the described embodiments, and other embodiments are possible, within the scope of the claims.