FIELD OF THE INVENTION

The invention relates to the field of luminaires with presence detectors for lighting networks for home, office, retail, hospitality and industry applications.

BACKGROUND OF THE INVENTION

Antenna devices for use in moving object detection are known, for example from United States patent application US2013/009805 Al, which discloses an antenna device which includes a radiator for radiating an electromagnetic wave, and a dielectric body arranged on an electromagnetic wave radiating side of the radiator, and having a plurality of dielectric members arrayed in a longitudinal direction of the radiator, wherein boundaries between the plurality of adjacent dielectric members are asymmetric with respect to a virtual line perpendicularly passing through the center of the dielectric body in the longitudinal direction.

United States patent application US2006/017636 Aldiscloses an electromagnetic wave transmitting device, which includes a direction-change element for changing the direction of travel of electromagnetic wave transmitting elements disposed on a substrate. The direction-change element has a periodic array of materials of different refractive indexes arranged in parallel to the substrate surface. The electromagnetic wave transmitting elements are positioned at opposite ends of the periodic array in the direction of its arrangement. By changing the relative intensity of electromagnetic waves to be sent from the two electromagnetic wave transmitting elements, it is possible to achieve high-accuracy control of the angle of emittance of the electromagnetic wave to be sent from the device.

Lighting networks comprise at least one luminaire device (e.g. a smart lamp such as a light emitting diode (LED) based lamp, gas-discharge lamp or filament bulb, plus any associated support, casing or other such housing) arranged to emit illumination in into an environment. The environment may be an indoor space such as one or more rooms and/or corridors of a building; or an outdoor space such as a park, garden, road, or outdoor parking area; or a partially covered space such as a stadium, structured parking facility or gazebo; or any other space such as an interior of a ship, train or other vehicle; or any combination of such possibilities. The luminaire device may take any suitable form such as a ceiling or wall mounted luminaire, a free-standing luminaire, a wall washer, a chandelier; or a less conventional form such as embedded lighting built into an item of furniture, a building material such as glass or concrete, or any other surface.

An area of interest for smart, connected lighting systems is human presence detection. Passive Infrared (PIR) sensors have been used for this purpose for many years but they require an infra-red transparent‘window’ to be present such that a direct view of the area for monitoring is provided. PIR systems are incapable of looking through translucent materials such as plastic, glass or fabrics. In addition, a desire to have lower cost LED drivers without galvanic isolation adds considerably to the cost and complexity of achieving a given international protection marking if an opening must also be present for the PIR sensor. PIR sensors are also not capable of achieving the kind of detection ranges required for such applications as high-bay lighting.

SUMMARY OF THE INVENTION

It is an object of the present invention to provide a luminaire device for enabling improved presence detection.

This object is achieved by a luminaire device for use in a lighting network as claimed in claims 1 and/or 3, and by a lighting system as claimed in claim 10.

Accordingly, an integrated RADAR system for luminaire devices is proposed, which does not cause any deterioration in the quality of light output, as the dielectric waveguide can have a physically low profile. This makes it amenable to fabrication by additive manufacturing. The resulting dielectric waveguide can provide high antenna gain and low losses. In addition when combined with simple frequency steered functionality the claimed invention will expand the capabilities and ease the installation and use of Doppler RADAR systems at microwave and millimeter-wave frequencies.

According to a first option, the dielectric waveguide may be made from an optically transparent or optically reflective material. Thereby, the dielectric waveguide can be constructed cheaply and easily from optically transparent or highly reflecting polymer materials so as to prevent deterioration of the quality of light output of the luminaire device.

According to a second option which can be combined with the first option, the perturbation pattern may comprise a thin-film metal strip grating. The grating can be easily generated and requires little additional space. Moreover, the grating pattern can be adapted to achieve a predetermined directivity and angle of the directional beam at a given frequency or frequency range.

According to a third option which can be combined with the first or second option, the dielectric waveguide may be excited from both ends simultaneously by a pair of the RADAR transceivers. This results in the generation of a pair of directional beams that point in opposite directions to one another.

According to a fourth option which can be combined with any one of the first to third options, the perturbation pattern may comprise a thin-film metal strip grating, wherein a pitch of the metal grating increases as the distance from the two ends of the dielectric waveguide increases to a maximum grating pitch value in the center of the dielectric waveguide.

According to a fifth option which can be combined with any one of the first to third options, the perturbation pattern may comprise a thin-film metal strip grating, wherein a gap spacing of the metal grating decreases as the distance from the two ends of the dielectric waveguide increases to a minimum gap spacing in the center of the dielectric waveguide.

According to a sixth option which can be combined with any one of the first to third options, the perturbation pattern may comprise a plurality of grooves cut into the dielectric waveguide, wherein a pitch of the grooves increases as the distance from the two ends of the dielectric waveguide increases to a maximum pitch value in the center of the dielectric waveguide.

According to a seventh option which can be combined with any one of the first to third options, the perturbation pattern may comprise a plurality of grooves cut into the dielectric waveguide, wherein a gap spacing of the grooves decreases as the distance from the two ends of the dielectric waveguide increases to a minimum gap spacing in the center of the dielectric waveguide.

In the above fourth to seventh options, the changed pitch or gap spacing of the metal grating or grooves provides a simple and efficient conversion of the energy guided by the dielectric waveguide into energy radiated as an antenna beam. From the above it will be clear that the thin-film metal strip grating causes an effect comparable to that of the grooves cut into the dielectric waveguide; i.e. both features cause energy of the RADAR signal to leak outwards from the dielectric waveguide. The respective pitch variation and/or gap spacing in relation to the RADAR transceiver causes the formation of a directional beam. In this manner these alternatives all are based on one and the same inventive concept. The direction beams radiated into space can be scanned in space simply by changing the frequency of the RADAR signal.

According to an eighth option which can be combined with any one of the first to seventh options, the housing part may be an LED circuit board of the luminaire device. Thereby, the proposed presence detection RADAR system can be simply placed on the LED circuit board and does not require any other modification of the luminaire housing.

According to a ninth option which can be combined with any one of the first to eighth options, the housing part may comprise a wall of a housing of the luminaire device. Thereby, the proposed presence detection RADAR system can be simply placed as a transparent or reflective module somewhere on the wall of the luminaire housing and does not influence the light quality of the luminaire device.

According to a tenth option which can be combined with any one of the first to ninth options, the at least one RADAR transceiver may be a CW Doppler RADAR transceiver. This measure simplifies signal processing, as no modulation is required and movements lead to a small Doppler shift of the frequency of the received reflected signal, which can be detected by a filter with suitably selected low-frequency passband.

According to an eleventh option which can be combined with any one of the first to tenth options, the at least one RADAR transceiver may be adapted to sweep the frequency of the RADAR signal in order to change an angle of the directional beam.

Thereby, both directional beams of the RADAR transceivers can be steered to another location by switching the frequency to another value. By operating the RADAR transceivers simultaneously at different frequencies, interference can be eliminated and they can operate independently from each other without the risk of injection locking.

According to a twelfth option which can be combined with any one of the first to eleventh options, the at least one RADAR transceiver may comprise a comb generator for generating a plurality of signals with different frequencies, and at least one bandpass filter for selecting a frequency of the RADAR signal and a resulting angle of the directional beam. Thereby, a greater area can be covered for presence detection, since a plurality of directional beams all pointing in different directions can be created simultaneously.

It shall be understood that a preferred embodiment of the invention can also be any combination of the dependent claims or above embodiments with the respective independent claim. These and other aspects of the invention will be apparent from and elucidated with reference to the embodiments described hereinafter.

BRIEF DESCRIPTION OF THE DRAWINGS

In the following drawings:

Fig. 1 schematically shows a side view of a leaky-wave antenna system mounted on a luminaire housing structure;

Fig. 2A schematically shows a top view of a luminaire circuit board with leaky -wave antenna driven by a pair of transceivers;

Fig. 2B schematically shows a top view of a luminaire circuit board with leaky -wave antenna driven by a pair of transceivers;

Fig. 3 schematically shows a block diagram of a dual-beam frequency-steered continuous wave Doppler RADAR system with frequency sweep control; and

Fig. 4 shows a block diagram of a multi-beam frequency-steered continuous wave Doppler RADAR system with frequency comb generator.

DETAILED DESCRIPTION OF EMBODIMENTS

Embodiments of the present invention are now described based on frequency- steered multi-beam antennas and associated circuit architectures for RADAR (Radio Detecting and Ranging) equipped luminaire devices.

In general, RADAR systems operating at microwave or millimeter-wave frequencies do not suffer from the initially mentioned problems of PIR sensors. They can transmit and receive signals, with appropriate choice of materials and thicknesses, directly through translucent materials of the types encountered in luminaire devices (e.g. bulbs, luminaires etc.). With a sufficient system design (analyzing antenna gain, transmitter output power, receiver bandwidth and gain, path losses etc.) RADAR presence detection systems also do not suffer the range limitations inherent to PIR sensors.

In order to minimize the space occupied in a luminaire device by the RADAR system’s antenna it is advantageous to use operating frequencies in the microwave and millimeter-wave bands (this also avoids problems of room confinement as seen at 2.4 and 5.8GHz) where the wavelengths are shorter and so the physical dimensions of the antenna and/or antenna elements are smaller. On the other hand, with increasing frequency the losses due to metallic conductors, both in rectangular waveguide and printed circuit antennas, rise and even the surface roughness of the metals becomes a significant source of additional loss. For longer range applications, such as high-bay lighting, arrays of printed circuit antenna elements (typical patch antennas) may be required in order to have sufficient antenna gain in the RADAR system for the signal-noise ratio to be sufficient for reliable motion detection.

Antenna array also occupy space within the luminaire device which could otherwise have been used for the placement of LEDs. The antenna arrays themselves need an appropriately designed feeding network which adds another source of loss to the antenna. As operating frequencies rise, simple“FR-4” grade glass-epoxy resin materials used for low frequency electronics may be replaced by specialized low-loss dielectric materials (e.g.

Taconic, Polyflon) may be used.

The substrate thickness of printed antenna elements (e.g. dipoles and patch antennas) has to be reduced as the operating frequency increases. For the case of a patch antenna, the substrate thickness should be less than around 0.11 times the free space operating wavelength. This is to ensure resonance of the patch antenna and also to limit the chance for the excitation of surface waves which can be an additional source of loss as well as causing mutual coupling between antenna elements in an array which can degrade the final overall antenna beam pattern.

For microstrip transmission lines, when the substrate thickness decreases the width of the metal tracks must shrink in order to maintain a given value of characteristic impedance of the transmission line. Given that conventional printed circuit board fabrication uses wet-etched metal layers, the minimum linewidth that can be achieved is around twice the metal thickness. This may present difficulties in the design and fabrication of printed circuit antennas at microwave and millimeter-wave frequencies, as it will limit the highest values of microstrip line impedance that can be achieved at reasonable cost which can present limitations in terms of circuit impedance matching performance between the antenna(s) and the active circuitry.

For greater system functionality (reconfigurability and easy installation), it could be advantageous to have a beam steered antenna array. The two traditional approaches to this are a passive and an active electronic beam steered array. In the former one (passive electronic beam steered array), a single transceiver drives multiple antennas through a power dividing network with each element in the array provided with a controllable phase shifter. In the latter one (active electronically beam steered) array each individual element has its own transceiver attached to it and the phases and amplitudes of the signals applied to each element are adjusted at the element level. This is considerably more complicated than the passive array but does provide built-in redundancy in case of individual active element failure unlike the passive array which is completely useless if the transceiver stops working. In both cases, a considerably quantity of additional active circuitry (phase shifters plus digital control or multiple transceiver modules all phase-locked together) is necessary to provide beam steering.

In order to provide a low thermal resistance support (heatsink) to help remove the waste heat generated by the light emitting diodes (LEDs) when they are switched on, a metal cored printed circuit board (so-called Level-2 board) may be used in the luminaire device to mount the LEDs on. This may be an aluminum plate onto which an electrically isolating, thin-film dielectric layer may be deposited followed by a low electrical resistivity conductor pattern (made of e.g. Copper) through which electrical power is applied to the LEDs.

Direct integration of an antenna or antenna array into luminaire devices may require the use of a separate printed circuit board. In case of microwave/millimeter-wave RADARs operating in the industrial-scientific-medical (ISM) bands (24.0 - 24.25GHz, 61 - 61.5GHz and 122 - 123GHz), special low-loss dielectric substrate materials may be used for the fabrication of circuitry and antennas. In addition, the antenna(s) of the resulting module should have a field of view that encompasses the space being illuminated, so that they should be placed within the light generating region of the luminaire device.

Leaky wave antennas are a class of traveling-wave antennas in which a wave propagates along a long structure as compared with the wavelength. They are very similar to surface wave antennas. Like most traveling-wave antennas, leaky wave antennas are long in the propagating direction and possess a cross-section with dimensions about the wavelength of the operation. A distinguished feature of these antennas is that the electromagnetically field is excited by a wave, which is incident on the interior or the exterior of a guiding structure, which produces currents that propagate along its longitudinal direction. When transmitting, the input traveling wave, often a fast wave, progresses along the guiding structure and leaks out energy from the structure so that only a negligible field is left at the termination end of the traveling wave antenna. Planar leaky -wave antennas that have been designed using simple microstrip lines on a dielectric substrate layer and a ground plane behave similarly, wherein the wave propagates along a transmission line and leaks when it encounters discontinuities or perturbations in the structure.

If such discontinuities or perturbations are introduced along the structure, for example slits or apertures for waveguides, patches, or stubs for microstrip lines etc., the traveling wave leaves the structure and radiates into free space. Thus, in the ideal case, no energy reaches the end of the structure. In a practical scenario, any energy that reaches the end is absorbed by a matching load. Such antennas can be constructed in the form of an insert that projects barely (low profile) from the support surface. This feature is very important for installations that are restricted by size or shape, as in luminaire devices.

Planar leaky -wave antennas have garnered interest due to their ability to efficiently scan the full-space. For single-frequency operation, electronically scanning leaky- wave antennas having no feeding network and low-cost varactors consuming no power, may be a viable alternative to phased-array antennas which utilize expensive and power-hungry phase shifters with bulky and lossy feeding networks. Such an electronically scanning leaky- wave antenna is excited at one port and is terminated at the other by a matched load. As an alternative, an end-switched electronically scanning leaky -wave antenna with two input ports at both ends may be provided, where the input signal is routed to either port via a switch.

Each port may provide half-space scanning and thus the full-space is scanned via the switching mechanism. This scheme may be applied to any two-port leaky wave antenna (with and without electronic-steering capabilities).

Fig. 1 schematically shows a side view of a leaky-wave antenna system mounted on a luminaire housing structure according to various embodiments.

The leaky wave antenna comprises a low profile, low-loss dielectric transmission line or waveguide 140 fabricated from an optically transparent or highly reflective (over the visible spectrum) opaque material onto which a perturbance 150 (e.g. a thin-film metal strip grating or other discontinuity) is placed or fabricated. The perturbance 150 perturbs the normal propagating mode (propagation direction indicated by the bold arrow in Fig. 1) of the dielectric waveguide causing energy to leak outwards from the guide forming giving a fan-shaped (antenna) beam 160. The dielectric waveguide 140 may be excited by a single continuous-wave (CW) Doppler RADAR transceiver 120 from one end, while the other end is connected to a waveguide-adapted load 130 (i.e., a termination load matched to the characteristic impedance of the system) suitable to absorb remaining wave energy without causing any reflections.

As an alternative, the waveguide 140 may be excited from both ends simultaneously by a pair of continuous-wave (CW) Doppler RADAR transceivers. This results in the generation of a pair of fan shaped antenna beams 160 that point in opposite directions to one another. Due to a fixed periodicity in the perturbation (e.g. pitch of metal grating or other discontinuity) and a high dispersion, which results in a frequency-dependent change of the complex propagation of the dielectric waveguide, a change in the excitation frequencies applied to the dielectric waveguide causes the angles of the antenna beam(s) 160 relative to broadside (horizontal plane in Fig. 1) to change as well. This means that the RADARs’ antenna beam(s) 160 can be scanned in space simply by changing the frequency of their local oscillator(s) of the transceiver(s) 120.

Thus, a considerably more compact, simpler and lower cost approach for a beam-steerable antenna setup can be provided, compared to the conventional use of phase shifters to build a phased antenna array which would require an array of simple antennas (such as patch antennas) all equipped with their own phase shifters and associated control electronics.

By using a low dielectric loss tangent, optically transparent or highly reflecting opaque material, the leaky-wave antenna can be directly integrated into a luminaire, e.g., housing wall 100 or circuit board or other housing portion, without causing a deterioration in the light output or light quality.

Given that a fairly long section of the dielectric waveguide 140 with its perturbance 150 may be necessary to ensure that the majority of the millimeter-wave energy is radiated as the antenna beam 160, the proposed structure is well suited to fluorescent tube replacement TLED-type light sources although by using a higher dielectric permittivity material the physical length of the dielectric waveguide 140 can be reduced which could make it fit into other form factors as well.

In addition, due to the channel bandwidths available at microwave and millimeter-wave frequencies, the losses of the dielectric waveguide 140 can be expected to be lower than for one of the more common metal-dielectric transmission-line antenna types such as microstrip patch antennas or (ground) coplanar waveguide radiators. In comparison with air-filled metallic rectangular waveguides, the proposed dielectric leaky-wave antenna approach can be expected to be cheaper to manufacture, lighter weight as well as offering optical transparency or high optical reflectivity in the visible light range. It also removes the need for large areas of expensive, low-loss microwave/millimeter-wave optimized printed circuit board materials to hold the RADAR circuitry and antenna(s).

Fig. 2A and 2B schematically show a top view of an LED circuit board 200 equipped with a dual-beam, frequency steered, leaky -wave antenna driven by a pair of CW Doppler RADAR transceivers 220.

The system of Fig. 2A can be provided on the circuit board 200 (e.g. a level-2 metal core printed circuit board) of the luminaire device with a plurality of LEDs 210 arranged in two parallel rows. Between the two rows, a low profile, low-loss dielectric transmission line or waveguide 240 fabricated from an optically transparent or highly reflective (over the visible spectrum) opaque material onto which a thin-film metal strip grating 250 is placed or fabricated. The pitch of the metal grating 250 increases as the distance from the two ends of the dielectric waveguide increases to a maximum grating pitch value pi in the center of the waveguide 240 while the gaps gs between the thin-film strip portions remain constant. The metal grating perturbs the normal propagating mode of the dielectric waveguide 240 causing energy to leak outwards from the guide forming giving a fan-shaped (antenna) beam directed into the viewing direction of the viewer of Fig. 2 A.

As an alternative option, Fig. 2B, the pitch p of the thin-film metal strip grating 250 could be maintained and the spacing or gaps between the strip portions could decrease as the distance from the two ends of the dielectric waveguide 240 increases to a minimum gap gsi in the center of the waveguide 240.

As the further alternative, the perturbance 150 of Fig. 1 could be achieved by cutting grooves into the dielectric waveguide 240 of Fig. 2A or 2B. Here, the same modification about pitch (center-to-center spacing) and gap between the grooves 240 applies as used for the thin-film metal patch perturbations above. I.e., as in Fig 2A, the pitch of the grooves increases as the distance from the two ends of the dielectric waveguide increases to a maximum groove pitch value in the center of the waveguide 240 while the gaps between the grooves remain constant. Or as in Fig. 2B, the length of the grooves could be maintained and the spacing or gaps between the grooves could decrease as the distance from the two ends of the dielectric waveguide 240 increases to a minimum gap in the center of the waveguide 240.

In the embodiment of Fig. 2A or 2B, the dielectric waveguide 240 is excited from both ends simultaneously by the pair of continuous-wave (CW) Doppler RADAR transceivers 220. This results in the generation of a pair of fan shaped antenna beams that point in opposite directions to one another. Due to the pitches of the thin-film metal grating 250 and the high dispersion, a change in the excitation frequencies applied to the dielectric waveguide 240 causes the antenna beams’ angles relative to broadside to change as well.

This means that the antenna beams can be scanned in space simply by changing the frequency of their local oscillators.

Fig. 3 schematically shows a block diagram of a generic dual-beam frequency- steered CW Doppler RADAR system with frequency sweep control for use with a dielectric leaky wave antenna 300 e.g. in a luminaire device, according to various embodiments.

The connections between two CW Doppler RADAR transceivers 222 and the ends of the dielectric waveguide 300 can be made using a set of multiplexers and switch matrices 226. In the basic example shown in Fig. 3, a diplexer is formed from the parallel connection of two filters 228, 229 designed as a pair such that there is a constant real impedance (ignoring parasitics and component tolerances) looking into the parallel filter connection at all frequencies covered by the combined passbands of the filter pair. More specifically, the filter pair comprises a low-pass filter 228 and high-pass filter 229, but it could just as easily be a lowpass and bandpass combination or a pair of bandpass filters. In principle there is no reason why more filter sections could not be used so as to permit more antenna beams to be generated without interference between the two transceivers.

The constant real input impedance of the diplexer ensures that all incident energy arriving at its input is transmitted through one of the filters 228, 229, or both filters 228, 229 for input frequencies equal to the crossover frequency. This means that there should be no energy reflected as a result of signals being in the stopband of either filter. This constant marched termination of the dielectric waveguide 300 prevents spurious radiation from the ends of the dielectric waveguide 300 which could otherwise degrade the desired antenna radiation pattern. In addition, by operating the two RADAR transceivers 222 simultaneously at different frequencies, each one located in the passband of a different filter in the diplexer, interference between the two transceivers 222 is eliminated and they can be operated and perform beam steering independently from one another without the risk of injection locking, generation of intermodulation distortion or incorrect RADAR target detection due to the down-conversion of one transceiver’s output signal in the other transceiver’s receiver stage.

In order to generate a broadside antenna beam the two transceivers 222 can be switched to the same nominal frequency that produces a near-broadside beam. If local oscillators of the two transceivers are phase coherent then the signals transmitted will add together and the two beams will add constructively to give a broadside antenna beam albeit with a wider half-power beam angle than the off-broadside beams. When changing from one filter passband to another, the switch matrix 226 is used to change the transceiver connection over and at the same time connect a termination load 224, matched to the characteristic impedance of the system, to the unused filter’s output port. The thin-film metal grating or other type of perturbation or discontinuity placed or deposited on top of the dielectric waveguide 300 can be designed to increase the amount of energy radiated as the antenna beam and decrease the energy that the terminating filter stage is required to absorb. For the case of a synthesized near-broadside antenna beam, the residue signal will be equally divided between the terminating filter stage and filter connected to the second transceiver. This halves the signal that would otherwise be down-converted to DC by a frequency mixer which greatly reduces the chance of saturating the subsequent receiver intermediate frequency (IF) circuitry.

During operation the system shown in Fig. 3 will use two different transmit frequencies. Each one will be located in a different filter passband e.g. the left-hand transceiver 222 generates a transmit frequency that is within the passband of the low-pass filter 228 while the right-hand transceiver 222 generates a transmit frequency that is within the passband of the high-pass filter 229. As a result, the left-hand high-pass filter 229 is connected via the switch matrix 226 to the characteristic impedance (Zo) terminating load 224 so that any residual power from the right-hand transceiver output is absorbed. Similarly, the right-hand lowpass filter 228 will be connected via the switch matrix 226 to the characteristic impedance termination load 224 so that any residual power from the left-hand transceiver output is absorbed. In order to steer both beams to another location the two transceivers switch or change their frequency and in so doing the switch matrices 226 are reconfigured so as to terminate the filter with a passband corresponding to the transmit frequency of the other transceiver 222.

Fig. 4 shows a block diagram of a multi-beam frequency-steered continuous wave Doppler RADAR system with frequency comb generator according to various embodiments.

In order to cover a greater area of space (e.g. a room with luminaire device(s)) with Doppler RADAR sensing beams without the requirement for duplicating all of the microwave/millimeter-wave circuitry, the architecture shown in Fig. 4 can be used. Here, instead of a single voltage-controlled oscillator or frequency synthesizer being used to generate the transmitter frequency, respective frequency comb generator circuits 421 are used at both ends of the dielectric waveguide 300. This simultaneously generates multiple frequency tones with the result that, due to the dispersion in the propagation constant of the dielectric leaky -wave antenna, multiple antenna beams all pointing in different directions are created. A wide variety of possible means of generating the frequency comb exist. These include the use of high-speed digital logic in the form of a logic inverter stage combined with either an XOR or NAND gate, a bipolar transistor driven into avalanche mode, a step- recovery diode or a pseudo-random sequence driven logical AND gate fed pulse generator.

Thereby, a multiple beam presence detection system for luminaire devices of lighting systems can be provided. The dense‘picket fence’ of frequency tones at the output of the comb generators 421 is passed through respective bandpass filters 422 to select only those tones of the appropriate frequency band for use in generating antenna beams from the leaky -wave antenna. Depending upon the implementation and the desired operating frequency band the output of the comb generator circuits 421 might first be passed through a frequency multiplier (not shown in Fig. 4), e.g., with integer multiplication factor, or via a single sideband upconverter in order to have tones in the required frequency band for the

RADAR/antenna combination.

The base frequency used to drive the frequency comb generator circuits 421 sets the frequency -frequency spacing of tones in the resulting comb. In Fig. 4 either a passive ferrite circulator 421 or an active circuitry quasi-circulator, enabling a single integrated circuit (IC) design incorporating all the circuit blocks shown, is used to separate out the transmitted and reflected return signals from each other. The return signals are applied to a filter multiplexer consisting of a number (three are shown in Fig. 4 but any integer number from 2 upwards is feasible) of bandpass filters 424, each one centered on a particular transmit frequency in the comb generator output.

Given that the Doppler shift from human movement will be relatively low these can be narrowband filters which will help to decrease the thermal noise in the receiver and mean that closely spaced comb frequencies can be used in the two RADAR transceivers. A single fixed-frequency local oscillator 426 is used to down-convert the signals reflected from targets in the room being monitored by the CW Doppler RADAR.

As the frequency comb generator circuits 421 provide a fixed-frequency design it should be easier to optimize its phase performance which will provide a higher level of overall sensitivity to slow moving (small Doppler shift) targets with low return signal strengths (i.e. distant targets or low reflectivity targets at close range) than for a tunable oscillator or synthesizer which is swept in frequency to steer a single leaky-wave antenna beam through the room, as shown in Fig. 2A or 2B.

In Fig. 4, single down-conversion mixers 427 with respective bandpass filters 428 are shown for each frequency. This is based upon the assumption that the in-phase (I) and quadrature (Q) information necessary to determine whether the Doppler shift is due to movement towards or away from the RADAR will be determined by digital signal processing through use of bandpass sampling and the Hilbert transform to determine the quadrature component. This removes the potential for phase and amplitude imbalances which would be unavoidable with an all analogue IQ down-conversion process. The conversion speed requirement of analogue-digital converters 429 used for bandpass sampling (which also performs the translation to baseband) is not high due to the very modest (few kHz at most) Doppler shifts that can be expected from typical human movement speeds and carrier frequencies in the ISM bands around 24, 61 and 122 GHz.

In the above or other embodiments, optically transparent materials for the waveguide could include such things as amorphous fluoropolymers. These have the advantage of low dielectric losses. The low real permittivity will result in larger physical dimensions of the dielectric waveguide than those obtained with a material with a higher real permittivity value. This larger physical size could be an advantage at higher millimeter-wave frequencies where the cost and difficulty of manufacturing a very small guide made from a higher permittivity material could be considerable.

Other optically transparent material options would include poly(methyl) methacrylate (which is commonly referred to as acrylic or acrylic glass or plexiglass) or polycarbonate. Fluorinated ethylene propylene would be an additional transparent polymer material that could be used. Various forms of glass with low dielectric loss tangent could be used as well to reduce losses in the waveguide. This would tend to imply the use of borosilicate glasses. Other possible transparent materials would be monocrystalline aluminum dioxide (“Sapphire”). The relatively high (~10) real permittivity may

advantageously lead to more beam shifting as a function of frequency change. Other options include fused quartz and exotic materials such as yttrium oxide, aluminum oxynitride spinel and magnesium aluminate spinel.

To increase the permittivity of the amorphous fluoropolymers, or indeed the other polymer materials, they could be‘loaded’ with another material (typically in the form of a fine powder) with a higher dielectric constant to create a dielectric composite material. Suitable loading materials would be such things as titanium (IV) oxide (often called titanium dioxide) which has a high dielectric permittivity as well as a high refractive index at optical wavelengths. It has an extremely bright white color and is excellent at reflecting visible light hence the loading of a transparent material with titanium dioxide could convert it into a highly reflective opaque material (depending upon the volume fraction of powder used). Other materials that could be used to Toad’ a low permittivity polymer host to make a higher permittivity composite would be silicon dioxide, aluminum dioxide, barium titanate or barium sulfate.

Highly reflective opaque material options could be polytetrafluoroetheylene (PTFE), titanium dioxide ceramic or polystyrene. Candidate materials for the thin-film metal strip grating could be electroless copper, electrodeposited copper, rolled copper, sputtered copper, evaporated copper, aluminum, aluminum-silicon-copper, gold, titanium tungsten or titanium tungsten with nitrogen‘stuffing”.

Aspect ratios from 1: 1 to 10: 1 (width:height) of the optically transparent or highly reflective opaque material of the waveguide can be used. For the dielectric waveguide placed directly onto a radio frequency (RF) ground plane, the aspect ratio determines the rate at which the antenna beam sweeps due to changes in frequency (number of degrees of change in the antenna beam direction per GHz of frequency shift). An aspect ratio of 5: 1 aspect ratio is expected to give the highest rate of antenna beam sweep as a function of frequency. This applies for all values of the real component of the dielectric permittivity of the material the waveguide is manufactured from. The single mode operating bandwidth of the dielectric waveguide will increase as the aspect ratio decreases towards 1 : 1 but the rate of antenna beam sweep will decrease.

The dielectric waveguide does not need to be placed on a ground plane but could be suspended in free space. Additionally, the dielectric waveguide may be made of two different dielectric portions, e.g., a lower dielectric portion placed directly on the RF ground plane could be chosen to have a lower dielectric constant than an upper portion of the waveguide and could be a relatively thin layer. The lower dielectric constant portion may then serve to lower the overall losses of the dielectric waveguide.

Typically, the direction in which the light is emitted from the luminaire device may also be the direction in which the antenna beam(s) are to be radiated as well. However, the directions may differ depending on the application of the luminaire device, e.g. whether it is used to generate light for a working place or for a picture at a wall or for a sculpture or other object at a specific position in a room.

To summarize, simple, low-loss dielectric waveguide leaky-wave antennas for multiple, frequency steered beams for use in CW Doppler RADAR systems integrated into luminaire devices have been described.

While the invention has been illustrated and described in detail in the drawings and foregoing description, such illustration and description are to be considered illustrative or exemplary and not restrictive. The invention is not limited to the disclosed embodiments. It may be relevant for any type of connected luminaires requiring Doppler RADAR sensors. Although the form factor appears suitable for TLED type luminaire devices, the approach can readily be adapted for more compact designs, e.g., by appropriate choice of dielectric waveguide material. Moreover, it could also be used for high data rate, point-to-point communications links.

Other variations to the disclosed embodiments can be understood and effected by those skilled in the art in practicing the claimed invention, from a study of the drawings, the disclosure and the appended claims. In the claims, the word“comprising” does not exclude other elements or steps, and the indefinite article“a” or“an” does not exclude a plurality. A single processor or other unit may fulfil the functions of several items recited in the claims. The mere fact that certain measures are recited in mutually different dependent claims does not indicate that a combination of these measures cannot be used to advantage.

The foregoing description details certain embodiments of the invention. It will be appreciated, however, that no matter how detailed the foregoing appears in the text, the invention may be practiced in many ways, and is therefore not limited to the embodiments disclosed. It should be noted that the use of particular terminology when describing certain features or aspects of the invention should not be taken to imply that the terminology is being re-defined herein to be restricted to include any specific characteristics of the features or aspects of the invention with which that terminology is associated.

A single unit or device may fulfill the functions of several items recited in the claims. The mere fact that certain measures are recited in mutually different dependent claims does not indicate that a combination of these measures cannot be used to advantage.