RANGE FROM EARTH-ORBITING SATELLITES

Contractual Origin of the Invention

The United States government has rights in this invention pursuant to Contract No. NAS7-100 between the National Aeronautics and Space Administration and the California Institute of Technology, Jet Propulsion Laboratory.

Background of the Invention

This invention relates to platform positioning, i.e., tracking and locating points relative to an earth coordi¬ nate system, and, more particularly, to a method and apparatus for deriving pseudo range from earth-orbiting, signal-transmitting satellites.

The U.S. Navy TRANSIT Navigational Satellite System comprises a number of satellites in near-earth polar orbit. The satellites transmit to user stations 150 and 400 MHz carriers on which satellite ephemeris information is modulated. The user measures the frequency of the received signal and calculates his range with respect to transmitting satellites based on the Doppler

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frequency shift observed by the user and the satellite ephemeris information. Specifically, if the positions of the observed satellite and its transmission frequency are known, the Doppler frequency shift of the signals received from the satellite at a point as it passes over the receiver in 15 minutes or less permits the determination of the location of that point in an earth-referenced coordinate system. A further descrip¬ tion of TRANSIT is given in an article by H.D. Black entitled "Satellites for Earth Surveying and Ocean

Navigating," Johns Hopkins APL Technical Digest, January- March, 1981, Volume 2, No. 1, pages 3-13.

The NAVSTAR Global Positioning System, which com¬ prises a plurality of coded signal transmitting satellites in far earth orbit, will provide, to authorized users in possession of the code, greater accuracy and more flexibility than TRANSIT. Although at present there are only six NAVSTAR satellites in orbit, it is planned that eventually there will be 18, so distributed that four satellites are visible from any point on earth. Each NAVSTAR satellite transmits a so-called Lj_ band radio signal centered about a frequncy of 1575.42 MHz and an L2 band radio signal centered about a frequency of 1227.6 MHz. The L^ band has a suppressed carrier ]_ upon which a protected information channel called P-code and a coarse acquisition channel called C/A code are modulated in double side band form. The L2 band has a suppressed carrier L2 upon which the P-code is modulated in double side band form. The P-code and C/A code channels both carry binary inform¬ ation signals at a given chip rate, i.e., 10.230 MHz in the case of the P-code, and 1.0230 MHz in the case of the C/A code. The P-code sequence is only known to

authorized users of the NAVSTAR system. With knowledge of the P-code sequence, an authorized user of NAVSTAR can derive his pseudo range relative to the satellites in his field of view by measuring the time it takes for the P-code sequence to traverse the distance between the satellites and the user, knowing that the signal travels at the speed of light. The true range of the user from a satellite equals the pseudo range plus the difference in the offset of the satellite clock and the user clock relative to a time reference such as the universal time coordinated scale. Having thus measured the pseudo range, a user can determine the satellite and user clock offsets and perform a number of tracking and location determinations termed platform positioning, by means of well-known techniques. To those not having access to the P-code sequence, the NAVSTAR satellite system cannot currently be used to determine high-precision pseudo range.

Summary of the Invention

The invention permits a user to derive his pseudo range from earth-orbiting, signal-transmitting satellites without knowledge of the code sequence of modulation carried by the signal, if any. A modulated radio frequency signal having a component at a given frequency, which is transmitted from a satellite is intercepted at a user position. The component is recovered from the intercepted signal. The phase and frequency of the component are measured. From these measurements _a nd similar measurements from other such satellites, the pseudo range of the satellite can be derived. Specifically, a fractional phase is derived from the measured phase and frequency of the intercepted signal.

A Doppler range value is also derived from the measured frequencies of the satellites. The Doppler range value is divided by the wavelength of the given frequency to produce an integer and a remainder. The integer is added to the fractional phase to produce a value propor¬ tional to the pseudo range.

From the pseudo range, a number of platform-positioning operations can be performed by means of well-known tech¬ niques — for example, point positioning of a user in an earth-referenced coordinate system, differential posi¬ tioning of a user relative to a stationary point, satellite positioning, and ionospheric calibration.

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Brief Description of the Drawings

The features of a specific embodiment of the best mo contemplated of carrying out the invention are illustrated in the drawings, in which: FIG. 1 is a diagram showing a single satellite in earth orbit for the purpose of depicting the relationships and terms used to explain the invention;

FIG. 2 is a diagram of three satellites orbiting the earth to depict the Doppler frequency shift; FIG. 3 is a schematic block diagram of a receiver incorporating principles of the invention;

FIGS. 4, 6, and 8 are sketches used to explain the invention;

FIGS. 5, 7, and 9 are schematic block diagrams of the signal processors of FIG. 3;

FIG. 10 is a schematic block diagram of the integer generator of FIGS 5, 7, and 9; and

FIG. 11 is a schematic diagram of a differential positioning system incorporating principles of the inventi

Detailed Description of the Specific Embodiment

Although the invention is applicable to other types of earth-orbiting, signal-transmitting satellite systems, it is described herein in connection with the NAVSTAR Global Positioning System, which furnishes a very advantageous signal for pseudo ranging by means of the invention.

FIG. 1 illustrates a signal-transmitting NAVSTAR satellite 10 orbiting the earth 12 along a path 14. A user on the surface of the earth is located at a point 16. In an earth-referenced coordinate system having its origin at the center of the earth 18, the position of user point 16 is R and the position of satellite 10 is p _s . The range of satellite 10 from the user is p. Point 16 lies in a horizontal plane of observation 20, which is normal to the radial line 22 passing through point 16 and center 18. As illustrated by the solid nature of path 14, satellite 10 is visible to a user at point 16, during the portion of path 14 lying on the opposite side of plane 20 from earth 12. While satellite 10 is visible, the user at point 16 is inter¬ cepting radio signals therefrom. The magnitude of the user's range Jp| from satellite 10 varies from a maximum where it rises above plane 20 to a minimum where it crosses radius 22, called herein time of closest approach (TCA), to another maximum where it sets below plane 20 (although R need not always lie in the satellite orbit plane). This relationship is shown by a curve 24. In the case of NAVSTAR, this range varies between about 20 million meters and 25 million meters. Due to Doppler frequency shift, the frequency intercepted at point 16 varies from the frequency transmitted by satellite 10, the measured frequency, f^, at point 16 being largest when satellite

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10 rises above plane 20, being smallest when satellite 10 sets below plane 20 and being the same as the trans¬ mitted frequency, fi _j , when satellite 10 is at its time of closest approach. This relationship and the Doppler frequency spread are shown by a curve 26.

FIG. 2 illustrates how the satellite position affects the frequency shift of the signal intercepted by the user

-*• at point 16. The component of the satellite velocity, V _s , lying on an imaginary line passing through the satellite and point 16, is proportional to the Doppler frequency shift. This component is proportional to the cosine , where α is the angle between the velocity vector of the satellite and the imaginary line passing through the satellite and point 16. Thus, a first satellite that has just risen above the horizontal plane of observation is at an angle c_j_, i.e., approximately

77°, and has the largest Doppler frequency shift. A second satellite at the time of closest approach has an angle c_2 of 90° and no Doppler frequency shift. third satellite midway between the time of closest approach and the horizontal plane of observation has an angle 0:3, i.e., approximately 103" and a negative Doppler frequency shift. In general, point 16 must lie on the surface of an imaginary cone whose vertex is at the satellite position with a half apex angle α. In a two-dimensional case, the cones of two satellites determine the location of point 16 and therefore two satellites are sufficiently unique to determine the location of point 16 by measuring the Doppler frequency shift at point 16. In the threedimensional case, ^' three satellites are required provided that the receiver frequency reference is identical to the frequency refernce aboard the NAVSTAR satellites.

In practicing the invention, selected frequency components of the signal transmitted from the satellites are recovered. To understand how the invention derives the pseudo range from a user to a satellite, the distance therebetween can be considered as comprising a large number of whole wavelengths, N, of one of the selected frequency component together with a fraction of this wavelength. The fraction of the wavelength called hereafter the fractional phase and designated φ , is derived by measuring the frequency, f _M , and time interval, T, between a given point of the selected com¬ ponent and a reference signal. Specifically, the measured frequency, f _M , is multiplied by the time interval T to produce a precise determination of the fractional phase, φ. The degree of precision depends on the wavelength of the selected component. The number of whole wavelengths is derived by measuring the Doppler frequency shift of a selected component from different satellites to determine a gross Doppler range value, hereafter called the Doppler range PQ, by well-known Doppler ranging techniques. The pseudo range is the sum of all the whole wavelengths between the satellite and the user and the fractional wavelength. Thus, the pseudo range can be expressed by the following equation:

p' = (c/f _τ )[f _M T + N]

where p' is the pseudo range, £ is the speed of light, f-T is the true frequency of the selected component transmit ted by the satellite, frø is the measured frequency of the selected component intercepted by the user reflecting the Doppler frequency shift, T is the time interval, and N

is the number of whole wavelengths.

By dividing the gross Doppler range value, PQ, by the wavelength of the selected component, λ , a whole number of wavelengths, N, plus a remainder are produced. If the measurements for deriving Doppler range, P _Q , are made to provide an accuracy of better than one-half of a wavelength, the whole number of wavelengths, N, is precise, leaving all the uncer¬ tainty, i.e., error, in the remainder. In general, the remainder is discarded and in its place the frac¬ tional phase, φ , which is very precise, is added to the whole number of wavelengths N to provide an accurate value of pseudo range, p ¹ , the extent of accuracy depend¬ ing upon the wavelength of the selected component, λ . (Note that some measurements must either be discarded or the quotient must be increased to the next whole number to produce the whole number of wavelengths, N, as discussed below in connection with FIG. 4.)

FIG. 3 illustrates a receiver for recovering selected components of the signal transmitted by a NAVSTAR satellit For convenience, the selected frequency components of the signal transmitted from the NAVSTAR satellites and their characteristics are set forth in the following table:

Selected Frequency Effective Recovered Doppler Component (MHz) Wavelength (m) Spread

^ carrier 1575.42 .095 + 8.3 kHz

L2 carrier 1227.6 .122 + 6.5 kHz

P-code chip rate 10.230 29.3 + 27 Hz

C/A-code chip rate 1.0230 293 + 2.7 Hz

The signal transmitted from a NAVSTAR satellite is intercepted by an antenna 30. As previously discussed, this signal comprises an ^_ band with a suppressed carrier _ upon which a P-code channel and a C/A code channel are modulated in double side band form and an 2 band with a suppressed carrier L2 upon which the P-code is modulated in double side band form. The underlined numbers in FIG. 3 represent the frequency at the designated locations, in MHz unless otherwise noted. The numbers in parentheses represent the pass bands of the filters and the time delays of the delay lines with which they are associated. Antenna 30 is connected by a band pass filter 32 and an amplifier 34 to the input of a power divider 36, which separates the L _j _ band and the L2 band. A frequency reference 38 comprises a stable atomic clock such as a Hewlett- Packard HP5065A, synchronized to the frequency reference of the satellite, although less sophisticated crystal oscillators such as a Hewlett-Packard HP105 can also perform this function. Frequency reference 38 is coupled to a frequency synthesizer 40, which generates eight different frequencies for use in the receiver by means of frequency multipliers and dividers. Thus, the eight frequencies generated by synthesizer 40, which are shown eminating therefrom are precisely synchronized to frequency reference 38. Frequency reference 38 is also coupled to a clock 41, which has an output where Uni¬ versal Time Coordinated (UTC) clock time is available and an output where one pulse per second (IPPS) is generated. From one output of power divider 36, the L _j _ band is coupled through a filter 42 to a mixer 44 where it is down converted by one of the tones from synthesizer 40. The output of mixer 44 is coupled by an amplifier 46 to the input of a power divider 48.

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1 The L _] _ carrier component is recovered by connecting two of the outputs of power divider 48 to a mixer 50. The output of mixer 50 is coupled by a filter 52 to a mixer 54, where it is down converted by a tone from "5 synthesizer 40. The output of filter 52 is centered at 70.84 MHz and has twice the Doppler frequency spread that occurs on the L ₁ carrier and thus an effective recovered wavelength of .095 meters. A low frequency narrow band sine wave signal, exhibiting the Doppler 10 frequency shift and fractional phase of the ]_ carrier, is coupled from mixer 54 to an L^ carrier signal processor 56. The zero Doppler frequency shift condition is exhibited as a 10 kHz sine wave.

The P-code chip rate component is recovered by con- 15 necting one output of power divider 48 directly to a mixer 58 and another output of power divider 48 to mixer 58 through a delay line 60 that introduces a time delay of one-half of the period of the P-code chip rate. The output of mixer 58 is coupled by a filter 20 62 to a mixer 64, where it is down converted by a tone from synthesizer 40. The output of filter 62 is a sine wave at the P-code chip rate. A low-frequency, narrow-band sine wave signal, exhibiting the Doppler frequency shift and fractional phase of the P-code chip 5 rate component, is coupled from mixer 64 to a P-code chip rate signal processor 66.

The C/A code chip rate component is recovered by connecting another output of power divider 48 directly to a mixer 68 and yet another output of power divider 0 48 to mixer 68 through a delay line 70, which introduces a delay equal to one-half the period of the C/A code chip rate. The output of mixer 68 is coupled by a filter 72 to a mixer 74, where it is down converted by a tone from synthesizer 40. The output of filter 72 5 is a sine wave at the .C/A code chip rate. A low-frequency,

14633/LTR -12- narrow-band sine wave signal exhibiting the Doppler frequency shift and fractional phase of the C/A code chip rate component is coupled from mixer 74 to a C/A code chip rate signal processor 76. From the other output of power divider 36, the 2 band is coupled through a filter 78 to a mixer 80, where it is down converted by one of the tones from synthesizer 40. The output of mixer 80 is coupled by an amplifier 82 to the input of a power divider 84. The L2 carrier component is recovered by connecting two of the outputs of power divider 84 to a mixer 86. The output of mixer 86 is coupled by a filter 88 to a mixer 90, where it is down converted by a tone from synthesizer 40. The output of filter 88 is centered at 70.84 MHz and has twice the Doppler frequency spread that occurs on the 2 carrier and thus the effective re¬ covered wavelength is .122 meters. A low frequency, narrow-band sine wave signal exhibiting the Doppler frequency shift and fractional phase of the 2 carrier is coupled from mixer 90 to an L2 carrier signal processor 92. The zero Doppler frequency condition is exhibited by a sine wave at 10 kHz.

The 2 channel P-code chip rate component is recovered by connecting one output of power divider 84 directly to a mixer 94 and another output of power divider 84 to mixer 94 through a delay line 96 that introduces a time delay of one-half of the period of the P-code chip rate. The output of mixer 94 is coupled by a filter 98 to a mixer 100, where it is down converted by a tone from synthesizer 40. The output of filter 98 is a sine wave at the P-code chip rate. A low- frequency, narrow-band sine wave signal, exhibiting the Doppler frequency shift and fractional phase of the P-code chip rate component, is coupled from mixer 100 to a P-code chip rate signal processor 102.

FIG. 4 depicts the distance, i.e., range, between a satellite 104 and a user 106 for the C/A code chip rate component. A large number, N _c / _A , of whole wavelengths, ^Λ c/ _A » of the C/A code component and a fractional phase, Φc/A' extend between satellite 104 and user 106. The fractional phase, ΦC/ » ^ ^S measured and the number of wavelengths, N / * ^ ^S determined by deriving a gross value of Doppler range, P _Q . In this case, the gross value of Doppler range, P r , is derived from the Doppler frequency shift of the P-code chip rate component because, assuming an amplitude signal- to-noise ratio of 10:1, this can readily be done to an accuracy of 50 meters or less, i.e., about one-sixth of a wavelength, ^Λ C/A* '^ ^ιe uncertainty i.e., possible error, of the gross value of Doppler range, PQ, is depicted by the parentheses 107, meaning that the measured gross value of the Doppler range p^ could fall anywhere within the parentheses. The gross value of the Doppler range, PQ, is divided by the wavelength, Λc/A' of the C/A code chip rate component to produce a quotient comprising a whole number plus a remainder. In general, the remainder is discarded and the whole number, which is treated as the whole number of wave¬ lengths, _c / _A , is added to the fractional phase, Φc/ ' ^to produce an accurate value of pseudo range ^P 'c/A* If ^tne remainder is larger than a given value depending on the signal-to-noise ratio, e.g., five- sixths of a wavelength, ^Λ c/ , ^{for an} amplitude signal- to-noise ratio of 10:1 and the fractional phase, Φc/A' is smaller than a given value, e.g. , one-sixth of a wavelength, λ _c / _A , for an amplitude signal-to-noise ratio of 10:1, this particular measurement of the gross value of the Doppler range must either be disregarded or one must be added to the whole number of the quotient

to produce the whole number of wavelengths, ^ / ' to be added to the fractional phase, ΦC/A _. » ^to produce an accurate value of pseudo range, P'C/A* With the assumed amplitude signal-to-noise ratio ratio of 10:1, the pseudo range value, P'c/A' ^{can be} determined to an accuracy of 5 meters or better.

FIG. 5 shows signal processor 76, which performs the operations described in connection with FIG. 4. The output of mixer 74 (FIG. 3) is coupled to a phase detector 108 and a frequency counter 110. The phase of the C/A code chip rate component relative to pulses occurring at one pulse per second generated by clock 41 is determined by phase detector 108. The frequency of this component is determined by frequency counter 110 at intervals of IPPS under the control of clock 41. Phase detector 108 and frequency counter 110 are connected to a multiplier 112, which produces the fractional phase, Q/A' ^{t ts out} P ^ut _' The output of mixer 64 (FIG. 3) is connected to a frequency counter 114. The output of frequency counter 114 is connected to a Doppler position finder 116 to which UTC clock time is also applied from clock 41. Position finder 116 could comprise commercially available equipment such as Magnavox Model 1502 adapted for data processing at 10.23 MHz instead of 400 MHz using principles described in "Satellite Surveying," by G. J. Hoar, Magnavox Document MX-TM-3346-81, No. 10058, February, 1982. Position finder 116 determines the gross value of Doppler range, pjj. The output of range finder 116 is connected o an integer generator 118, which produces the value of the whole number of wavelengths, ^Λ c/ _A « The outputs of multiplier 112 and integer generator are applied to an adder circuit 120 and the output of adder circuit 120 is connected to a multiplier 122, where the value

of the sum of the fractional phase, ΦC/A» P ^{1US tϊιe} whole number of wavelengths, N _c / _A , is multiplied by the speed of light, £, divided by the frequency, f _c / _A , of the C/A code chip rate component without Doppler shift.

FIG. 6 depicts the distance, i.e., range, between satellite 104 and user 106 for the P-code chip rate component. A large number, Np, of whole wavelengths, λp, of the P-code component and a fractional phase, p» ^extend between satellite 104 and user 106. The fractional phase, φp, is measured and the number of wavelengths, Np, is determined by using the already derived pseudo range value, p'c/A _* which is accurate to less than one-sixth of a wavelength, λp, of the P-code chip rate component. The uncertainty, i.e., possible error, of the pseudo range, p'c/A» ^ ^s depicted by the parentheses 109, meaning that it could fall anywhere within the parentheses. The value of the pseudo range, p'c/A' ^ ^S divided by the wavelength, λp, of the P-code chip rate component to produce a quotient comprising a whole number of wavelengths, Np, plus a remainder. In general, the remainder is discarded the whole number, which is treated as the whole number of wavelengths, Np, is added to the fractional phase, ΦP' ^to Produce an accurate value of pseudo range, p'p. If the remainder is larger than a given value depending on the signal-to-noise ratio, e.g., five-sixths of a wavelength, λp, for an amplitude signal-to-noise ratio of 10:1 and the fractional phase φp is smaller than a given value, e.g., one-sixth of a wavelength, λp, for an amplitude signal-to-noise ratio of 10:1, the gross value of the Doppler range must either be disregarded or one must be added to the whole number of the quotient to produce the whole number of wavelengths, Np, to be

added to the fractional phase, φ _p , to produce an accurate value of pseudo range, p'p. With the assumed amplitude signal-to-noise ratio of 10:1, the pseudo range value, p'p can be determined to an accuracy of 30 cm, or better.

FIG. 7 shows signal processor 66 or 102, which performs the operations described in connection with FIG. 6. The output of mixer 64 or 100 (FIG. 3) is coupled to a phase detector 124 and to frequency counter 114 (FIG. 5). The phase of the P-code chip rate component relative to pulses occurring at one pulse per second generated by clock 41 is determined by phase detector 124. The frequency of this component is determined by frequency counter 114 at intervals of IPPS under the control of clock 41. Phase detector 124 and frequency counter 114 are connected to a multiplier 126, which produces the fractional phase, Φp, at its output. The output of multiplier 122 (FIG. 5) is connected to an integer generator 128, which produces the value of the whole number of wavelengths, λp. The outputs of multiplier 126 and integer generator 128 are applied to an adder circuit 130 and the output of adder circuit 130 is connected to a multiplier 132, where the value of the sum of the fractional phase, φp, plus a whole number of wavelengths, Np, is multiplied by the speed of light, £, divided by the frequency, fp, of the P-code chip rate component without Doppler shift.

FIG. 8 depicts the distance, i.e., range, between satellite 104 and user 106 for the L-carrier compo- nent. A large number Nj_. of whole wavelengths, λ _j ^, of the L-carrier component and a fractional phase, φ^, extend between satellite 104 and user 106. The fractional phase, Φ _L , is measured and the number of wavelengths, Np, is determined by using the already derived pseudo range

value, P'p, which is accurate to less than one-sixth of a wavelength, λ _L , of the L-carrier component. The uncertainty, i.e., possible error, of the pseudo range, p' _p , range is depicted by the parentheses 111, meaning that it could fall anywhere within the parentheses; however, a relatively high signal-to-noise ratio of 230:1 is required. The value of the pseudo range, p'p, is divided by the wavelength, λ^, of the L-carrier component to produce a quotient comprising a whole number of wavelengths, Nj _j , plus a remainder. In general, the remainder is discarded and the whole number, which is treated as the whole number of wave¬ lengths, L, is added to the fractional phase, L, to produce an accurate value of pseudo range, 'L _* if the remainder is larger than a given value depending on the signal-to-noise ratio, e.g., five-sixths of a wavelength, λ _L , for an amplitude signal-to-noise ratio of 10:1 and the fractional phase, Φ^, is smaller than a given value, e.g. , one-sixth of a wavelength, λ _L , for an amplitude signal-to-noise ratio of 10:1, the gross value of the Doppler range must either be disregarded or one must be added to the whole number of the quotient to produce the whole number of wavelengths, N _L , to be added to the fractional, Φ _L , to produce an accurate value of pseudo range, λ'L. With the as¬ sumed amplitude signal-to-noise ratio of 10:1, the pseudo range value, P'L* ^{an De} determined to an accuracy of 2 mm or better.

FIG. 9 shows signal processor 56 or 92, which per- forms the operations described in connection with FIG. 8, The output of mixer 54 or 90 (FIG. 3) is coupled to a phase detector 134 and a frequency counter 136. The phase of the L-carrier component relative to pulses occurring at. one pulse per second generated by clock

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41 is determined by a phase detector 134. The frequency of this component is determined by a frequency counter 136 at intervals of IPPS under the control of clock 41. Phase detector 134 and frequency counter 136 are connected to a multiplier 138, which produces the fractional phase, ΦL, at its output. The output of multiplier 132 (FIG. 7) is connected to an integer generator 140, which produces the value of the whole number of wavelengths, λp. The output of multiplier 138 and integer generator 140 are applied to an adder circuit 142 and the output of adder circuit 142 is connected to a multiplier 144, where the value of the sum of the fractional phase, φ^, plus a whole number of wavelengths, T _^ , is multiplied by the speed of light, C, divided by the frequency, fj., of the L-carrier component without Doppler shift.

Referring once again to FIGS. 4, 6, and 8, the fractional phase, φ , is measured accurately in each case, but there is an ambiguity, namely, the number of whole cycles, N, in addition to the fractional phase, φ, that makes up the total path length between satellite 104 and user 106, i.e., the pseudo range, p'. In the case of each selected component, the ambiguity is resolved by a measurement which, although not as accurate as the fractional phase measurement, is accurate to within less than about one-sixth of a wavelength of the selected component, thereby permitting precise determination of the number of wavelengths, N. Thus, in the described example, the ambiguity of the C/A chip rate code component is resolved by a Doppler position measurement, the ambiguity of the P-code chip rate component is resolved by using the pseudo range derived from the measurements in connection with the C/A chip rate component, and the ambiguity of the L carrier component is resolved

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by using the pseudo range derived from the measurement in connection with the P-code chip rate component.

By way of example, the described phase detectors could be so-called time interval counters comprising a clock pulse source (e.g., at a frequency of 5 MHz) that feeds pulses to a counter. The counter starts counting the clock pulses when a 1 PPS pulse from clock 41 is applied thereto and stops counting the clock pulses at the positive going zero crossing of a component whose phase is being detected. Similarly, the described frequency detectors could comprise a counter that counts the number of cycles of the selected component occurring during a given time period, e.g., one second, as determined by clock 41.

FIG. 10 shows integer generator 118 in more detail. (Integer generators 128 and 140 are identical thereto.) A divider 170 to which the output of Doppler position finder 116 (multiplier 122 in FIG. 7, and multipilier 132 in FIG. 9) is connected divides the input value by the value of the wavelength, λ _c / _A (λp in FIG. 7, and λ _L in FIG. 9). The output of divider 170 is directly connected to adder ciruit 120 (132 in FIG. 7, and 144 in FIG. 9) via a transmission gate 172 and an isolation gate 174. The output of divider 170 is also coupled through an add one circuit 176, a transmission gate 178, and isolation gate 174 to adder circuit 120 (132 in FIG. 7, and 144 in FIG. 9). The same signal applied to divider 170 is also applied to a threshold detector 180. The output of multiplier 112 (126 in FIG. 7, and 138 in FIG. 9) is applied to a threshold detector 182. The outputs of threshold detectors 180 and 182 are coupled to the inputs of an AND gate 184. The output of AND gate 184 is directly connected to the control terminal of transmission gate 172 and is

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connected via an inverter 186 to the control terminal of transmission gate 178. Threshold detectors 180 and 182, AND gate 184, and inverter 186 are digital control circuits, while divider 170, add one circuit 176, transmission gate 172 and 178, and isolation gate 174 are analog or digital signal transmission circuits. Threshold detector 180 produces a high binary value when the signal applied to its input exceeds a threshold representative of a quotient with a remainder larger than five-sixth of a wavelength, ^Λ C/A _* Threshold detector 182 produces a high binary value when the signal applied to its input is representative of a fractional phase less than one-sixth of a wavelength, λ _c / _A . When both these conditions are met, AND gate 184 produces a high binary value, thereby opening transmission gate 178 to transmit therethrough a value representative of the whole number of the quotient from divider 170 plus one. Otherwise, trans¬ mission gate 172 couples a value representative of the whole number of the quotient from divider 17Q. The signals and components in FIGS. 5, 7, and 9 could either be analog or digital. In the latter case, sequential timing circuits not shown would be provided. Alternatively, these operations could be carried out on a programmed digital computer. In the case of the selected components in the L2 band, which does not have a C/A code channel, the Doppler range is derived on the basis of the Doppler frequency shift of the P-code component to sufficient ac¬ curacy to permit the derivation of the whole number Np of wavelengths, λ _p# rather than, ^as described in con¬ nection with FIGS. 4 and 5. This whole number of wavelengths, Np, is added to the fractional phase, Φp, of the P-code chip rate component rather than the C/A code chip rate component which will require a higher

signal-to-noise ratio to resolve one-sixth of a wave¬ length, λp (5 m), rather than the requirements to position to 50 m to resolve C/A code phase ambiguities.

In processing signals from a plurality of satellites with a directional antenna, only one set of signal proces¬ sors such as processors 56, 66, 76, 92, and 102, need be provided because only the signal from one satellite at a time is being received. If an omnidirectional antenna is employed, then a separate set of signal processors must be provided for each of the satellites, e.g., four, in view at one time. The signals received from the different satellites are distinguishable from each other by their difference in frequency due to Doppler frequency shift. They could be separated from each other by, for example, by a comb filter.

FIG. 11 depicts an application of the invention to differential positioning, i.e., the positioning of a user relative to a stationary point in an earth-referenced coordinate system. A NAVSTAR satellite 150 transmits signals to first and second stationary earth stations. The range from satellite 150 to the first station is Pl and to the second station is p 2• The unit vector from the first station to the satellite is S. The baseline vector from the first station to the second

-_- station is B, the quantity to be derived. At the first stationary station, the satellite signal is intercepted by an antenna 152, which is connected to a receiver and signal processor 154. Receiver and signal processor 154 derives frequency and time interval values for a selected component of the intercepted signal and sends this information to a remote computer 156 via a transmission line 158, although by time tag¬ ging these values the information can be delayed by any amount in its transmission. Similarly, at the second

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user station, the satellite signal is intercepted by an antenna 160, which is connected to a receiver and signal processor 162. Receiver and signal processor 162 derives frequency and time interval values for a selected component of the intercepted signal and sends this information to computer 156 via a transmission line 164. The known quantities S^, R, and p _Sj _ are also supplied to computer 156 which by means of a computer program, a listing of which is attached hereto as Appendix A, generates the baseline vector B. For each of the four satellites in view, computer 156 solves the following equation:

Δp = - S * B + <5 + c(ΔTui - ΔTu2-

+ cT^ = c/f(f ₂ T ₂ - fχT _χ ) + c/f(N ₂ - N _χ )

where ΔTyi is the offset of the clock at the first station, Δ Tu2 i the offset of the clock at the second station, T^M is the differential atmospheric transmission media error for the signals arriving at each station, f _j _ is the measured frequency of the selected component at the first station, T]_ is the measured time interval of the selected component at the first station, f2, is the measured frequency of the selected component at the second station, and T2 is the measured time interval of the selected component of the second station, and δ = Δp - B /2p^« Receiver and signal processors 154 and 162 are constructed in the manner described in connection with FIG. 3 together with the frequency counters and phase detectors as described in connection with FIGS. 5, 7, and 9.

The described embodiment of the invention is only con sidered to be preferred and illustrative of the inventive concept; the scope of the invention is not to be restricte

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to such embodiment. Various and numerous other arrange¬ ments may be devised by one skilled in the art without departing from the spirit and scope of this invention. One aspect of the invention is to derive pseudo range by Doppler positioning alone utilizing concurrent reception of modulated radio signals from plural satellites together with recovery of a component therefrom for comparison with the frequency of the transmitted component. Such comparison permits the position of the user to be determined as described above. Attached hereto as Appendix B is further description of the invention, the disclosure of which is fully incorporated into this application by reference.

*Λ-W.

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