FIELD OF THE INVENTION The field of invention relates generally to wireless communications; and more specifically to calibrating a wireless device placed in the field.

BACKGROUND Super Heterodyne and Frequency Shift Keved (FSK) Modulation/Demodulation Figure 1 shows a portion 106 of a receiving device 166 referred to as a demodulator. A demodulator 106 provides a signal (commonly referred to as a baseband signal b (t) in various applications) that is representative of the information being sent from a transmitting device 165 to a receiving device 166.

The demodulator 106 extracts (i. e., demodulates) the baseband signal b (t) from a high frequency wireless signal that"carries"the baseband signal b (t) through the medium (e. g., airspace) separating the transmitting and receiving devices 165,166.

The particular demodulator 106 example of Figure 1 is designed according to: 1) a demodulation approach that is commonly referred to as super heterodyne detection (hereinafter referred to as a heterodyne detection for simplicity); and 3) a modulation/demodulation scheme referred to as Frequency Shift Keying (FSK). The industry standard referred to as"BLUETOOTH" (the requirements of which may be found in"Specification of the Bluetooth System", Core v. l. OB, 12/1/99, and published by the Bluetooth Special Interest Group (SIG)) can apply to both of these approaches and, accordingly, will be used below as a basis for reviewing the following background material.

Heterodyne detection is normally used when dedicated channels are allocated within a range of frequencies 111 (where a range of frequencies may also be referred to as a"band"111). For BLUETOOTH applications within the United States, 89 channels 110l, 1102, 1103,... 11079 are carried within a 2.400 GHz to 2.482 GHz band 111. Each of the 79 channels are approximately 1 Mhz wide and are centered at frequencies 1 Mhz apart.

The first channel 110, is centered at 2.402 Ghz, the second channel 1102 is centered at 2.403 Ghz, the third channel 1103 is centered at 2.404 Ghz, etc., and the seventy ninth channel 11079 is centered at 2.480 Ghz. The heterodyne demodulator 106 accurately receives a single channel while providing good suppression of the other channels present within the band 111. For example, if channel 1102 is the channel to be received, the baseband signal b (t) within channel 1102 will be presented while the baseband signals carried by channels 1101, and 1103 through 11079 will be suppressed.

An FSK modulation/demodulation approach is commonly used to transmit digital data over a wireless system. An example of an FSK modulation approach is shown in Figure 1. A transmitting modulator 105 within a transmitting device 165 modulates a baseband signal at a carrier frequency f « ; er into an antennae 102. That is (referring to the frequency domain representation 150 of the signal launched into the antennae 102) if the baseband signal corresponds to a first logic value (e. g.,"1"), the signal 150 has a frequency of carner+ o. If the data to be transmitted corresponds to a second logic value (e. g., "0"), the signal has a frequency of fcamer'o.

Thus, the signal launched into the antennae 102 alternates between frequencies of fcarrier+ fo and fcar"er-fo depending on the value of the data being transmitted. Note that in actual practice the transmitted signal 150 may have a profile 151 that is distributed over a range of frequencies in order to prevent large, instantaneous changes in frequency. The carrier frequency fcamer corresponds to the particular wireless channel that the digital information is being transmitted within. For example, within the BLUETOOTH wireless system, camer corresponds to 2.402 Ghz for the first channel 1101. The difference between the carrier frequency and the frequency used to represent a logical value is referred to as the deviation frequency fo.

Referring now to the heterodyne demodulator 106, note that the signal received by antennae 103, may contain not only every channel within the frequency band of interest 111, but also extraneous signals (e. g., AM and FM radio stations, TV stations, etc.) outside the frequency band 111. The extraneous signals are filtered by filter 113 such that only the frequency band of interest 111 is passed. The filter 113 output signal is then amplified by an amplifier 114.

The amplified signal is directed to a first mixer 116 and a second mixer 117. A pair of downconversion signals dl (t), d2 (t) that are 90° out of phase with respect to each other are generated. A first downconversion signal dl (t) is directed to the first mixer 116 and a second downconversion signal d2 (t) is directed to the second mixer 117. Each mixer multiples its pair of input signals to produce a mixer output signal. Note that the transmitting modulator 105 may also have dual out of phase signals that are not shown in Figure 1 for simplicity.

Transmitting a pair of signals that are 90° out of phase with respect to one another conserves airborne frequency space by a technique referred to in the art as single sideband transmission.

The frequency fdown of both downconversion signals dl (t), d2 (t) is designed to be fcarrier - fIF. The difference between the downconversion frequency fdo,,, and the carrier frequency f « ; er is referred to as the intermediate frequency f, F. Because it is easier to design filters 118a, b and 127a, b that operate around the intermediate frequency, designing the downconversion that occurs at mixers 116,117 to have an output term at the intermediate frequency f,, enhances channel isolation.

The mixer 117 output signal may be approximately expressed as kbFSK (t) cos (27T f t) cos (2E fdownt). Eqn. 1 Note that Equation 1 is equal to kbFsK (t) [cos (2# (fcarrer - fdown) t) +cos (27r (f+f) t)] Eqn. 2 which is also equal to kbFSK (t) cos (27r fiat) + kbFSK (t) cos (27r (fcarrier + fdown)t) Eqn. 3 using known mathematical relationships. The bF5K (t) term represents a frequency shift keyed form of the baseband signal (e. g., a signal that alternates in frequency between +fo for a logical"1"and-fo for a logical"0"). The constant k is related to the signal strength of the received signal and the amplification of amplifier 114. For approximately equal transmission powers, signals received from a nearby transmitting device are apt to have a large k value while signals received from a distant transmitting device are apt to have a small k value.

Equation 3 may be viewed as having two terms: a lower frequency term expressed by kbFsK (t) cos (27T fiat) and a higher frequency term expressed by kbF5K (t) cos (2# (fcarrier+fdowl) t) Filter 118b filters away the high frequency term leaving the lower frequency term kbFSx (t) cos (27r flFt) to be presented at input 119 of amplification stage 125. Note that, in an analogous fashion, a signal kbpsK (t) sin (2sr f, Ft) is presented at the input 126 of amplification stage 170.

Amplification stage 125 has sufficient amplification to clip the mixer 117 output signal. Filter 127b filters away higher frequency harmonics from the clipping performed by amplification stage 125. Thus, amplification stage 125 and filter 127b act to produce a sinusoidal-like waveform having approximately uniform amplitude for any received signal regardless of the distance (e. g., k factor) between the transmitting device and the receiving device.

After filter 127, a signal s (t) corresponding to AbFsK (t) cos (27 fiat) is presented to the frequency to voltage converter 128 input 129 (where A reflects the uniform amplitude discussed above). The spectral content S (f) of the signal s (t) at the frequency to voltage converter 128 input 129 is shown at Figure 1. The signal s (t) alternates between a frequency of fn + fo (for a logical value of"1") and a frequency of flF-fo (for a logical value of"0"). The spectral content S (0 of the signal s (t) at the frequency to voltage converter 128 input 129 is mapped against the transfer function 160 of the frequency to voltage converter 128 in order to reproduce the baseband signal b (t) at the demodulator output.

Frequency Synthesis Referring back to the pair of downconversion signals dl (t), d2 (t) that are directed to mixers 116,117, recall that the downconversion signals dl (t) and d2 (t) should have a downconversion frequency fdo,, n equal to fcarrl fIF for each of the channels 1101 through 11079. For example, for an intermediate frequency flF of 3Mhz, the frequency synthesizer 140 is responsible for generating a frequency of 2.399 Ghz in order to receive the first channel 1101 (i. e, f,., i,-fF= 2.402-0.003 Ghz = 2.399Ghz); a frequency of 2.400 Ghz in order to receive the second channel 1102 ; a frequency of 2.401 Ghz in order to receive the third channel 1103 ;... etc., and a frequency of 2.477 Ghz in order to receive the 79i channel 11079. A channel select input 141 presents an indication of the desired channel to the frequency synthesizer 140.

Both the transmitting device 165 and the receiving device 166 typically have a frequency synthesizer. A frequency synthesizer 140 is shown in the receiving device 166 (but not the transmitting device 165 for simplicity).

Frequency synthesizers typically create their output signals by multiplying a reference frequency (such as the frequency of a local oscillator). As seen in Figure 1, frequency synthesizer 140 multiplies the frequency of local oscillator 142 to produce downconversion signals dl (t) and d2 (t). For example, for a local oscillator 142 reference frequency of 13.000 MHz, frequency synthesizer 140 should have a multiplication factor of 184.53846 to produce downconversion signals dl (t), d2 (t) used to receive the first channel 110, (i. e., 184.53846 x 13. = 2.399 GHz).

A problem with wireless technology involves deviation from the "designed for"carrier fc er and/or downconversion fdown frequencies (e. g., from non zero tolerances associated with the local oscillator 140 reference frequency).

As either (or both) of the carrier and/or downconversion frequencies deviate from their"designed for"values, offsets may be observed in the baseband signal b (t) at the demodulator 106 output.

Figure 2a shows a baseband signal 250 if the carrier and downconverting frequencies are ideal. As discussed above, the spectral content 253 of the signals produced by filters 127a, b will be centered at the intermediate frequency f,,.

Since the origin 250 of the frequency to voltage converter transfer curve 260 is centered at the intermediate frequency fjp, the output signal 250 has no offset (e. g., has an offset positioned at O. Ovolts) Errors in the carrier and/or downconversion frequency, however, will cause the spectral content of the signals produced by filters 127a, b to be centered at an offset 254 from the intermediate frequency fjp. That is, because fiv in equation 3 corresponds to fcarrier - fdown, if either fcarrier or fdown (or both) are in error the value of fjpin equation 3 does not correspond to the designed for fiv value (e. g., 3 Mhz) that is centered at the origin of the transfer curve 260. As such, the baseband signal 255 will have an offset 256 with respect to O. Ovolts.

Drifting of Synthesized Frequencies and Calibration of the Deviation Frequency (fo) A problem with wireless technology involves optimization of the deviation frequency fo in light of"drifting", after the transmitting and receiving devices 165,166 have been manufactured, of the synthesized carrier and downconversion frequencies fcarr, er, fdOw Typically, the deviation frequency fo is measured and/or calibrated during the manufacture of a wireless device such that transmitting devices shipped into the field transmit ls and Os at frequencies of flF +/-fo within an expected, allowable range.

However, after devices are shipped into the field, the carrier and/or downconversion frequencies fcarr, er, fdO"%, may drift (e. g., as a result of aging, temperature, etc. of a frequency synthesizer's crystal oscillator). The drifting of these frequencies can cause a drifting offset 254 in the intermediate frequency f as seen in Figure 2b. As a result, the manufactured setting of the deviation frequency fo may cause the flF +/-fo frequencies to deviate toward or beyond the limits of their allowable range. In still other cases, the deviation frequency fo may separately drift on its own (i. e., whether or not the intermediate frequency f Fis or is not drifting). As a consequence, the performance of wireless links implemented with the device may begin to degrade or fail.

SUMMARY OF INVENTION A method that comprises demodulating a received wireless signal with a demodulator. Then, tapping a tapped receive signal from the demodulator.

Then, measuring a frequency of the tapped receive signal.

BRIEF DESCRIPTION OF THE DRAWINGS The present invention is illustrated by way of example, and not limitation, in the Figures of the accompanying drawings in which: Figure 1 shows a wireless transmitting device and a demodulator design associated with a wireless receiving device; Figure 2a shows a demodulator output signal without an offset; Figure 2b shows a demodulator output signal with an offset; Figure 3a shows a technique for measuring the deviation frequency of a wireless transmitting device; Figure 3b shows a wireless receiving device design that measures the deviation frequency of a transmitting device; Figure 4 shows a portion of a possible embodiment for the design of Figure 3b; Figure 5 shows an approach for measuring the frequency of received signal; Figure 6a shows an embodiment for a frequency measurement circuit; Figure 6b shows an alternative embodiment for a frequency measurement circuit; Figure 7 shows a frequency synthesizer and Voltage Controlled Oscillator (VCO) transfer curve associated with a transmitting device; Figure 8 shows a VCO, a capacitance vs. voltage curve for a VCO and frequency vs. voltage curve for a VCO; Figure 9 shows a technique for determining the deviation frequency adjustment to be made to a plurality of wireless channels; and Figure 10 shows measured frequencies for a wireless link.

DETAILED DESCRIPTION A method is described that involves demodulating a received wireless signal with a demodulator. Then, tapping a tapped receive signal from the demodulator. Then, measuring a frequency of the tapped receive signal. An apparatus is described that includes a wireless receiving device having a demodulator where the demodulator has a tapped receive signal output. The tapped receive signal output is coupled to a frequency measurement circuit input.

Recall from the background discussion that drifting of the carrier fermer/ downconversion fdO and/or deviation fo frequencies, in combination or separately, after a wireless device has been shipped into the field, may cause the performance of a wireless link that employs the device to degrade or fail. A solution involves effectively measuring one or more of these frequencies, after devices have been shipped into the field, so that they may be calibrated.

For example, in a case where the deviation frequency fo is drifting (but the intermediate frequency fi ils not drifting or is assumed not to be drifting), the transmitted deviation frequency fo may be measured and calibrated in the field.

The following discussion provides a description of such an approach.

Afterward, additional cases (e. g., where a drifting intermediate frequency fn is accounted for) and approaches that handle them are discussed.

Referring to Figure 3a, a wireless link is formed between a wireless transmitting device 365 (or simply"transmitting device") and a wireless receiving device 366 (or simply"receiving device"). A wireless transmitting device sends information in a wireless fashion to a wireless receiving device. A wireless receiving device receives the information sent by the wireless transmitting device.

The transmitting device 365 sends, with a transmit signal 301, a tone frequency corresponding to fcarner-fo (e. g., a logical 0) and/or a tone frequency corresponding to fc,, ,, + fo (e. g., a logical 1). The receiving device 366 measures the downconverted tone frequency flF-fo and/or flF + fo associated with the received transmit signal 301. The receiving device 366 then sends to the transmitting device 365, in a return message 307, the deviation frequency fo of the transmitting device 365. Upon receiving this information, the transmitting device 366 adjusts its deviation frequency fo.

Figure 3b shows an embodiment of a design that may be used by a receiving device 366 to measure received, downconverted tone frequencies of f,, -fo and/or f¢ + fo. Demodulator Demodulator corresponds corresponds to demodulator, demodulator, such demodulator 106 of Figure 1, that demodulates a wireless transmit signal 301. A frequency measurement circuit 310 receives a signal 311 that is tapped from the demodulator 306 (e. g., between downconverting mixers 116,117 and frequency to voltage converter 128 referring briefly back to Figure 1).

The frequency measurement circuit 310 measures the tone frequency of the tapped receive signal 311 in order to determine the deviation frequency fo.

In various embodiments, the demodulator 306 location where a signal 311 is tapped for presentation to the frequency measurement circuit 310 allows for easy and/or accurate frequency measurement. A discussion of some of the possible embodiments for the form of the tapped receive signal 311 follow.

Recall from the discussion in the background pertaining to the demodulator 106, that amplifiers 125,170 clip the downconverted, received signal. Figure 4 shows an example in greater detail. Signal 401 corresponds to the input signal presented to the input of amplification stage 425 (e. g., which corresponds to amplification stage 125 of Figure 1), the output signal 411 of the amplification stage 425 appears more like a digital signal than a sinusoidal waveform.

Typically, amplification stage 425 clips its output waveform 411 to its power rails 426,427 as seen in Figure 4. Amplification stage 425 may also be referred to as a limiter because its gain is sufficiently large to clip most all received signals, regardless of its associated k factor (referring briefly to equations 1 through 3 and the related discussion in the background). As such, an approximately uniform received signal amplitude is created for most all received signals, regardless of the distance separating the receiving device and the transmitting device. A limiter output signal 411 may be used to measure the offset frequency fo as described in more detail below.

It should be understood that it is possible to develop demodulator designs that are: 1) different than the particular design 106 of Figure 1; and 2) employ a limiter (which would include one or more circuits that help perform the limiting function discussed above). As a result, the calibration technique (s) discussed below may be used with demodulator designs other than the particular demodulator design 106 shown in Figure 1.

Referring to Figures 1 and 3a, b, note that if the transmitting modulator 105 transmits digital data corresponding to one logical value (e. g., a continuous stream of"1"s or a continuous stream of"0"s) the FSK modulated signal corresponds to a signal having one frequency (i. e., a tone) rather than a signal that alternates between two frequencies. For example, if the transmitting modulator 105 of Figure 1 sends a continuous stream of logical"0"s, the transmit signal 301 will resemble a tone having a single frequency of fcarlier-fo. Similarly, if the transmitting modulator 105 of Figure 1 sends a continuous stream of logical"1"s, the transmit signal 301 will resemble a tone having a single frequency of f, + fo.

For a transmit signal 301 resembling a tone having only one frequency, a clipped limiter output signal 411 will resemble a digital signal having a fixed frequency (e. g., a dock signal). For the demodulator design 106 shown in Figure 1, where a limiter 125,170 is positioned subsequent to the mixers 116,117 that downconvert the carrier frequency of the received airborne signal from fcarl, er to fjp (as discussed with respect to equations 1 through 3), the frequency fii w of the limiter 125,170 output signal will be fjp-fo or fF+ fo depending upon the continuous logical value being transmitted by the transmitting device.

Figure 4 may be viewed as one possible embodiment of the approach shown in Figure 3. Referring to Figures 3 and 4, limiter 425 corresponds to a limiter in a demodulator 306,406 where the demodulator 306,406 is receiving a transmit signal 301 that corresponds to a continuous stream of logical values (e. g., a continuous stream of"1"s or a continuous stream of"0"s). The limiter output signal 311,411 is"tapped"from the demodulator 406 and its frequency is measured by the frequency measurement circuit 310 in order to calculate the deviation frequency fo of the transmit signal 301.

That is, because fF is a known constant for a particular demodulator 306, 406 design, the deviation frequency fo may be determined according to: fo = f,, m ;, er-f¢ or fo = flF flimiter Eqns 4a, 4b depending on which logical value is used during transmission (note that flirniter corresponds to the frequency of the tapped limiter output signal 311,411 as seen in Figure 4).

Consistent with a transmitting device embodiment where a logical"0" corresponds to f,-fo, fo can be determined according to fo = flF-flimiter Referring back to Figure 3, the continuous stream of logical values may be encapsulated in a packet sent from the transmitting device 365 to the receiving device 366. That is, in an embodiment, the"data"carried by the packet is loaded only with ls or only with Os. Note that in various applications the data sent between the transmitting and receiving devices is scrambled or otherwise encoded before being transmitted so that it is balanced (i. e., has equal or approximately equal amounts of ls and Os). In order to construct a packet having only ls or only Os, therefore, such scrambling/decoding functionality should be disabled or routed around.

Because digital signals operating at or near the intermediate frequency fiF may be processed by inexpensive digital circuitry (e. g., circuitry fabricated upon a large wafer with a CMOS logic manufacturing process), deliberately creating a digital tapped signal 311 from a portion of the demodulator 306 channel following the demodulation performed by mixers 116,117, allows the offset frequency fo measurement to be cost effectively implemented within a frequency measurement circuit 310 formed with such digital circuitry. Note that other approaches, alternative to taking a clipped output from a demodulator's limiter, may be used to formulate a digital tapped receive signal 311.

For example, an analog received signal tapped from within the demodulator (e. g., a received signal resembling a sinusoidal waveform such as a limiter's input signal 401 seen in Figure 4) may be formed into a digital signal.

That is, an analog signal may be tapped, amplified and/or clipped by an amplifier other than a demodulator's limiter. Alternatively or in combination, an analog received signal within the demodulator may also be tapped and coupled to a comparator that produces logic values based upon a comparison of tapped receive signal against another voltage (which may be referred to as a threshold).

Furthermore tapped signal 311, as presented to the frequency measurement circuit 310, is not limited to a digital signal. That is, the frequency measurement circuit 310 may also be designed with analog or mixed signal circuitry designed to process an analog tapped receive signal. Furthermore the frequency of the tapped receive signal may at/near the intermediate frequency (such as fjp +/-fo) or at/near the carrier frequency (e. g., fer +/-to).

The tapped receive signal 311 therefore does not require to be tapped from the output of a limiter. Generally, in order to measure the deviation frequency fo, the tapped receive signal 311 may be tapped from anywhere along the demodulator channel prior to the frequency to voltage converter 128. Note that some degree of processing may be performed upon the tapped receive signal 311 before its frequency is measured by the frequency measurement circuit 310.

In an embodiment, the tapped receive signal 311 may be directed to a sample and hold circuit so that samples of the tapped receive signal are produced. These samples may be further processed by a frequency measurement circuit 310 that processes signal samples rather than a continuous signal. For example, in an embodiment, the frequency measurement circuit 310 is implemented as a digital signaling processor (DSP) (coupled to a memory) that executes software having associated algorithms that measure the deviation frequency fo).

As another example, if the tapped receive signal 311 is a clipped limiter output (as shown back in Figure 4) output signal it will typically swing between the V+ and V-voltage rails 426,427 of the limiter. If V-corresponds to a negative voltage, the tapped receive signal 311 may require some form of effective level shifting and/or amplitude adjustment before being processed by frequency measurement circuit 310.

Because digital circuits typically use a ground reference, continuous application of negative voltages at an input may cause damage in some cases (depending on the protection offered by the technology used to implement the digital circuit). Level shifting and/or amplitude adjustment changes the minimum and maximum voltage levels of the tapped receive signal 311 before it is presented to the frequency measurement circuit 310 so that the risk of damage to the measurement circuit is reduced.

In embodiments where a negative supply voltage is not employed, V-in Figure 4 may correspond to a ground reference. In these environments, processing of the tapped receive signal 311 before presentation to the frequency measurement circuit 310 may be desired. For example, if the limiter is designed to drive the next stage in the demodulator 306, a circuit may be introduced between the demodulator 306 and the frequency measurement circuit 310 to make the load of the frequency measurement circuit 310 insignificant to the limiter output. In other embodiments, the tapped receive signal is differential (e. g., because the limiter has a differential output). A circuit that converts the differential signal into a single ended signal may be used to provide a single ended tapped receive signal 311 to the frequency measurement circuit 310.

Figure 5 relates to an approach that may be used by the frequency measurement circuit 310 of Figure 3b to measure the frequency of the tapped receive signal 311. As seen in Figure 5, the tapped receive signal 511 resembles a digital dock. Thus for simplicity, in the example of Figure 5, the tapped receive signal 511 corresponds to a digital signal.

The approach of Figure 5 also shows a reference signal 512 having a reference frequency fus : that is equal to 1/TREF. A reference signal is a signal within the receiving device having a known frequency. The reference signal may have a frequency known to a fair degree of precision such as the frequency of a crystal oscillator. A crystal oscillator frequency, or a fixed frequency derived from it, may therefore be used as the reference signal 512 frequency. In a sense, the reference signal 512 acts as a yardstick that measures the time consumed by the tapped receive signal 511.

In the approach of Figure 5, the tapped receive signal 511 is observed and the number of observed cycles (i. e., the number of periods T ;, , that elapse) are counted. Simultaneously, the reference signal 512 is observed and the number of observed cycles (i. e., the number of periods TREF that elapse) are also counted.

When the number of observed cycles in the tapped receive signal 511 reaches a predetermined amount, P, the counting activity is stopped for each signal 511, 512.

At this point, the count for the tapped receive signal 511 will be P and the count for the reference signal 512 will be Q. In the approach of Figure 5, the reference signal frequency fREF is greater than the tapped receive signal 511 frequency As a result, Q will be greater than P when the counting ceases.

The approximate frequency flirniter of the tapped receive signal 511 is determinable because the number of cycles executed by the tapped receive signal 511 is known (P), and the amount of time that was required to execute P cycles can be measured.

The amount of time required to execute P cycles by the tapped receive signal 511 may be approximated as: Q TREF = Q/fEF. Eqn. 5 Furthermore, because the amount of time required to execute P cycles by the tapped receive signal 511 is also equal to PT, (as seen in Figure 5), using Equation 5 above: flimiter = 1 / Tlimiter # (P/Q) FREF Eqn. 6 Thus, because P and f are known before the measurement and Q is the result of the measurement, the frequency of the tapped receive signal may be determined. The offset frequency fo of the transmit signal may finally be approximated by substituting Eqn. 6 into Equations 4a and/or 4b.

Note that equation 6 is expressed as an approximation. The approximation stems from the fact that, in this example, the source of the tapped receive signal 511 frequency fiirniter (e. g., an oscillator in the transmitting device) is different than the source of the reference signal 512 frequency f (e. g., an oscillator in the receiving device).

Since these sources are uncorrelated, in various embodiments the reference signal 512 and tapped receive signal 511 will have frequencies fREF, flimuter that are not integer multiples of another. If so, the edges of these signals 511,512 can not be kept at a constant phase relationship with respect to each other. If the cycles for each signal are counted by counting the observed signal edges (i. e., counting each rising edge), there may be some error as a result of the signal edges not being able to maintain a constant phase relationship.

Specifically, the count Q determined for the reference signal 512 may be terminated within + or-TREF of the actual time in which the Pth cycle of the tapped receive signal 511 is reached (if only one edge per cycle is (counted).

This is analogous to Q being in error at a maximum by +1 or-1.

The worst case accuracy of the fl ; n, ; ter frequency measurement may be expressed as: Accuracy = 1. 00 +/- (f/ (fpP)) Eqn. 7 An accuracy value of 1.00 corresponds to a perfect measurement, thus the (fl ter/(fREFP)) term in Equation 7 corresponds to the error in the flimiter frequency measurement. Note that the error in the fl ter measurement approaches zero as P increases and as fREF becomes greater than fl ter. Using equation 7, as some examples, the worst case error may be kept within: 45% of the actual f, r, ter value if (flimiter/fREFP) is less than or equal to 0.35,10% of the actual flimiter value if (fl ter/fREFP) is less than or equal to 0.10,6% of the actual limiter value if (fl ter/fREFP) is less than or equal to 0.05, etc. The error term in Eqn. 7 may be reduced by half if both edges are counted rather than a single edge.

Also note that, in an embodiment where a continuous stream of Os corresponds to an flimier value of fjp-fo (and a continuous stream of 1s corresponds to an flimiter value of fjp + fo), the fl tervalue for a continuous stream of Os is less than the flimiter value for a continuous stream of ls (i. e., tip-to < fF + fo). As such, noting that the accuracy of the frequency measurement improves as fREF becomes greater than flimiter, for a fixed valueoffpthe accuracy of a measurement made with a continuous stream of Os should be better than the accuracy of a measurement made with a continuous stream of ls.

Figure 6a shows an embodiment for a frequency measurement circuit 610 that can perform the measurement technique discussed above. A first counter 601 is clocked by the tapped receive signal 611 and a second counter 602 is clocked by the reference signal 612. Both counters may be reset by the same signal so that the counting for both signals is initiated at approximately the same time.

When the first counter 601 reaches a value of P, a comparator 603 triggers the output value Q of the second counter 602 to be held (e. g., by shutting off the reference signal 612 to the clock input of the second reference counter). Once the Q value is known, Equations 6 and 4a or 4b may be used to calculate the frequency offset fo of the transmit signal by a digital logic circuit 604.

Figure 6b shows an alternative embodiment 620 that helps minimize the error referred to above with respect the frequency measurement. The embodiment 620 of Figure 6 is similar to the embodiment 610 of Figure 6a.

However, a pair of tapped receive signals 611a, 611b are counted separately and their count value is averaged by an averaging circuit 630. The pair of tapped receive signals 611a, b may be taken from opposite quadrature arms within the demodulator.

For example, referring to Figures 1 and 6, tapped receive signal 611a may be taken from the output of limiter 170 while tapped receive signal 611b may be taken from the output of limiter 125. Averaging the counts from both quadrature arms should minimize frequency measurement errors caused by imperfect transmitting device or receiving device phase splitters (e. g., an 89 or 91 degree phase shift rather than a 90 degress phase shift) because the phase errors of the pair of tapped receive signals 611a, 611b with respect to the reference signal 612 will offset one another.

Referring back to Figure 3, once the frequency offset fo is known, the frequency offset fo is sent back to the transmitting device 365 (e. g., in a packet sent from the receiving device 366 to the transmitting device 355). The transmitting device can then readjust its deviation frequency to a more desirable location (e. g., closer to the middle of an allowable deviation frequency range).

Figure 7 shows a modulation technique performed in the transmitting device referred to as open loop modulation. It is important to note, however, that modulation schemes other than an open loop modulation scheme can be employed by a transmitting device consistent with the teachings herein. Open loop modulation has been chosen for the present discussion because of its relative ease of comprehension.

Within the transmitting device (e. g., transmitting device 365 of Figure 3a), a frequency synthesizer 700 is used to generate the proper carrier frequency for a wireless channel (as indicated by channel select value 741). More details about the operation of such a frequency synthesis circuit 700 may be found a U. S.

Patent entitled Method and Apparatus for Offset Cancellation In a Wireless Receiver, filed on September 27,2000.

A voltage controlled oscillator (VCO) 721 within a phase lock loop circuit 795 produces an output signal that is eventually fed, in some form, to an antennae 802 as modulated data. A corresponding frequency domain representation 150 of the VCO 721 output signal is also shown in Figure 1. As discussed above, the frequency of the VCO 721 output signal is a function of the data being transmitted. That is, if the data to be transmitted corresponds to a first logic value (e. g.,"1"), the VCO 721 output signal has a frequency of fcarr, er+ fo. If the data to be transmitted corresponds to a second logic value (e. g.,"0"), the VCO 721 output signal has a frequency of fcarrier-fo. Thus, the VCO 721 output signal 150 alternates between frequencies of fcarrier + fo or fcarrier - fo depending on the data being transmitted.

The carrier frequency fcarrier corresponds to the particular wireless channel that the digital information is being transmitted within. For example, within a BLUETOOTH wireless system, fc« ; er corresponds to 3.402 Ghz for the first channel 110, (referring briefly to Figure 1). The frequency of the VCO 721 output signal 150 is a function of the VCO 721 input signal voltage as shown in the VCO transfer curve 799. Thus, two frequencies fo, feamer associated with the VCO 721 output signal may be formed by two separate input signals.

Said another way, the overall VCO input signal may be seen as the combination of a first and second input signals. The first input signal is a DC signal 797 that determines fcarl, er and therefore corresponds to the channel that information is being transmitted over. The second input signal is an AC signal 798 that corresponds to the data being transmitted (i. e., an analog baseband signal) and whose amplitude 799 determines the deviation frequency fo. A brief discussion of each follows.

The first VCO input signal is a DC signal 797 used to set fcarr, er, The first VCO input signal corresponds to the output of a loop filter 794 found within a phase lock loop circuit 795 that includes the VCO 721. Phase lock loop circuit 795 is responsible for multiplying the frequency of a local oscillator 732 to the proper fcurler frequency according to a channel select input 741 value. The channel select input 741 value indicates the channel over which the baseband signal b (t) is to be transmitted. Thus, if the channel select input 741 value corresponds to channel 1101 for a BLUETOOTH application, the loop filter 794 will produce a DC signal 797 that corresponds to a fc., i,, of 3.402 GHz at the VCO 721 output.

The second VCO input signal is an AC signal 798 that corresponds to the data being transmitted (i. e., an analog baseband signal) and whose amplitude 799 determines fo. That is, if the amplitude 799 of the AC signal 798 is reduced, fo is reduced (i. e., frequencies 791,793 migrate closer to the fc, 6,, frequency); and, as the amplitude 199 of the first input signal is increased, fo is increased (i. e., signals 791,793 migrate farther away from the carrier frequency).

A digital baseband signal b (t) corresponding to the digital information to be transmitted is presented to the input of a filter 792 (such as a filter having a gaussian transfer curve). AC signal 792 corresponds to the output of filter 792.

A such, the output of filter 792 is applied to the input of the VCO 721. Note that it is common practice to apply the DC and AC signals to the VCO 721 at different times. That is, initially only the DC signal 797 is applied to the VCO 721 for the selected channel.

Then, the DC signal 797 is removed from the VCO 721 input (e. g., by opening switch 789). Then, the AC signal 798 is applied (e. g., by closing switch 788). The VCO 721 should have sufficient internal capacitance to hold the DC signal 797 for the time during which switch 788 is closed and switch 789 is open.

The DC and AC input signals 797,798 are applied at different times because the alternating frequency behavior associated with VCO output signal causes the loop filter 194 output to become unstable (via the feedback within phase lock loop circuit 795).

In order to adjust the deviation frequency fo in light of the measurement taken by a receiving device, the gain of the VCO 721 (i. e., the slope of the VCO transfer curve 799) should be accounted for when determining the appropriate adjustment. Figure 8 shows an exemplary VCO 821 having a first input 801 which corresponds to the input from the loop filter 794 of Figure 7; and a second input 802 which corresponds to the input from the filter 792 in Figure 7.

The LC tank circuit 803 produces a resonance if a voltage is applied at either or both inputs 801,802. Thus, the VCO 821 operates by applying a voltage at either inputs 801,802 which results in the creation of a resonance across the LC tank circuit 803. A resonance is the repeated transfer of energy from the inductor L to the capacitor C and back again (from the capacitor C to the inductor L). Because the rate of energy transfer is constant (for constant voltage (s) placed at inputs 801,802), the frequency of the signal observed across the LC tank circuit 803 is a constant.

The frequency of the resonance is given by f = 1/(27r (LC) g Eqn. 8 where L is the inductance of the inductor and C is the capacitance of the capacitor. In an embodiment, the VCO 821 is responsive (i. e., changes output signal frequency with) the voltage (s) presented at inputs 801,802 because the capacitor C changes its capacitance as a function of the applied input voltage.

For example, the capacitor may be implemented as a reverse biased diode whose depletion width varies as function of the reverse bias voltage (i. e., the voltage from inputs 801,802). The depletion width of a diode determines its capacitance.

Thus, if the voltage at either input 801,802 changes, the capacitance in the tank circuit 803 changes. This will cause the frequency of the resonance to change in light of Equation 9 above. For reversed biased diodes, however, the change in capacitance is not linear with the change in voltage. For example, for diodes formed with an integrated circuit manufacturing process, the capacitance varies approximately as C Eqn. 9 where V is the voltage applied across the diode and k is a constant. An exemplary curve 800 of Eqn. 9 is shown in Figure 8. Note that resistors 807,808 result in the mean voltage applied across the tank circuit 803 being different than the mean voltage applied at the VCO inputs 801,802. Nevertheless, Equation 9 may also be used to approximately model the change in capacitance with the VCO input voltage where V corresponds to the summation of the DC signal and the AC signal.

Note the non linear behavior of curve 800. Due to this non-linear behavior, greater change in capacitance per unit change in voltage is observed in one region (e. g., region 811) than in another region (e. g., region 812). That is, the slope of curve 800 changes. Figure 8 also shows an exemplary range of VCO frequencies 804 (and corresponding RF channels) that may be synthesized from the range of capacitance shown in curve 800. Because the capacitance is non linear with VCO input voltage, the range of synthesized RF channels will be non-linear with VCO input voltage as seen in Figure 8b.

Because of the non linearity discussed above, identical adjustments in VCO frequency may be implemented with different adjustments made to the VCO input voltage. For example, referring to curves 800 and 804, note that for equal adjustments in VCO frequency at opposite ends of the frequency range (e. g., from frequency fl to frequency f2 and from frequency f3 to f4 where f2- fl=f4-f3), different voltage adjustments will be needed (e. g., V4-V3 > V2-V1).

Similarly, the same adjustment in VCO input voltage can produce different VCO frequency adjustments at opposite ends of the VCO frequency range 804.

The adjustment (s) to be made to the deviation frequency fo (in response to the measurement made by a receiving device) may be implemented by changing the gain of the filter 792 of Figure 7. That is, by increasing the gain of the filter 792, the amplitude 799 of the AC signal 798 will increase (causing an increase in the deviation frequency fo) ; and by decreasing the gain of the filter 792, the amplitude 799 of the DC will decrease (causing a decrease in the deviation frequency fo).

One may calculate the adjustment needed for a particular channel in light of the non linearity exhibited in curves 800 and 804. For example, in one approach shown in Figure 9, a first correcting adjustment 901 to the deviation frequency fo is made at one end of the frequency band of interest (e. g.,, near fl and f2 in curve 804 of Figure 8) and a second correcting adjustment 902 is made at the opposite end of the frequency band of interest (e. g., near f3 and f4 in curve 804 Figure 8).

The correcting adjustments may be made by recursive techniques as between the transmitting devices. That is, after the transmitting device receives a first measurement from the receiving device, it calculates a new filter gain and sends another transmit signal to the receiving device for a second measurement.

The receiving device performs a second measurement and forwards it to the transmitting device. The transmitting device calculates a second filter gain in response. The process repeats until the deviation frequency reaches a desired value within its allowable range. At this point, the correcting adjustment 901 of Figure 9 is made.

The process is repeated to determine a second correcting adjustment made at the opposite end of the frequency band. Once the appropriate adjustments 901,902 are calculated for both of these band regions, the remaining adjustments are calculated as a linear interpolation 903 that connects both points 901,902. Configuring the adjustments in this manner adjusts each of the channels properly in view of curve 804 of Figure 8.

As mentioned initially, the above described approach may be used in a case where the deviation frequency fo is drifting but the intermediate frequency fiv ils not drifting or is assumed not to be drifting. As such, the transmitted deviation frequency fo is measured and calibrated. The following discussion provides a description of additional cases (e. g., where a drifting intermediate frequency fjp is accounted for) and approaches that can handle them.

Referring to Figure 3b, the frequency measurement circuit 310 may be configured to not only measure a tone corresponding to fF-fo (e. g., as produced by a continuous stream of logical 0's) but also a tone corresponding to fIF + fo (e. g., as produced by a continuous stream of logical 1's).

In order to measure a tone corresponding to fjp + fo, in an embodiment, a packet having a payload comprising only 1s is sent from the transmitting device 355 to the receiving device 356. The frequency measurement circuit 310 can measure the tone by the same approach as outlined with respect to Figures 5 and 6. Note that the inaccuracy associated with the frequency measurement, as originally discussed with respect to Equation 7, exists. However, the inaccuracy may be kept to within tolerable limits by keeping filter sufficiently smaller than FIEF- Referring to Figures 3 and 10, by measuring both tones, the frequency measurement circuit 310 can determine the range 1001 of frequencies over which the baseband signal is transmitted as well as the particular tone frequencies 1002, 1003 at the edges of the range 1001 that correspond to a particular baseband logic value (e. g., a 1 or a 0). Furthermore, the frequency measurement 310 circuit may be configured to measure the intermediate frequency f.

For example, if the transmitting device 365 sends only a carrier signal (as described in more detail below), the tapped receive signal 311 measured by the frequency measurement circuit 356 will have a frequency 1004 corresponding to the intermediate frequency f ; p (if tapped downstream from the mixers 116,117 of Figure 1). As such, the frequency measurement circuit 310 can develop a complete understanding of the wireless link between the transmitting and receiving devices 365,366.

That is, the frequency measurement circuit 310 can detect drifting of the intermediate frequency fla if the measured intermediate frequency 1004 has an offset (such as offset 254 seen in Figure 2b) with respect to its designed for value (e. g., 3 MHz). Furthermore, by also measuring the tone frequencies 1002,1003 as described above, the frequency measurement circuit 310 can monitor or otherwise observe the effect on the position of the tone frequencies 1002,1003 as caused by drifting in the intermediate frequency f, and/or deviation frequency fo.

The receiving device may 366 adjust its downconversion frequency fd.,,, in response to this understanding of the wireless link (e. g., in order to properly position the frequencies 1002,1003,1004) and/or send its understanding of the wireless link (i. e., the measured intermediate 1004 and tone frequencies 1002, 1003) to the transmitting device 365. In response, the transmitting device 365 can also adjust its carrier frequency fcarli « = and/or deviation frequency fo.

Recall from above that the intermediate frequency f¢ can be measured by the frequency measurement circuit 310 if the transmitting device 365 sends an unmodulated carrier. Referring to Figure 7, sending an unmodulated carrier corresponds to applying only the DC signal 797. That is, switch 788 is never closed. Alternatively, a balanced AC signal 798 (e. g., a signal that corresponds to a 1010 pattern) may be modulated and the offset of the baseband signal as received from the demodulator 306 output may be measured by offset detection circuitry downstream from the demodulator. Referring briefly back to Figure 1, with knowledge of the frequency to voltage conversion 160 performed by the frequency to voltage converter 128 in the demodulator 106, the intermediate frequency f, F of the wireless link may be deduced.

Note that typical wireless devices have transmitting circuitry and receiving circuitry. That is, typical wireless devices may act as a transmitting device or a receiving device depending upon which particular role (i. e., transmitting or receiving) they currently have to perform with respect to communication over a wireless link. Referring to Figures 3a and 3b, in an alternate embodiment, the transmit signal 301 corresponds to a signal passed from the transmit circuitry portion of a wireless device to the receive circuitry portion of a wireless device. That is, the transmit signal 301 is kept within the same wireless device rather than passed between two separate wireless devices.

It is important to point out that the discussion above is applicable to other applications besides frequency measurement associated with BLUETOOTH devices. That is, the teachings above may be applied to other frequency shift keyed wireless technologies besides BLUETOOTH such, as just a few examples, HomeRF, IEEE 802.11, GSM and Digitally Enhanced Cordless Telephony (DECT).

Embodiments of the present description may be manufactured as part of a semiconductor chip (e. g., by a planar semiconductor manufacturing process).

Note also that embodiments of the present description may be implemented not only as part of a semiconductor chip but also within machine readable media.

For example, the designs discussed above may be stored upon and/or embedded within machine readable media associated with a design tool used for designing semiconductor devices. Examples include a netlist formatted in the VHSIC Hardware Description Language (VHDL) language, Verilog language or SPICE language. Some netlist examples include: a behaviorial level netlist, a register transfer level (RTL) netlist, a gate level netlist and a transistor level netlist. Machine readable media also include media having layout information such as a GDS-II file. Furthermore, netlist files or other machine readable media for semiconductor chip design may be used in a simulation environment to perform the methods of the teachings described above.

Thus, it is also to be understood that embodiments of this invention may be used as or to support a software program executed upon some form of processing core (such as the CPU of a computer) or otherwise implemented or realized upon or within a machine readable medium. A machine readable medium includes any mechanism for storing or transmitting information in a form readable by a machine (e. g., a computer). For example, a machine readable medium includes read only memory (ROM); random access memory (RAM); magnetic disk storage media; optical storage media; flash memory devices; electrical, optical, acoustical or other form of propagated signals (e. g., carrier waves, infrared signals, digital signals, etc.); etc.

In the foregoing specification, the invention has been described with reference to specific exemplary embodiments thereof. It will, however, be evident that various modifications and changes may be made thereto without departing from the broader spirit and scope of the invention as set forth in the appended claims. The specification and drawings are, accordingly, to be regarded in an illustrative rather than a restrictive sense.