SIGNALS TECHNICAL FIELD

[0001] The present invention relates to satellite radio-navigation signals, and more particularly to a method and receiver for processing Offset Carrier Modulated (OCM) ranging signals. BACKGROUND

[0002] Through the use of systems such as GPS, satellite navigation has become a critical element of society and economy. Global Navigation Satellite Systems (GNSS) involve the transmission of radionavigation signals to (typically, but not exclusively, ground-based) receivers where they are processed and used for ranging purposes, or to calculate a position, velocity, time (PVT) solution. [0003] Many modern Global Navigation Satellite System (GNSS) signals broadcast composite Code Division Multiple Access (CDMA) signals which use an Offset Carrier Modulation (OCM). These signals incorporate varying numbers of baseband components and a range of sub-carriers. Examples include (i) Binary Phase-Shift Keyed (BPSK) baseband signals modulated by sinusoidal sub-carriers resulting in OCM signals, (ii) BPSK baseband signals modulated by square-wave sub-carriers, resulting in Binary Offset Carrier (BOC) sig- nals, and (iii) Quadrature Phase-Shift Keyed (QPSK) signals using sinusoidal sub-carriers. In general, these signals exhibit a symmetric Power Spectrum Density (PSD) with little power located at the centre frequency and two main lobes, located at either side of the signal centre frequency, which contain the majority of the signal power. [0004] This spectral shape, coupled with the autocorrelation properties of the baseband CDMA components, yields a signal which can provide high accuracy ranging. The autocorrelation function of such signals is typically steep and exhibits numerous zero- crossings. As the ranging accuracy provided by these signals is directly related to the signal autocorrelation function, these signals are often tuned to have a high slope near around the zero-offset point. However, this comes at a cost, which is generally manifest as difficulties experienced by the receiver in the initial signal acquisition phase, and when strong multipath conditions (involving reflected signals) prevail. [0005] As many GNSS signals are broadcast from each satellite, it is not uncommon that the centre-frequency of offset-carrier modulated signal coincides with a second signal which either has been modulated with either (a) no sub-carrier, or (b) a sub-carrier of a low frequency. [0006] To demonstrate the challenges of processing offset-carrier modulated signals, an example OCM signal configuration will now be discussed, with reference to Figs 1 to 5 (PRIOR ART). [0007] The particular signal chosen for illustration purposes is an OCM which uses a square- wave sub-carrier, typically termed a BOC modulation, with a primary code rate of (2.5 x 1.023) Mcps and a cosine-phased sub-carrier rate of (15 x 1.023) MHz. The composite modulation, denoted BOC _c (15,2.5) has a normalized PSD and autocorrelation function depicted in Figs 1 and 2, respectively. [0008] Specifically, the signal of interest (a down-converted and digitized version of the radionavigation signal received at the receiver’s antenna) is denoted s _A (t) which is modelled as follows:

where P _A denotes the nominal received power, F _A is the nominal broadcast centre frequency, CA (t) is the CDMA spreading sequence, and SC _A (t) is the square-wave sub-carrier. Estimates of various signal parameters including, for example, F _A and θ _A , are generally extracted via correlation of the received signal and a local replica, the result, typically termed the correlator value and denoted Y _A (f, τ, θ), is computed via:

where T _I , often termed the pre-detection integration period is generally of short duration, perhaps some milliseconds, and is generally chosen in accordance with the period of sCDMA spreading sequence, CA. [0009] One feature of this modulation that can be challenging for a receiver is the presence of multiple, so-called, side-peaks in the autocorrelation function, leading to acquisition ambiguity. When a receiver attempts to acquire such a signal, it typically implements a search across the code-delay τ, striving to detect the largest autocorrelation peak. Ideally this will correspond to the alignment between the received signal and the local replica signal. A problem is that, due the large relative magnitude of the adjacent peaks, both positive and negative, of the BOC _c (15, 2.5) autocorrelation function, the presence of thermal noise interference can lead a receiver to identify one of the adjacent local-maxima as the maximum value. In terms of receiver operation, this can correspond to a bias in the measured range and, thereby, degrade positioning accuracy. [0010] As a demonstration of this particular problem, we consider that the signal has been acquired by detecting and tracking each of its components parts, the upper and lower side- lobes, separately. This corresponds to the individual or joint acquisition of one or both of the BPSK signals centred at F _c (15 1.023) MHz. Given this coarse acquisition estimate, a receiver may begin to track the BPSK signals to refine the delay and frequency alignment and, subsequently, attempt a fine acquisition of the composite BOC _C (15, 2.5) signal. In doing so, the receiver may populate an acquisition search space, across the delay uncertainty. Typically this search will have a finite range and finite delay resolution, such that the uncertainty space occupies samples of the autocorrelation function, depicted in Fig. 2. As an example, we assume that the receiver may not be coherently tracking the signal, such that there may be a phase uncertainty and, therefore, might implement a non-coherent detection scheme. [0011] The decision variable ( ^Y _A ^2) produced by examining the square magnitude of a complex correlation Y _A between a received signal and a local replica, having perfect frequency synchronization, unaligned phase and a range of code-delays is presented in Fig. 3. When attempting to align the local replica signals with the received GNSS signals, the receiver may observe a range of code delays around the current best estimate. This range will depend on the uncertainty of the current code delay estimate.

[0012] As an example of this problem, Fig. 4 depicts the probability of choosing the correct code-delay when examining a range of correlator values, spaced at 1 metre intervals across a range extending ^30 metres for a selection of received C/N0 values. While in the absence of thermal noise, selection of the appropriate code delay will be trivial, upon inspection of Fig.4 it is clear that the performance may degrade rapidly with reduced signal quality. In particular, and as seen also in Fig. 3, it is noteworthy that the local maxima immediately adjacent to the (central) global maximum have relative magnitudes of almost 0.8. [0013] Results are presented in Figure 4 wherein it is clear that a receiver will experience significant difficulty in acquiring the appropriate code delay under weak signal conditions. Of course, the results presented here correspond only to the case where a receiver integrates over a period of T _I = 1 ms. The performance can be improved by extending the integration period, however, this period is ultimately limited by the signal design and receiver operating conditions. [0014] One further challenge experienced by receivers processing BOC signals is that of false-lock of the code tracking architecture: multiple stable lock points. Generally, a receiver will form some sort of discriminator to estimate misalignment spreading sequence, CA, and secondary code, SC _A between the received signal and the local replica. This is typically done by generating correlator values that are equally spaced, early and late, relative to the best estimate of the code delay. Differencing these early and late correlator values, respectively denoted YE and YL , can generate the code-delay error estimate. [0015] Depending on the receiver design, it may or may not coherently track the phase of the received signal. In cases where the received signal is tracked a coherent estimate can be made and if the signal phase is not tracked or if it is likely to be misaligned, then a non-coherent estimate can be made. For example, basic coherent and non-coherent delay estimates can be made via:

where A _coh and A _non-coh are normalizing gains, generally a function of both the received signal strength, the signal modulation type and the relative spacing between the early and late correlator values; and denotes the real part of a complex value x. Functions e _coh and e _{non- coh} generally produce an error estimate that is proportional to the true delay for a small range of delay values, centred around zero. A problem is that, outside this range, the error function can exhibit positive-sloped zero-crossings at which a code tracking scheme may experience a stable lock. These, so called, false-lock points can lead to biases in the measured range. The more complex the signal modulation, the greater the number of these false-lock points. Also, in the case of the BOC modulation, the non-coherent case will exhibit more false-lock points than the coherent case. [0016] Figure 5(a) depicts the coherent code error estimate, and Fig.5(b) depicts the non- coherent code error estimate, of a BOCc(15, 2.5) signal given an early-to-late correlator spacing of 5 m. In the coherent case, the modulation results in twelve stable lock points which do not correspond to the true signal delay, although perhaps only ten of these are significant. More troubling is that in the non-coherent case this number increases to twenty-four and the range, over which the error estimate is proportional to the true error, shrinks by a factor of two. The implications of this are that a receiver, when operating in non-deal conditions, such as fading or high-dynamics, may struggle to converge to the correct stable lock point, resulting in biased range measurements. [0016a] US2014119392A discloses a receiver for receiving a composite signal transmitted from a satellite, such as a navigation satellite (e.g., a multiplexed binary offset carrier signal or pilot component of the L1C signal for the Global Positioning System(GPS)) the receiver being capable of at least partially decoding the received composite signal that is received. In one embodiment, the received composite signal is from a Galileo-compatible navigation satellite or Global Positioning System satellite. In one embodiment, the received composite signal refers to a first binary offset carrier signal that is multiplexed with a second binary offset carrier signal.

[0016b] EP2402787A1 discloses a GNSS receiver that can perform correlation processing on a positioning signal phase-modulated by a CBOC signal. A correlation processing module performs correlation processing between a baseband signal and a BOC(1, 1) replica code to output a BOC(1, 1) correlation data, and also performs correlation processing between the baseband signal and a BOC(6, 1) replica code to output a BOC(6, 1) correlation data. SUMMARY OF THE INVENTION [0017] In one aspect of the invention there is provided a method of processing offset carrier modulated (OCM) ranging signals in a radionavigation system comprising a plurality of satellite-borne transmitters and at least one ground-based receiver, the receiver being adapted to carry out the method, the method comprising: receiving a first radionavigation signal from at least one of the plurality of transmitters and deriving therefrom a first OCM signal S _A ; receiving a second signal S _B synchronously broadcast with the first OCM signal S _A , the second signal S _B having the same or nearby centre frequency to the first OCM signal S _A ; generating a combined correlation value Y _C , the combined correlation value Y _C corresponding to the correlation of a combined signal S _C with a replica of the first OCM signal, the combined signal S _C resulting from the coherent combination at the receiver of first OCM signal S _A with the second signal S _B ; and deriving ranging information based on the combined correlation value YC. [0018] In one embodiment, the centre frequency of the second signal S _B is selected such that the power spectral density (PSD) of the second signal S _B occupies the bandwidth contained between two lobes of the first OCM signal S _A . The second signal S _B may have (i) no subcarrier or (ii) a subcarrier SC _B , the subcarrier SC _B being of lower frequency than a subcarrier SC _A of the first signal S _A . The subcarrier SC _B of the second signal S _B may be a square wave [0019] In one embodiment, the centre frequencies of the first OCM signal S _A and the second signal S _B differ by no more than the sum of the sub-carriers of the first OCM signal S _A and the second signal. [0020] In one embodiment, the centre frequencies of the first OCM signal S _A and the second signal S _B satisfy

where the first OCM signal SA and the second signal SB have centre frequencies , respectively, and have sub-carrier frequencies , respectively. [0021] In one embodiment, the centre frequencies of the first OCM signal S _A and the second signal S _B satisfy where the first OCM signal S _A and the second signal S _B have centre frequencies , respectively, and have sub-carrier frequencies , respectively. [0022] The second signal S _B may be synchronously broadcast with the first OCM signal S _A . [0023] The second signal S _B may comprise one of (i) an OCM signal and (ii) a BOC signal. [0024] In one embodiment, generating a combined correlation value Y _C comprises:

coherently combining the first OCM signal SA with the second signal SB according to ; and:

generating, using a combined integrate and dump function, the combined correlation value Y _C from s _C (t) and the replica signal. [0025] In one embodiment, generating a combined correlation value Y _C comprises:

generating, using a first integrate and dump function, a first correlation value Y _A from the first OCM signal s _A (t) and the replica signal according to ; where C _A (t) is the CDMA spreading sequence, and SC _A (t) is the sub-carrier, of the first OCM signal S _A ;

generating, using a second integrate and dump function, a second correlation value Y _B from s _B (t) and the replica signal according to

where C _B (t) is the CDMA spreading sequence, and SC _B (t) is the sub-carrier, of the second signal S _B ; and

coherently combining the first correlation value Y _A and the second correlation value YB to form the combined correlation value YC. The subcarrier SCA of the first signal SA, and/or the subcarrier SC _B of the second signal S _B , may be a square wave. [0025a] In one embodiment, generating a combined correlation value Y _C comprises generating Y _C as the weighted sum

where Y _A and Y _B are correlation values derived from the first OCM signal S _A and the second signal S _B , respectively, and k _A and k _B are weighting factors. [0026] Generating a combined correlation value Y _C may comprise generating Y _C according to

where C _A (t) is the CDMA spreading sequence, and SC _A (t) is the sub-carrier, of the first OCM signal S _A ;

where C _B (t) is the CDMA spreading sequence, and SC _B (t) is the sub-carrier, of the second signal S _B ; and

where k _A and k _B are weighting factors and In one embodiment, k _A = k _B . The

subcarrier SC _A of the first signal S _A , and/or the subcarrier SC _B of the second signal S _B , may be a square wave. [0027] The method may further comprise providing a module for generating, based on the combined correlation value Y _C , a code-delay error function; wherein the ratio k _A : k _B is selected such that a plot of the code-delay error function has only one positive slope zero- crossing. [0028] The method may further comprise: operating the receiver in a first mode for a first period in which the ratio k _A : k _B is varied until a condition is satisfied that a plot of the code- delay error function, determined based on the combined correlation value Y _C , has only one positive slope zero-crossing; and operating the receiver in a second mode after the condition is satisfied, in which the ratio k _A : k _B has a predetermined value. Preferably, the predetermined value is in a range defined by

[0029] In one embodiment, the ratio k _A : k _B is continuously varied in response to environmental factors, signal strength factors, and/or user dynamics factors. [0030] According to another aspect, there is provided a receiver for processing offset carrier modulated (OCM) ranging signals in a radionavigation system comprising a plurality of satellite-borne transmitters and at least one ground-based receiver, the receiver being comprising: an antenna for receiving a first radionavigation signal from at least one of the plurality of transmitters; and processing circuitry, coupled for receiving the first radionavigation signal, the processing circuitry being operable to perform the method of any of claims 1 to 16 of the appended claims. [0031] According to another aspect, there is provided a recordable, rewritable or storable medium having recorded or stored thereon data defining or transformable into instructions for execution by processing circuitry and corresponding to at least the steps of any of claims 1 to 16 of the appended claims. [0032] According to another aspect, there is provided a server computer incorporating a communications device and a memory device and being adapted for transmission on demand or otherwise of data defining or transformable into instructions for execution by processing circuitry and corresponding to at least the steps of any of claims 1 to 16 of the appended claims. BRIEF DESCRIPTION OF THE DRAWINGS [0033] Embodiments of the invention will now be described by way of reference example to the accompanying drawings, in which: [0034] Figure 1 (PRIOR ART) shows the normalized PSD of a BOC _c (15,2.5) signal; [0035] Figure 2 (PRIOR ART) shows the normalized autocorrelation function of a BOC C(15, 2.5) signal; [0036] Figure 3 (PRIOR ART) shows the normalized non-coherent decision variable

of a BOC _C (15, 2.5) signal; [0037] Figure 4 (PRIOR ART) shows the probability of selecting the correct code delay for a BOCc(15, 2.5) signal, given code-delay values in the range ^30 metres at a 1 metre spacing and coherent integration period of 1 ms; [0038] Figure 5 (PRIOR ART) depicts (a) the coherent code error estimate, and (b) the non- coherent code error estimate, of a BOCc(15, 2.5) signal given an early-to-late correlator spacing of 5 m; [0039] Figure 6 is a schematic block diagram of a receiver according to an embodiment of the invention, illustrating the combined processing of two signals, s _A (t) and s _B (t) as one coherent signal, s _C (t); [0040] Figure 7 shows the normalized PSD of the combined signal s _C (t) formed by the coherently combination of a BOCc(15,2.5) signal and a concentric BOCs(1,1) signal; [0041] Figure 8 shows the normalized autocorrelation function YC the combined signal sC(t); [0042] Figure 9 is a schematic block diagram of a receiver according to another embodiment of the invention, illustrating the combined processing of two signals, s _A (t) and s _B (t) via the weighted linear addition of the corresponding correlator values Y _A and Y _B ; [0043] Figure 10 shows the normalized non-coherent decision variable ^Y _C ^2 of the combined signal s _C (t); [0044] Figure 11 is a plot showing probability of selecting the correct code delay for each of of the first signal, s _A (t), and the combined signal, s _C (t), given code-delay values in the range ^30 metres at a 1 metre spacing and coherent integration period of 1 ms; [0045] Figure 12 shows the coherent code error estimate of (a) the BOCc(15, 2.5) signal and (b) the combined signal s _C (t), given an early-to-late correlator spacing of 5 metres; [0046] Figure 13 shows the non-coherent code error estimate of (a) the BOCc(15, 2.5) signal and (b) the combined signal s _C (t), given an early-to-late correlator spacing of 5 metres; [0047] Figure 14 shows the multipath envelope of (a) the BOCc(15, 2.5) signal and (b) the combined signal s _C (t), assuming an early-to-late correlator spacing of 5 m and a signal specular reflection having a power of -6 dB relative to the line-of-sight signal; and [0048] Figure 15 shows (a) cross correlation, (b) acquisition decision variable, and (c) code- delay estimate of the combined signal s _C (t) for signal weighting ratios ^ k _A , k _B ^ ^ ^at each of 1:100, 1:20, 1:1 and 100:1. DETAILED DESCRIPTION OF EMBODIMENTS [0049] In the following, like numerals will be used to denote like elements. As used herein, the“coherent combination” of two signals is a linear addition of the time-domain signals, as complex numbers, respecting the relative phasing of the signals, as broadcast by the transmitter. [0050] As mentioned above, as many GNSS signals are broadcast from each satellite in a GNSS, it is not uncommon that the centre-frequency of offset-carrier modulated signal coincides with a second signal which either has been modulated with either (a) no sub-carrier, or (b) a sub-carrier of a low frequency. The present disclosure describes a technique for processing offset-carrier modulated signals in the presence of these second signals. This technique eliminates some of the challenges experienced by receivers providing a reduction in the likelihood of side-peak acquisition and a reduced sensitivity to multipath propagation. Thus, it is not necessary that the second signal S _B be modulated by a subcarrier, as the present invention will function and provide the noted improvements in performance if the second signal S _B is not modulated by a subcarrier. If the second signal S _B is modulated by a subcarrier, the present invention will function, however, the improvements will only be achieved if the subcarrier has a frequency lower than the frequency of the subcarrier of the first signal S _A . [0051] Figure 6 is a schematic block diagram of a receiver 600 according to an embodiment of the invention, illustrating the combined processing of two signals, s _A (t) and s _B (t) as one coherent signal, s _C (t). [0052] In an embodiment, the second signal, s _B (t), is broadcast on the same centre frequency as s _A (t). In the present embodiment, another BOC modulation is used as the second signal, s _B (t), although in principle any modulation type could be used for the second signal s _B (t). [0053] Thus, in the present embodiment, the second signal s _B (t) comprises a BOCs _c (1, 1), having the following signal model: where the notation is analogous to that of (1). In the present embodiment, both s _A and s _B are centred at F _A . Advantageously, the present embodiment is based on the receiver processing a single combined signal, following:

[0054] As seen in Fig. 6, antenna 602 receives radionavigation signal which is supplied to downconverting and digitizing module 604 which outputs a digitized (sampled) signal s _A (t). As schematically illustrated, s _A (t) and s _B (t) are effectively coherently combined, whereby a first component C _B (t) of second signal s _B (t) is combined with CDMA spreading sequence component C _A (t) of first signal s _A (t) at first mixer 606, and a second component SC _B (t) of second signal s _B (t) is combined with square wave subcarrier SC _A (t) of first signal s _A (t) at second mixer 608. The resulting coherently combined signal 610 is fed to first integrate and dump module 612, at which the correlation with a local replica signal is performed, to derive the correlator value Y _C of the combined signal s _C (t). In embodiments, the signals S _A and S _B are separate and different, most notably in that they may use a different CDMA spreading code, such that C _A and C _B are different. In embodiments, in the present invention two genuinely different signals are used, which may have two different carriers, two different spreading codes, and/or two different data modulations (if they are present). [0055] Following (2), the correlator values Y _C computed for this combined signal are generated via:

where k _A and k _B are weighting factors, and

as depicted in Fig.6. Thus, signal component S _A (t) is correlated with components C _A and SC _A , while the signal component S _B (t) is correlated with components C _B and SC _B . [0056] For simplicity and for purposes of illustration, in the present embodiment, the nominal received power for s _A (t) and s _B (t), i.e. P _A and P _B , are equal; however, this need not necessarily be the case. Also, in the present embodiment, k _A = k _B . Cases where k _A ≠k _B are discussed later in this disclosure. [0057] The inventors have discovered that an improvement over the receiver processing performance over the techniques shown in Figs 1 to 5 can be attained by combining the received OCM signal with another signal that has the same, or nearby centre frequency. [0058] In one embodiment, the PSD of the second signal occupies the bandwidth contained between the two lobes of the PSD of the OCM signals. This generally implies that the centre frequencies of the two signals should differ by no more than the sum of the sub-carriers of the two signals. For example, if the (PSDs of) signals A and B have centre frequencies , respectively, and those signals have sub-carrier frequencies , respectively, then the most pronounced improvements are achieved when ; however, the techniques according to embodiments of the invention still provide a significant improvement when the following, less restrictive, condition is satisfied: [0059] It is to be noted, however, that the requirements presented in (5) and (6) represent conditions which provide optimal or near optimal performance, but do not represent absolute or mandatory requirements. To demonstrate these potential improvements, another example is discussed in the following. [0060] Figure 7 shows the normalized PSD of the combined signal s _C (t) formed by the coherently combination of a BOCc(15,2.5) signal (s _A ) and a concentric BOCs(1,1) signal (s _B ), along with that for the signal s _A . It is to be noted that the addition of the second signal concentrates a significant amount of power near the signal centre frequency. [0061] Figure 8 shows the normalized autocorrelation function YC the combined signal sC(t), along with that for the signal s _A . It is apparent that, as a result of the use of the combination, that the signal autocorrelation function, Y _C , has a positive bias and has excursions below zero of lower magnitude than that of s _A . [0062] Figure 9 is a schematic block diagram of a receiver according to another embodiment of the invention, illustrating the combined processing of two signals, s _A (t) and s _B (t) via the weighted linear addition of the corresponding correlator values YA and YB. It is to be noted that, as the correlation process is linear, the combination of the two signal components can also be done after the correlation operation, according to the alternative combination method depicted in Fig.9. [0063] More particularly, antenna 602 receives radionavigation signal which is supplied to downconverting and digitizing module 904, which outputs a digitized (sampled) signal s _A (t) at 905, as well as s _B (t) at 907. As schematically illustrated, s _A (t) is effectively combined with CDMA spreading sequence component C _A (t) of first signal s _A (t) at third mixer 906, and is combined with square wave subcarrier SC _A (t) of first signal s _A (t) at fourth mixer 908. The resulting combined signal 910 is fed to second integrate and dump module 912, at which the correlation with a local replica signal is performed, to derive the first correlator value Y _A of the first signal s _A (t). [0064] As schematically illustrated, s _B (t) is effectively combined with a first component C _B (t) of second signal s _B (t) at fifth mixer 914, and is combined with square wave subcarrier SC _B (t) of first signal s _B (t) at sixth mixer 916. The resulting combined signal 918 is fed to third integrate and dump module 920, at which the correlation with a local replica signal is performed, to derive the correlator value Y _B of the first signal s _B (t). [0065] Next, at first correlator value Y _A is multiplied by first weighting factor k _A at first amplifier 922, and second correlator value Y _B is multiplied by second weighting factor k _B at second amplifier 924. [0066] Finally, the weighted outputs, k _A Y _A and are k _B Y _B coherently combined at combiner 926, producing combined correlator value Y _C . [0067] Figure 10 shows (a) the normalized non-coherent decision variable of the first signal s and (b) the normalized non-coherent decision variable of the combined signal s _C (t) (coherently combined BOCc(15,2.5) signal and a concentric BOCs(1,1) signal). This illustrates the advantageous reduction in acquisition ambiguity, according to embodiments of the invention. [0068] Although the autocorrelation function YC of the combined signal sC(t) is similar in complexity and number of vertices, to that of the BOC _c (15, 2.5) signal s _A (t), the positive bias and lack of large negative excursions means that the square magnitude of the autocorrelation function is quite different. As is apparent from Fig.10, wherein the square magnitudes of each of the autocorrelation functions and are depicted, the combined signal s _C (t) results in a function which has far fewer local maxima, reducing in acquisition ambiguity. [0069] Reference is made again to the problems mentioned hereinabove– that the signal has been acquired by detecting and tracking each of its components parts, the upper, lower and now central lobes, separately. Given this coarse acquisition estimate, a receiver may begin to track the individual signals to refine the delay and frequency alignment and, subsequently, attempt a fine acquisition of the composite signal, s _C . [0070] Figure 11 is a plot showing probability of selecting the correct code delay for combined signal, s _C (t), given code-delay values in the range ^30 metres at a 1 metre spacing and coherent integration period of 1 m, for a selection of received C/N ₀ values. Included also, for comparison, are the results for the BOCc(15, 2.5)-only case (s _A (t)). [0071] It is worth nothing that two factors are involved. Firstly, the inclusion of the second signal component, s _B increases the received signal power by a factor of two, assuming that P _A = P _B . Thus, it is reasonable to assume that the detection probability curves should be similar in shape, but that that of the combined signal should be shifted by approximately 3 dB. The second factor is that there are far fewer local maxima (in Y _C ) in the combined case, in this embodiment, approximately half as many. For this reason, the receiver is significantly less likely to mistakenly designate a local maxima as the global maximum. [0072] The plots in Fig.11 demonstrate the improvement that can be achieved by combining both signals coherently. Indeed, in some cases the improvement is of the order of 5 dB. Advantageously, as the locations of the local maxima are further from the global maximum, appropriate weighting of ^Y _C ^2 (selection of k _A and k _B ) based on an a priori error distribution yields more improvement in the combined case than in the case of the BOCc(15, 2.5) signal alone. [0073] Embodiments of the inventions also provide reduced tracking ambiguity. In embodiments, the method of processing the first signal s _A (t) (BOCc(15, 2.5)) by coherently combining it with a concentric second signal s _B (t) (BOCs(1, 1)) can also provide some improvements in the code-delay tracking performance. As with acquisition ambiguity, the reduced complexity of the square magnitude of the autocorrelation function contributes to this improvement. [0074] Figure 12 shows the coherent code error estimate of (a) the BOCc(15, 2.5) signal and (b) the combined signal s _C (t), given an early-to-late correlator spacing of 5 m. Figure 13 shows the non-coherent code error estimate of (a) the BOCc(15, 2.5) signal and (b) the combined signal s _C (t), given an early-to-late correlator spacing of 5 m. As can be seen in Fig. 12, the coherent discriminator when applied to the combined signal can provide similar performance to that of the traditional, BOCc(15, 2.5)-only, case. However, in the non- coherent case, the addition of the second signal, s _B , reduces the number of false-lock points by a factor of two, as illustrated in Fig.13. This can significantly improve the robustness in harsh propagation environments. [0075] One consideration in receiver systems is the multipath envelope. Figure 14 shows the multipath envelope of (a) the BOCc(15, 2.5) signal and (b) the combined signal s _C (t), assuming an early-to-late correlator spacing of 5 m and a signal specular reflection having a power of -6 dB relative to the line-of-sight signal. [0076] In embodiments, despite changing the characteristics of the code-delay estimator, the performance of the combined signal s _C (t) is similar to that of the BOCc(15, 2.5) (s _A (t)) when processed alone. One common way of assessing the sensitivity of a ranging signal to multipath propagation is to examine its, so-called, multipath envelope. Here, a single specular multipath reflection is considered. It is assumed that it arrives at the receiver with a power equal to one quarter (-6 dB) of that of the line-of sight signal. A range of relative delays of the multipath signal are considered, and the largest excursions of the central zero-crossing of code-delay estimate is found. Figure 14 depicts this multipath envelope for both the BOCc(15, 2.5) signal (s _A (t)) and the combined processing of BOCc(15, 2.5) signal and a concentric BOCs(1, 1) signal (s _C (t)), wherein it can be seen that the multipath envelopes are almost identical in shape and magnitude. [0077] Advantageous embodiments of the type of signal combining discussed above enable a user to define the signal correlation properties. As indicated in (11), the combined correlator value (Y _C ) is formed as the weighted sum of the correlation of each of the signal components, s _A and s _B . By manipulating the weighting factors the user can achieve a variety of different correlation properties. These can be exploited both in the acquisition and the tracking stages. Briefly, they can be summarized as follows. [0078] Figure 15 shows (a) cross correlation, (b) acquisition decision variable, and (c) code- delay estimate of the combined signal s _C (t) for signal weighting ratios at each of

1:100, 1:20, 1:1 and 100:1. The autocorrelation function can be manipulated, as depicted in Figure 15(a), to be that of either signal component, or anything in between. In this case, it corresponds to approximately that of a BOCs(1, 1) signal for a ratio

through various forms that resemble a Composite Binary Offset Carrier (CBOC) modulation, to finally assuming approximately that of a B signal for a ratio ^k k ^

These various weightings can be leveraged in the acquisition stage, where the cases of k _B ≥ k _A have fewer local maxima, as depicted in Figure 15(b), thereby reducing the likelihood of side-peak acquisition. [0079] In embodiments, in the tracking domain also, it is possible to exploit the above technique to produce various modulation properties, depending on the application. It may be desirable for the user to avail themselves of the low ambiguity of the BOCs(1, 1) component, under certain circumstances, while the high precision offered by the BOCs(15, 2.5) may be desirable in others. Embodiments allow a user to vary of select a particular weighting configuration. [0080] Embodiments also provide a method of finding the stable lock point that corresponds to the true signal delay. In one embodiment, by selecting certain weighting options, it is possible to ensure that the code-delay error function has only one positive slope zero-crossing. In the particular embodiment chosen here, this corresponds to the case of k _B ≥ 20 k _A , as depicted in Figure 15(c). In addition, in one embodiment, once the appropriate zero-crossing has been identified, the ratio can be reduced to unity, or beyond that to that of the BOCs(15, 2.5) signal alone. [0081] In embodiments, continuously adaptive variation of the factors in response

to factors such as environment, signal strength, and user dynamics may be performed (e.g. by the user), both in the acquisition and tracking phases. [0082] In summary, there is presented herein, at least in embodiments, a novel method of processing an OCM signal as part of a coherent combination another synchronously broadcast signal having a nearby centre frequency. The technique provides a number of benefits to a user, including: (i) reduced acquisition ambiguity; (ii) reduced likelihood of false code-lock; (iii) improved sensitivity; and (iv) user-side tuning of the signal correlation properties. An example embodiment has been presented including a BOCc(15, 2.5) signal and a concentric BOCs(1, 1) signal, and the particular results, and corresponding receiver parameters provided, are specific to that embodiment. However, the concept of coherently combining multiple signals in such a manner can, of course, be extended to any selection of two or more appropriate signals. [0083] While embodiments have been described by reference to embodiments having various components in their respective implementations, it will be appreciated that other embodiments make use of other combinations and permutations of these and other components. [0084] Furthermore, some of the embodiments are described herein as a method or combination of elements of a method that can be implemented by a processor of a computer system or by other means of carrying out the function. Thus, a processor with the necessary instructions for carrying out such a method or element of a method forms a means for carrying out the method or element of a method. Furthermore, an element described herein of an apparatus embodiment is an example of a means for carrying out the function performed by the element for the purpose of carrying out the invention. [0085] In the description provided herein, numerous specific details are set forth. However, it is understood that embodiments of the invention may be practiced without these specific details. In other instances, well-known methods, structures and techniques have not been shown in detail in order not to obscure an understanding of this description. [0086] Thus, while there has been described what are believed to be the preferred embodiments of the invention, those skilled in the art will recognize that other and further modifications may be made thereto without departing from the spirit and scope of the invention, and it is intended to claim all such changes and modifications as fall within the scope of the invention. For example, any formulas given above are merely representative of procedures that may be used. Functionality may be added or deleted from the block diagrams and operations may be interchanged among functional blocks. Steps may be added or deleted to methods described within the scope of the present invention.