POSITION SENSING APPARATUS AND METHOD

This invention relates to a position sensing apparatus and method for generating information indicative of the relative position between two members.

Various inductive sensors are known for detenmning the relative position between two members. hi these inductive sensors, the electromagnetic coupling between a transmit aerial and a receive aerial, typically via an intermediate coupling element, varies along a measurement path. Therefore, the signal induced in the receive aerial in response to an excitation signal being applied to the transmit aerial is indicative of the relative position of the first and second members.

A problem with such inductive sensors is that there are a number of other factors which can affect the electromagnetic coupling between the transmit aerial and the receive aerial, and this can affect the accuracy of the position measurement. To address this problem, various ratiometric techniques have been utilised in which the strength of two electromagnetic couplings is compared.

International Patent Publication WO 95/31696 discusses a position sensor in which a transmit winding, a first receive winding and a second receive winding are formed on a first member and a resonant circuit is formed on a second member. The electromagnetic coupling between the transmit winding and the first receive winding via the resonant circuit varies with the relative position of the first and second members in accordance with a first sinusoidal function, whereas the electromagnetic coupling between the transmit winding and the second receive winding via the resonant circuit varies with the relative position of the first and second members in accordance with a second sinusoidal function which is a quarter of a cycle out of phase with the first sinusoidal function. A signal representative of the relative position between the first and second members is determined by calculating the arctangent of the ratio between the amplitude of the signal

induced in the first receive winding and the amplitude of the signal induced in the second receive winding in response to a drive signal at the resonant frequency of .the resonant circuit being applied to the transmit winding. However, this calculation is not straightforward to implement. International patent publication WO 03/038379 discusses an inductive position sensor in which first and second transmit windings and a receive winding are formed on a first member and a resonant circuit is formed on a second member. The first transmit winding is arranged to generate a magnetic field having a component which varies along a measurement direction in accordance with a first sinusoidal function, and the second transmit winding is arranged to generate a magnetic field having a component which varies along the measurement direction in accordance with a second sinusoidal function which is one quarter of a cycle out of phase with the first sinusoidal function. By applying respective oscillating excitation signals to the first and second transmit windings which are one quarter of a cycle out of phase with each other, a signal is induced in the receive winding having a phase which is indicative of the relative position of the first and second members.

In order to be able to measure the phase of the induced signal in a straightforward manner, the frequency of the oscillating signal needs to be relatively low, whereas for high signal strength the frequency of the oscillating signal needs to be relatively high. The position sensor of WO 03/038379 addresses these conflicting requirements by using an oscillating signal at a relatively low frequency to modulate a carrier signal at a relatively high frequency, with the frequency of the carrier signal being set so as to induce a resonant signal in the resonant circuit.

While the performance of the position sensor of WO 03/038379 is high, the excitation signal generation and the detected signal processing is fairly complicated and accordingly quite expensive to implement. In accordance with an aspect of the present invention, there is provided a position sensor in which a first aerial and a second aerial are fixed

relative to a first member and a magnetic field generator is fixed relative to a second member. An excitation signal generator generates an excitation signal which causes the magnetic field generator to generate a magnetic field which induces a signal in the first aerial indicative of the relative position between the first and second members. The excitation signal generator includes a feedback controller which varies the excitation signal in accordance with a signal representative of the electromagnetic coupling between the magnetic field generator and the second aerial.

Other aspects of the invention are set out in the accompanying claims. Exemplary embodiments of the present invention will now be described with reference to the accompanying figures in which:

Figure 1 schematically shows a perspective view of a position sensor according to a first embodiment of the invention;

Figure 2 shows the main components of the position sensor illustrated in Figure 1;

Figure 3 shows in more detail the conductive tracks formed on a first printed circuit board forming part of the position sensor illustrated in Figure

1;

Figure 4A shows the layout of a transmit aerial formed on the first printed circuit board;

Figure 4B shows the layout of a receive aerial formed on the first printed circuit board;

Figure 4C shows the layout of a feedback aerial formed on the first printed circuit board; and Figure 5 shows in more detail the conductive tracks formed on a second printed circuit board forming part of the position sensor illustrated in Figure 1 ;

Figure 6 is a circuit diagram showing in more detail an oscillator forming part of the position sensor illustrated in Figure 1 ; Figure 7 is a circuit diagram showing in more detail a divide-by-four circuit forming part of the position sensor illustrated in Figure 1 ;

Figure 8 is a circuit diagram showing in more detail a feedback loop employed in the position sensor illustrated in Figure 1 ;

Figure 9 is a circuit diagram showing in more detail circuitry employed to generate a position-dependent signal in the position sensor illustrated in Figure 1 ;

Figure 10 shows the main components of a position sensor forming a second embodiment of the invention;

Figure 11 shows in more detail components of drive circuitry forming part of the position sensor illustrated in Figure 10; Figure 12 shows the main components of a position sensor forming a third embodiment of the invention; and

Figure 13 shows alternative drive circuitry for the first embodiment of the invention.

FIRST EMBODIMENT System Overview

Figure 1 schematically shows a position sensor for sensing the position of a sensor element 1 which is slidably movable relative to a support 3 to allow linear movement along a measurement direction (the direction X in Figure 1). A first printed circuit board (PCB) 5 extends along the measurement direction adjacent to the support 3 and has thereon conductive tracks which form a receive aerial 7, a transmit aerial 9 and a feedback aerial 11, each of which are connected to a control unit 13. In this exemplary embodiment, a display 15 is also connected to the control unit 13 for displaying a number representative of the position of the sensor element 1 along the support 3.

As shown in Figure 1 , the first PCB 5 is generally rectangular in shape with the lengthwise axis aligned with the measurement direction and the widthwise axis aligned perpendicular to the measurement direction. The receive aerial 1, the transmit aerial 9 and the feedback aerial 11 are connected to the control unit 13 via a lengthwise edjge of the first PCB 5, which

corresponds to the position value of x equals zero, with the position value increasing along the length of the first PCB 5 from the lengthwise edge corresponding to x equals zero.

Figure 2 schematically shows the main components of the position sensor illustrated in Figure 1. As shown, an oscillator 21 generates an oscillating signal at 16 MHz. The signal output by the oscillator is input to a conventional divide-by-four circuit 23 which outputs four signals at 4 MHz, namely: a first signal (hereafter called the 0° signal); - a second signal (hereafter called the 90° signal) which lags the first signal by a quarter of a cycle; a third signal (hereafter called the 180° signal) which lags the first signal by half a cycle; and a fourth signal (hereafter called the 270° signal) which lags the first signal by three-quarters of a cycle.

The 0° and 180° signals output by the divide-by- four circuit 23 are input to drive circuitry 25 which generates an excitation signal which is applied to the transmit aerial 9, which results in the generation of a magnetic field which oscillates at 4MHz. As will be described in more detail hereafter, in accordance with the invention, the amplitude of the excitation signal is controlled by a feedback loop.

In this embodiment, the sensor element 1 has a resonant circuit 27 which is formed by an inductive winding 29 and a capacitor 31 mounted on a second PCB. The resonant frequency of the resonant circuit is at 4MHz, and accordingly the oscillating magnetic field generated by the transmit aerial induces an oscillating signal at 4MHz in the resonant circuit 27. As those skilled in the art will appreciate, the resonant signal induced in the resonant circuit 27 is shifted in phase by a quarter of a cycle from that of the oscillating magnetic field. The resonant signal induced in the resonant circuit 27 itself generates an oscillating magnetic field, and this oscillating magnetic field induces

signals in the receive aerial 7 and the feedback aerial 11. In this embodiment, the amplitude of the oscillating signal induced in the feedback aerial 11 by the resonant signal in the resonant circuit 27 does not vary with the position of the sensor element 1 along the measurement direction, while the amplitude of the of the oscillating signal induced in the receive aerial 7 by the resonant signal in the resonant circuit 27 varies linearly in accordance with the position of the sensor element 1 along the measurement direction.

Although the signal induced in the feedback aerial 11 does not vary in accordance with the position of the sensor element 1 along the measurement direction, it may vary for a number of different factors including: variations in the distance between the sensor element 1 and the first PCB 5 in a direction perpendicular to the measurement direction ofthe first PCB 5; variations in the frequency of the ceramic resonator in the oscillator 21; variation in the amplitude of the current supplied to the transmit aerial 9 as a result of variation in the impedance of the drive circuitry 25 and the impedance of the conductive track forming the transmit aerial with environmental factors such as temperature; and - variations in the resonant properties of the resonant circuit 27 with changing environmental factors, e.g. temperature, humidity.

In a similar fashion, these factors will cause a variation in the amplitude of the signal induced in the receive aerial 7, and uncorrected those variations would lead to inaccuracy in the measurement reading. Accordingly, in this embodiment a feedback control loop is used to vary the amplitude of the excitation signal to cause the amplitude of the signal induced in the feedback aerial 11 to be set at a reference level.

As shown in Figure 2, the signal induced in the feedback aerial 11 is input to a synchronous detector 33 which performs synchronous detection at the excitation frequency using the 90° and 270° outputs of the divide-by-four circuit 23 in order to take into account the phase shift introduced by the

resonant circuit 27. The output of the synchronous detector 33 approximates a full-wave rectified signal, and is passed through a low pass filter 35 to generate a DC signal corresponding to the amplitude of the signal induced in the feedback winding. The output of the low pass filter 35 is input to a servo-system 37 along with a reference voltage level generated by a reference voltage level generator 39. The output of the servo-system 37 is input to the drive circuitry 25 to control the amplitude of the excitation signal applied to the transmit aerial 9. hi operation, the servo-system 37 controls the amplitude of the excitation signal so that the signal output by the low pass filter 35 matches the reference voltage level, thereby forming a feedback control loop.

As a result of this feedback control, the amplitude of the signal induced in the receive aerial 7 is predominantly dependent on the position of the sensor element along the measurement direction. As shown in Figure 2, the signal induced in the receive aerial 7 is input to a synchronous detector 41 which performs synchronous detection at the excitation frequency using the 90° and 270° outputs of the divide-by-four circuit 23 in order to take into account the phase shift introduced by the resonant circuit 27. The output of the synchronous detector 41 is then input to a low-pass filter and buffer 43, whose output is a DC signal having an amplitude which is dependent on the position of the sensor element 1 along the measurement direction.

In the exemplary embodiment, the signal output by the low pass filter and buffer 43 is converted to a numerical value which is then displayed on the display 15. The layouts of the receive aerial 7, the transmit aerial 9, the feedback aerial 11 and the resonant circuit 27 will now be described in more detail, followed by a more detailed description of the main components of the circuitry.

Aerial Layouts

Figure 3 shows the layout of the conductive tracks on the first PCB 5 which form the receive aerial 7, the transmit aerial 9 and the feedback aerial 11. hi this embodiment, the first PCB 5 is a multi-layer PCB with conductive tracks being deposited on both planar surfaces (referred to hereafter as the top surface and the bottom surface for ease of reference), with connection between the conductive tracks on the top and bottom surfaces occurring through via holes. In Figure 3 and the following Figures, conductive tracks deposited on the top surface are shown by solid lines whereas conductive tracks deposited on the bottom surface are shown by dashed lines.

For ease of explanation, Figures 4A to 4C respectively show just the conductive tracks forming the transmit aerial 9, the receive aerial 7 and the feedback aerial 11. As shown in Figures 4A to 4C, the transmit aerial 9 is almost exclusively formed by conductive tracks deposited on the bottom surface of the first PCB 5, whereas the receive aerial 7 and the feedback aerial 11 are predominantly formed on the top surface of the first PCB 5.

The transmit aerial 9 has a first set of three current loops around the periphery of the first PCB 5, with a second set of three current loops in the same sense as the first set positioned adjacent the widthwise edge of the first PCB 5 away from the widthwise edge corresponding to x equals zero. The second set of three current loops extend over about one fifth of the length of the first PCB 5.

The feedback aerial 11 has a first set of three current loops which extend from the widthwise edge of the first PCB 5 corresponding to x equals zero over about four fifths of the length of the first PCB 5, and a second set of six current loops in the opposite sense to the first set of three current loops which extend over the remaining fifth of the length of the first PCB 5 so that they are substantially aligned with the second set of three current loops of the transmit aerial. In this way, the transmit aerial 9 and the feedback aerial 11 are balanced (i.e. an alternating electromagnetic field generated by the

transmit aerial 9 directly induces substantially no signal in the feedback aerial 11) in a manner which is well known to those skilled in the art.

The receive aerial 7 extends over about four fifths of the length of the first PCB 5 from the widthwise edge corresponding to x equals zero. The pattern of the conductive track of the receive aerial 7 is such that any perpendicular electromagnetic field component close to the position x equals zero passes through six current loops in one sense, and then moving in the measurement direction without changing sense the number of current loops reduces from five, then four, then three, then two and then one until finally at the central point of the length of the receive aerial 7 any perpendicular electromagnetic field component passes through no current loops. Continuing along the measurement direction, the sense of the current loops changes and the number of current loops increase from one, to two, to three, to four, to five and finally to six. The particular arrangement of the receive aerial 7 results in the signal induced in the receive aerial 7 varying in dependence on the position of the sensor element 1 in a linear manner from a maximum positive value adjacent to x equals zero, reducing to zero in the middle of the receive winding, and then reducing further to a maximum negative value at the end of the receive aerial 7 away from x equals zero. hi addition, the anti-symmetric arrangement of the current loops of the receive aerial 7 leads to the receive aerial 7 being substantially balanced with respect to the transmit aerial 9.

For completeness, Figure 5 shows that the inductor 29 of the resonant circuit 27 of the sensor element 1 is formed by conductive track on the second PCB arranged in a concentric set of current loops. The ends of the conductive track are joined by the capacitor 31 (not shown in Figure 5).

Circuitry Components

As shown in Figure 6, the oscillator 21 employs a standard oscillator circuit, hi this embodiment, the oscillator circuit includes a ceramic oscillator

51, which combines both good stability and relatively low cost. As shown in

Figure 7, the divide-by- four circuit 23 is a conventional circuit using two flip- flops 61a,61b. The outputs of the flip-flop 61a supply the 90° and 270° signals, while the outputs of flip-flop 61b supply the 0° and 180° signals.

Figure 8 shows the circuitry within the feedback loop from the feedback aerial 11 to the drive circuitry 25 which supplies the excitation signal to the transmit aerial 9. As shown, the signal induced in the feedback aerial 11 is input to a cross-over analogue switch which forms the synchronous detector 33. As discussed above, the cross-over analogue switch is controlled by the 90° signal and 270° signal to take account of the phase shift introduced by the resonant circuit 27.

The output of the synchronous detector 33 is input, via the low pass filter 35, to the inverting input of an operational amplifier forming part of the servo-system 37. A potential divider is used as the reference voltage level generator 39, and the resultant reference voltage level is input to the non- inverting input of the operational amplifier of the servo-system 37.

As those skilled in the art will recognise, the servo system 37 is effectively a P-I-D (proportional - integral - derivative) controller in which the integral component dominates. In other words, the historical variation between the signals input to the non-inverting and inverting inputs of the operational amplifier largely determine any change in the output of the servo- system 37. For the particular components used in this embodiment, this was found to produce good results.

The output of the servo-system 37 supplies the power rail for the drive circuitry 25, which as discussed above supplies an oscillating excitation signal to the transmit aerial 9 in accordance with the 0° and 180° signals. The drive circuitry itself is a well-known arrangement employing a pair of MOSFETS.

In use, if the signal induced in the feedback aerial 11 exceeds a desired level, then the signal input to the inverting input of the operational amplifier of the servo system 37 will rise above the reference voltage level. This leads to the output of the servo system 37 dropping, resulting in a decrease in the amplitude of the oscillating excitation signal a'nd accordingly a decrease in the

amplitude of the signal induced in the feedback aerial 11. Similarly, if the signal induced in the feedback aerial 11 falls below a desired level, then the signal input to the inverting input of the operational amplifier of the servo system 37 will drop below the reference voltage level. This leads to the output of the servo system 37 increasing, resulting in an increase in the amplitude of the oscillating excitation signal and accordingly an increase in the amplitude of the signal induced in the feedback aerial 11.

Figure 9 shows the circuitry used to process the signal from the receive aerial 7. As shown, the signal induced in the receive aerial 7 is input to the synchronous detector 41, which in this embodiment comprises an analogue cross-over switch. As discussed above, the analogue cross-over switch is driven by the 90° and 270° signals to take account of the phase shift introduced by the resonant circuit 27. The potential divider circuit in the synchronous detector 41 is used to set a virtual ground about which the signal received from the receive aerial 7 oscillates.

The output of the synchronous detector 41 is input to a low pass filter and buffer 43, which outputs a DC signal which varies in accordance with the position of the sensor element along the measurement direction.

SECOND EMBODIMENT

A second embodiment of the invention will now be described with reference to Figures 10 and 11. In the second embodiment, the drive circuitry of the first embodiment is modified so as to drive the transmit aerial with a substantially sinusoidal signal with a view to reducing EMC emissions and increasing efficiency. Potentially this approach may also provide improved accuracy due to the reduction in harmonic frequencies.

As shown in Figure 10, in this embodiment an oscillator 51 provides a 4 MHz signal to drive circuitry 53, and the drive circuitry 53 provides the 90° and 270° signals to the synchronous detectors 33 and 41. Apart from the oscillator 51 and the drive circuitry 53, the remaining components are the

same as the corresponding components in the first embodiment and will not, therefore, be described again in detail.

In this embodiment, the oscillator 51 outputs a sine wave signal at 4 MHz, although it is not essential for the output of the oscillator 51 to be sinusoidal to achieve correct operation.

Figure 11 is a circuit diagram showing components of the drive circuitry 53. The signal from the oscillator 51 is input to the terminal labelled osc_in and is attenuated using a MOSFET Ml which is arranged to act as a voltage controlled resistor (VCR) whose resistance is varied in accordance with a control signal received at the VDDJTX terminal. The control signal applied to the VDD_TX terminal corresponds to a sign inversion of the signal output by the servo system 37.

Although the voltage control of the VCR is non-linear, this is not a problem in the context of a feedback system. For the same reason, the fact that the turn-on voltage of the MOSFET Ml in the VCR will vary from device to device is not a problem.

The attenuated signal is supplied to the base of an n-type transistor Tl which forms part of a current amplifier arrangement. The capacitor C9 connected to the emitter ensures high gain for alternating currents, while the resistor R20 sets a DC bias level. The transmit aerial 9 and a capacitor C5 are connected in parallel to the collector of the n-type transistor. The capacitance of the transistor C5 is set for resonance at the carrier frequency (i.e. 4MHz) to improve the sinusoidal nature of the current flowing through the transmit aerial 9. The collector of the n-type transistor is also connected, via capacitor

C6 and resistor R12, to the non-inverting input of a comparator Ul with a DC bias voltage set by resistors R23 and R24. A reference voltage, set by resistors R21 and R22, substantially equal to that DC bias voltage is applied to the inverting input of the comparator Ul so that the output of the comparator Ul is a square wave having a fundamental frequency equal to the frequency of the carrier signal. This square Wave output by the comparator

Ul is then processed in a straightforward manner by components not shown in Figure 11 to generate the 90° and 270° signals which are supplied to the synchronous detectors 33 and 41.

THIRD EMBODIMENT

A third embodiment will now be described with reference to Figure 12 in which the transmit aerial 9 also functions as the feedback aerial, and therefore a separate feedback aerial is not needed, hi the absence of a separate feedback aerial, the balancing coils at one end of the transmit aerial can be omitted thereby reducing the area of printed circuit board required. hi this embodiment, by placing the transmit aerial 9 in one arm of a Wheatstone bridge arrangement, the signal across the centre of the Wheatstone bridge is representative of the electromagnetic coupling between the resonator 27 and the transmit aerial 9. As shown in Figure 12, in this embodiment the drive circuitry applies the excitation signal to the top of a Wheatstone bridge 61. The top two arms of the Wheatstone bridge 61 are respectively formed by the transmit aerial 9 and a reference inductor 63 having the same inductance and resistance as the transmit aerial 9. The lower two arms of the Wheatstone bridge 61 are also respectively formed by reference inductors 65,67 having the same inductance and resistance as the transmit aerial 9. In this embodiment, the three reference inductors 63,65,67 are off-the-shelf components.

The signal across the centre of the Wheatstone bridge 61 is input to a differential amplifier 69, and the output of the differential amplifier 69 is input to the synchronous detector 33. Apart from the removal of the feedback aerial and the balancing coils of the transmit aerial, putting the transmit aerial in a Wheatstone bridge 61 and the addition of the differential amplifier 69, the remaining electrical components are the same as the corresponding components in the first embodiment. In the absence of the resonator 27, the Wheatstone bridge 61 is balanced and therefore no significant signal is input into the differential

amplifier 69. As the resonator 27 approaches the transmit aerial 9, the impedance of the transmit aerial 9 changes due to interaction with the resonator 27, thereby unbalancing the Wheatstone bridge causing a signal to be present across the centre of the Wheatstone bridge which is phase-shifted by 90° from the excitation signal. Therefore, the signal across the centre of the Wheatstone bridge has analogous properties to the signal induced in the feedback aerial 11 in the first embodiment.

By removing the need for a feedback aerial, the design of the position sensor is simplified. In particular, the three reference inductors 63,65,67 can be discrete inductors which are simpler and less expensive to implement.

MODIFICATIONS AND FURTHER EMBODIMENTS

In an alternative version of the first embodiment, the transmit aerial is made resonant at the carrier frequency by connecting a capacitor in parallel with the transmit aerial 9. In this way, the efficiency is improved and the EMC performance is improved by suppressing harmonics of the resonant frequency. Such an arrangement is shown in Figure 13. As shown, the drive circuitry 81 includes resistors 83 and 85 to make the FET stage have higher impedance, and the capacitor 87 which is selected to make the transmit aerial 9 resonant at 4 MHz. The position of the resistors 83 and 85 allows maximum turn-on of the FETs while minimising shoot-through current. The resistances of the resistors 83 and 85 are chosen for optimal efficiency and satisfactory coil current drive amplitudes. La this embodiment, the capacitor is preferably a COG type capacitor in order to prevent excessive energy loss. In the third embodiment, the lower two anus of the Wheatstone bridge

61 respectively have reference inductors 65,67. Alternatively, these reference inductors 65,67 could be replaced by capacitors having the identical capacitances, which results in a reduction of the drive voltage required to achieve the desired current flow. In addition, replacing the inductors by capacitors typically reduces the component cost. Further, an additional

external capacitor could be provided in parallel with the Wheatstone bridge 61 to cause resonance at the carrier frequency.

In the described embodiments, the transmit aerial 9 is designed so that an alternating electromagnetic field generated by the transmit aerial 9 does not directly induce a signal in the receive aerial 7. In an alternative embodiment a proportion of the reference voltage level determined by the amplitude of the transmission drive signal is subtracted. Ih this way, any signal breakthrough proportional to the transmission drive signal may be compensated.

In the first embodiment, a feedback loop controls the amplitude of a drive signal applied to the transmit aerial 9 at a substantially fixed frequency. Alternatively, a voltage-controlled oscillator could be used with the feedback loop controlling the frequency of the oscillating signal generated by the voltage-controlled oscillator. In this way, the strength of the signal induced in the resonant circuit would be varied as the frequency of the excitation signal moved relative to the resonant frequency (i.e. increasing as the excitation frequency approaches the resonant frequency and decreasing as the excitation frequency moves away from the resonant frequency), and therefore the strength of the signal induced in the feedback aerial would vary.

In the exemplary embodiments, a passive resonant circuit 27 on the sensor element 1 acts as an intermediate coupling element between the transmit aerial and both the receive aerial and the feedback aerial. Alternatively an active resonant circuit, including a power source for amplification, could be used. A non-resonant intermediate coupling element could alternatively be used, for example a ferrite element, a conductive disk or a simple wire loop. Unlike the resonant circuit, these other intermediate coupling elements do not introduce a 90° phase shift and therefore the circuitry can be slightly simplified. However, the use of a resonant circuit as an intermediate coupling element gives higher signal strength.

The intermediate coupling element could include a non-linear element, such as a diode, so that a magnetic field oscillating at an excitation frequency induces a current in the intermediate coupling element having frequency

components away from the excitation frequency. This would result in a signal being induced in the receive aerial at these new frequency components, which could be detected and have improved noise immunity with respect to the excitation signal generating circuitry. The intermediate coupling element effectively acts as a magnetic field generator which is powered through inductive coupling with the transmit aerial. In an alternative embodiment, a magnetic field generator on the sensor element could be directly powered, with the excitation signal applied being controlled in accordance with the signal induced in the feedback aerial. Such an arrangement has the advantage of removing the need of a transmit aerial, but also has the significant disadvantage of requiring electrical connections to the sensor element.

Two synchronous detectors are employed in the first and second embodiments, one to detect the signal induced in the receive aerial and one to detect the signal induced in the feedback aerial. This allows the feedback loop to work continuously. ha an alternative embodiment, a single feedback/receive aerial could be used in a first mode as a feedback aerial and in a second mode as a receive aerial, the first mode and the second mode being sequentially selected. The amplitude of the drive signal applied in the second mode is determined by the strength of the signal induced in the feedback/receive aerial in the first mode. This requires using in the first mode a first transmit aerial whose electromagnetic coupling via the resonant circuit with the feedback/receive aerial does not vary with the position of the sensor element, and in the second mode a second transmit aerial whose electromagnetic coupling via the resonant circuit with the feedback/receive aerial does vary with position of the sensor element.

Another alternative would be to apply an excitation signal at a first frequency to a first transmit aerial, whose electromagnetic coupling via the resonant circuit with a feedback/receive aerial does not vary with position of the sensor element along the measurement direction, and concurrently to apply an excitation signal at a second frequency (which is different from the

first frequency) to a second transmit aerial whose electromagnetic coupling via the resonant circuit with the feedback/receive aerial does vary with position. In this way, the strength of the signal induced in the feedback receive aerial at the first frequency can be used to adjust the amplitude of the excitation signal at the second frequency applied to the second transmit aerial. Those skilled in the art will appreciate that there are many different possible layouts of the aerials. Those skilled in the art will also recognise that synchronous detection is not essential, and for example could be replaced by a full- wave or half-wave rectifier arrangement. ' While 4MHz has been found to be a convenient frequency of operation, other frequencies could be used. Typically, the operational frequency will be in the range from 1 OkHz to 10MHz. hi the exemplary embodiments described above, the position sensor measures the position of a first member (the sensor element 1) relative to a second member (the first PCB 5) along a rectilinear measurement path. Alternatively, the position sensor could be adapted to measure linear position along a curved measurement path, for example a circle (i.e. a rotary position sensor), by varying the layout of the transmit aerial, the receive aerial and the feedback aerial in a manner which would be apparent to a person skilled in the art. The position sensor could also be used as a speed detector by taking a series of measurements of the position of the first member relative to the second member at known timings.

In the exemplary embodiment, the aerials are formed by conductive tracks on a printed circuit board. Alternatively, a different planar substrate could be used. Further, if the aerials are themselves sufficiently rigid then they could be fixed relative to the first member and the resonant circuit could be fixed relative to the second member without the use of a substrate. It is also not essential that the aerials be planar because, for example, a cylindrical geometry could be used.

Of course, as the position sensor detects the relative position between first and second members, it does not matter which of the first member and the second member are moved, or even if both are moved.