Field This invention relates to circuits, systems, and methods for converting power obtained from a power generator. The circuits, systems, and methods described herein may be used to interface a power generator to a power distribution grid.

Background Grid-connected power generation systems typically include two major parts: power generators that produce the power and converters that receive, condition, and inject the power into the power distribution grid. Power generators include, for example, photovoltaic (PV) cells, fuel cells, and wind turbines.

To increase the overall efficiency of power generators under different circumstances, such as partial shadowing of PV cells, or mismatches between PV cells or wind turbines, independent control and power extraction is required for each power generator. This requires using a separate converter for each power generator. This may be referred to as micro- inverter technology.

In single-phase grid-connected power generation systems, the instantaneous output power oscillates at twice the grid frequency. In systems where the power generators are PV cells, the input power generation is DC and thus oscillation of the instantaneous power at the converter output, if reflected in the input, causes the input operating point to deviate from DC. If there is any power oscillation on the PV cell side, maximum power is only achievable at the peak of oscillation, which translates into less average power extraction than the available maximum power [1,2]. This is a power loss that reduces the efficiency of the PV cell system. Therefore, power pulsation is a key problem in such systems and the PV cell converter should decouple the output power pulsation from the input DC power generation to maximize the efficiency.

As noted above, if there is no power decoupling in a single-phase inverter, the power generation at the PV cell terminal will contain oscillations that result in a deviation from the optimum point. Energy storage in the circuit may supply oscillatory power and reduce power pulsation at the PV cell terminal. The decoupling problem is normally resolved by using large electrolytic capacitors (e.g., in the range of milli-Farads) to minimize the effect of the output power pulsation on the input operating point. However, this is highly undesirable because it decreases the life-time and increases the volume, weight, and cost of the inverter.

Depending on the topology, different locations of the energy storage are possible. For example, for a single-stage topology energy storage may be implemented at the PV cell terminals. For multi-stage topologies, when a voltage source inverter is employed at the output, the power decoupling capacitor may be placed at the input terminals and/or at the DC bus (e.g., between DC-DC converter and DC-AC inverter stages). It is beneficial to have most of the decoupling capacitance on the DC bus because the voltage level is higher and the same amount of energy storage can be achieved with a smaller capacitor.

The generation of a high DC voltage is not efficient and it poses an excessive voltage stress on the inverter and on the output of the first stage. Moreover, the high voltage on the bus enlarges high frequency ripples on the output current, which requires large passive filters for compensation. Further, in such an approach relatively large electrolytic capacitors at the input are still needed to remove any oscillations at the PV cell input.

Use of a voltage source inverter at the output requires a bulky inductor for connection to the grid. To avoid this, a micro-inverter may use an unfolding power circuit in the last stage. However, with this approach, a large electrolytic capacitor bank is still required at the PV cell terminals because the voltage level is very low and the amount of capacitance required becomes large. In general, topologies that use a transformer as an energy buffer employ such a configuration for power decoupling [1,2,5]. To reduce the amount of input capacitance required, a multistage approach may be used as in [3] which processes the full output power. However, this reduces the efficiency and increases the size and weight of the converter. Moreover, since the DC bus voltage is very high the stress on the switches of the converter stages is very high and also the filter that shapes the current becomes relatively large.

In other approaches [5-10], an auxiliary power circuit is introduced that absorbs power and provides energy when needed. As a result a large electrolytic capacitor is not required. The auxiliary power circuit usually operates at high voltage to reduce the energy storage component. Such approaches generally have low efficiency and have high number of power processing stages.

Summary

Described herein is a power converter for use with a power generator, comprising: a high side input point and a low side input point for receiving substantially DC power from the power generator; a capacitor connected across the high side and low side input points; a first means that receives voltage across the capacitor and converts the voltage into a high frequency AC voltage while rejecting or minimizing oscillations in the received voltage; a second means that converts the high frequency AC voltage into a substantially sinusoidal AC voltage or current; an isolating transformer that receives the substantially sinusoidal AC voltage or current; a third means that converts a high frequency substantially sinusoidal AC voltage or current from the transformer into (i) a low frequency substantially sinusoidal AC output voltage or current; and (ii) a DC output voltage or current; and a high side output point and a low side output point for outputting the output voltage or current.

In one embodiment the high frequency AC voltage may be amplitude modulated by a substantially sinusoidal voltage at a frequency that is related to a power distribution grid frequency.

The first means may comprise a chopper circuit. Switches of the chopper circuit may be controlled using pulse width modulation. The second means may comprise a resonant circuit. The resonant circuit may be a series resonant circuit. The third means may comprise a rectifier and optional filter, wherein the output is a DC voltage or current. The third means may comprise a rectifier and an inverter, wherein the output is an AC voltage or current. Also described herein is a micro-inverter for a distributed power generator, comprising: the power converter described above; and a controller that controls operation of switches in the first means by comparing the voltage across the capacitor with a reference voltage so as to minimize voltage fluctuations across the capacitor. The controller may include one or more of a modified pulse width modulator, a duty cycle controller, and a maximum power point tracker. The duty cycle controller may comprise a proportional integral-derivative compensator.

Also described herein is a power generation system, comprising: at least one micro- inverter as described above; and at least one power generator. Each power generator may be connected to a micro-inverter.

Also described herein is a method for obtaining power from a power generator, comprising: converting substantially DC output power from the power generator into a high frequency AC voltage while rejecting or minimizing oscillations in the output power from the power generator; converting the high frequency AC voltage into a high frequency

substantially sinusoidal voltage or current; and converting the high frequency substantially sinusoidal AC voltage or current into (i) a DC voltage or current, and (ii) a low frequency substantially sinusoidal AC voltage or current; wherein the high frequency substantially sinusoidal AC voltage or current is isolated from the DC voltage or current or the low frequency substantially sinusoidal AC voltage or current. In one embodiment the method may include amplitude modulating the high frequency

AC voltage by a substantially sinusoidal voltage at a frequency that is related to a power distribution grid frequency.

The method may comprise using a chopper to convert output power from the power generator into a high frequency AC voltage. The method may comprise using pulse width modulation to control one or more switches of the chopper circuit. The method may include comparing output voltage of the power generator with a reference voltage so as to minimize voltage fluctuations of the power generator. The method may comprise using a resonant circuit to convert the high frequency AC voltage into a high frequency substantially sinusoidal AC voltage or current. The resonant circuit may comprise a series resonant circuit. The method may comprise connecting the low frequency substantially sinusoidal output voltage or current to a power distribution grid. In the above circuits, systems, and methods, the power generator may be a photovoltaic (PV) cell or a fuel cell.

Brief Description of the Drawings

For a better understanding of the invention, and to show more clearly how it may be carried into effect, embodiments will now be described, by way of example, with reference to the accompanying drawings, wherein:

Figure 1 is a generalized block diagram of a micro-inverter.

Figure 2 is a block diagram of a micro-inverter according to one embodiment.

Figure 3(a) is a circuit diagram of a micro-inverter with maximum power point tracking according to one embodiment.

Figure 3(b) is a circuit diagram of a micro-inverter with maximum power point tracking according to another embodiment.

Figure 3(c) is a plot showing operating waveforms of the circuit of Figure 3(b).

Figure 4(a) is a plot showing typical current-voltage and power-voltage characteristics and the maximum power point of a PV cell, and Figure 4(b) is a plot showing how the characteristics change with amount of irradiation.

Figure 5 is a plot showing typical steady-state operating waveforms of the

embodiment of Figure 2.

Figures 6(a) to (f) show schematic diagrams of equivalent circuits of the embodiment of Figure 3(a), for each interval of the plot of Figure 5.

Figure 7 is a circuit diagram showing the kth harmonic equivalent circuit of the circuit in Figure 2.

Figures 8(a) and (b) are plots showing zero voltage switching of the switches Mj (a) and M ₂ (b) of Figure 3(a), obtained from a simulation.

Figure 9 is a plot showing the bus and grid voltage waveforms and the gate signals for the embodiment of Figure 3(a) using modified pulse width modulation as described herein. Figure 10 is a plot showing output grid current waveform with conventional pulse width modulation and with modified pulse width modulation.

Figure 11 is a plot showing zero voltage switching of a simulation based on the circuit of Figure 3(a). Figure 12 is a plot showing bus voltage and resonant tank current and voltage of a simulation based on the circuit of Figure 3(a).

Detailed Description of Embodiments

Described herein are systems, circuits, and methods for obtaining power from a power generator. A power generator may be, for example, a wind turbine, a fuel cell, or a photovoltaic cell. The power generator may be a distributed power generator. Whereas embodiments of the systems, circuits, and methods are described herein primarily with respect to photovoltaic cells, it will be appreciated that the systems, circuits, and methods are not limited thereto. The systems, circuits, and methods described herein may be used in a micro-inverter for a power generator. As used herein, the term "micro-inverter" refers to a device that interfaces a power generator with a load, such as a power distribution grid. A system including a micro-inverter is shown in the generalized block diagram of Figure 1. The micro- inverter 20 receives power from a power generator 10, and outputs power to a load 30. The micro-inverter 20 may include a power section 200 that may include one or more stages and perform one or more functions, such as, for example, DC-DC conversion, DC- AC

conversion, or a combination thereof. The micro-inverter may include a control section 300 that may perform one or more functions such as, for example, maximum power point tracking of the power generator, and/or providing gating signals to the power section 200. The gating signals may be determined by sensing the power generator voltage and/or current, and/or the load voltage and/or current.

A micro-inverter as described herein increases the overall efficiency of power generation under different circumstances and conditions. For example, in the case of a PV cell, partial shadowing of the PV cell or mismatches between PV cells can degrade the overall efficiency of the system. However, use of a micro-inverter for each PV cell, or for a PV cell string or module, permits independent control and power extraction from each PV cell or PV cell string or module, maximizing power extraction from PV cells despite varying conditions of individual PV cells.

A micro-inverter as described herein is compact, so as to be attached to a PV cell (e.g., to the back of a PV cell). Since micro-inverters are exposed to a wide range of environmental conditions, such as extremes of temperature and humidity, reliability and maintenance of are major issues. This exposure also adversely affects the life expectancy and performance of the inverter. These factors demand robust design and construction, and may require more expensive components that lead to a higher manufacturing cost. Consequently, challenges in the design of a micro-inverter include achieving compactness and low cost, e.g., by reducing the number and size of circuit components. Advantageously, a micro-inverter as described herein does not require costly high voltage components and wiring.

In one embodiment, a micro-inverter may include one or more feature such as, for example, communication, inversion (i.e., DC to AC conversion), peak power tracking, islanding, etc. Such a micro-inverter may be integrated into a PV module. The term "PV module" refers to one or more PV cells connected in series, parallel, and/or series-parallel to effect a desired electrical output. For example, a PV module may be the smallest discrete unit provided (e.g., sold) to an end user of a photovoltaic conversion system. Typically, PV modules are supplied with connectors or other suitable hardware to extract the DC output from the module. However, as described herein, a micro-inverter may be integrated into a PV module, wherein such connectors or other suitable hardware to extract the DC output are replaced with a micro-inverter so as to obtain one or more electrical features, such as those described above. Moreover, such integration of the micro-inverter with the PV module may include packaging of the micro-inverter either substantially together or incorporated into the module packaging, or in a manner that substantially unifies the micro-inverter with the module packaging. The PV module packaging may be pre-existing or may be designed to accommodate the micro-inverter. It will be appreciated that the inversion portion of the micro-inverter may be of any design, such as, for example, but not limited to, current source, voltage source, resonant, etc., and may include single or multiple power stages. The control section 300 of a micro-inverter as described herein may include maximum power point tracking (MPPT). As a result, MPPT may be performed on each power generator independently. In the case of PV cells, this is useful where partial shading cannot be avoided since MPPT allows the maximum power to be extracted from each PV cell for any instantaneous condition. MPPT removes any mismatch losses between PV cells in the system. Further, micro-inverters as described herein provide modularity to individual power generators, allowing a "plug and play" approach to their use in a distributed power generation system. The control section may be implemented in whole or in part using discrete components, using digital technology (e.g., in a digital signal processor (DSP), field programmable gate array (FPGA), or application specific integrated circuit (ASIC) device), or using a combination thereof. For example, one or more components of the control section may be implemented in an algorithm using a suitable hardware language such as, for example, very high speed integrated circuit (VHSIC) hardware descriptive language (VHDL), register transfer language (RTL), or Verilog. Such an algorithm may be implemented in, for example, a FPGA or ASIC device, or other suitable logic device. Use of digital technology provides a controller that is compact and robust.

As used herein, the terms "maximum power point tracking (MPPT)" and "maximum power point tracker (MPP tracker)" are distinct. "MPPT" refers to an algorithm and "MPP tracker" refers to hardware (i.e., a circuit). The MPPT calculates the optimum operating point for a distributed power generator such as a photovoltaic cell, and provides a reference point for MPP tracker to steer the system toward the optimum operating point.

As used herein, the term "photovoltaic cell (PV cell)" refers to any cell having a light absorbing material to absorb photons and generate electrons via a photoelectric effect. A non-limiting example of a photovoltaic cell is a solar cell. The light absorbing material may absorb light in any wavelength or combination of wavelengths, including, for example, wavelengths of solar light that reach the earth's surface, and/or wavelengths of solar light beyond the earth's atmosphere. Two or more light absorbing materials having specific wavelengths of light absorption may be used in combination to take advantage of different light absorption and charge separation mechanisms. The light absorbing material may be configured as, for example, bulk material, thin-film (e.g., inorganic layers, organic dyes, and organic polymers), and/or nanocrystals. Photovoltaic cells may be combined into arrays, strings, modules, or panels. For example, a photovoltaic cell string may include a plurality of photovoltaic cells connected together in a series, parallel, series-parallel, or other

configuration. For simplicity, the term "PV cell" as used herein will refer to a single cell or any such combination of cells. A review of existing micro-inverter topologies reveals that for power decoupling, electrolytic capacitors, very high bus voltage, or auxiliary power circuits are used. The former has life-time, size, and maintenance problems and the latter approaches mainly exhibit low efficiency, high number of stages, and/or high voltage stress which compromise performance.

A micro-inverter topology as described herein overcomes the aforementioned problems. Figure 2 shows a block diagram of an embodiment of the power circuit 200 together with a control section 300. Referring to Figure 2, a capacitor Cpvi is connected across the power generator 10 and the input of a voltage chopper 210. The voltage chopper topology may be, for example, half bridge or full bridge. From the voltage chopper, the circuit includes a resonant tank circuit 220, an isolation transformer 230, a first converter 240, a capacitor C _bUS , a second converter 250, and an output filter 260. The converters may be voltage source converters. A half bridge or full bridge rectifier may be provided at the output of the isolation transformer. The control section 300 provides gate signals for the voltage chopper 210 and may also provide maximum power point tracking. The control section also provides decoupling and high switching frequency of the voltage chopper stage 210, to ensure that main power pulsations are drawn from the bus capacitor C _bus and input power extraction is constant, while using only a small capacitor Cpvi at the PV terminal. The decoupling method creates a DC plus AC voltage at the middle stage converter 240 and then uses a modified pulse width modulation (PWM) technique to remove double frequency harmonics from the output current. Therefore, the power pulsations have minimum impact on the input power extraction or output power injection. By accepting a large AC oscillation on the bus and controlling the average bus voltage, both the value of the bus capacitor and the voltage stress on the inverter may be optimized. For example, the input and bus capacitors may be reduced to less than 20 microFarad, an improvement of two or three orders of magnitude, relative to prior designs. As a result, the topology and control systems described herein yield a compact design suitable for micro-inverter application.

A micro-inverter as described herein employs an isolated resonant-mode topology using a high switching frequency. The high switching frequency (e.g., 10 kHz or higher) allows use of small circuit components, relative to a lower switching frequency, resulting in a high power density. In general, as the switching frequency increases, switching losses associated with turning on and off the active device(s) also increase, typically resulting in low conversion efficiency. However, in the embodiments described herein, the high switching frequency does not affect the efficiency of the circuit because a soft switching technique is employed. Soft switching refers to the switching of a device (i.e., turning the device on or off) when there is zero voltage across the device (i.e., zero voltage switching (ZVS)) or zero current through the device (i.e., zero current switching (ZCS)). That is, in a resonant-mode topology the switching losses are low, allowing operation of a resonant converter at a high switching frequency with low switching loss.

In one embodiment the isolated resonant-mode converter is a series resonant converter. This embodiment exhibits near zero switching losses while operating at constant and very high frequencies (e.g., 100 kHz or higher). The component count is low, making it feasible for micro-inverter applications.

The embodiment of Figure 2 is suitable for obtaining power from a power generator and interfacing the power with an AC load, such as a power distribution system. Figure 3(a) shows an exemplary circuit diagram of an embodiment without a converter stage 250 (see Figure 2), but with a load RL, wherein the output may be DC. This embodiment also employs a series resonant DC-DC converter. As shown in Figure 3(a), the embodiment may include a control section 300, wherein maximum power point tracking (MPPT) may be employed. The power generator 10 may be a PV cell. The circuit includes a chopper (Mi, M ₂ , shown with their body diodes DMI and DM ₂ ), a series resonant tank (C _s , L _s ), a high frequency isolation transformer (T), a rectifying circuit (D _1; D ₂ ) and an output filter (C _f ). The function of the chopper is to convert the DC input voltage (V _g ) to a high frequency AC voltage. The series resonant tank converts this AC voltage into oscillatory (i.e., substantially sinusoidal) current (i), with low frequency voltage components. This feature is useful, when, for example, the load is an inverter which is injecting the power into a power distribution grid. Low frequency ripple generated by the inverter may be substantially blocked by the resonant tank, preventing it from appearing at the input and allowing use of small energy-storage capacitors. The high frequency transformer provides matching and isolation for the output voltage. The rectifying circuit and the output filter convert the high frequency resonant current into a DC output voltage. The load RL may be a DC load, a converter stage, or a converter and/or inverter stage, as shown in Figure 2. For example, the load may be an inverter that produces AC output power suitable for injection into a power distribution grid.

As used herein, the term "low frequency" refers to a harmonic frequency that is derived from the AC grid frequency. As used herein, the term "high frequency" refers to a switching frequency of the converter and one or more harmonics thereof.

An alternative embodiment is shown in Figure 3(b). This embodiment is based on a full bridge resonant converter with a series resonant tank. The full bridge may be controlled using a modified phase shift modulation technique. Soft switching of the power MOSFET switches Mi-M ₄ may be achieved by tuning the resonant frequency of the series resonant tank circuit C ₃ -L _s lower than the operation frequency of the resonant inverter. This is an inductive mode where the impedance viewed from the input port of the resonant tank ¾„ becomes inductive. Resonant current lags the fundamental voltage and as a result Θ becomes positive. Switches Mi and M ₃ turn on during negative current, therefore, they have zero voltage turn on because current always transfers from the anti-parallel diode to the MOSFET. The scenario is the same for switches M ₂ and M ₄ . All the switches turn off under non-zero resonant current. Snubber capacitors may be used to reduce turn-off switching losses.

Principle operating waveforms of the embodiment of Figure 3(b) are shown in Figure 3(c). The output voltage of the converter (v _ab ) is regulated through the change of the phase shift a between the two pulses for Mi and M ₃ in Figure 3(c). The phase shift angle increases when the input capacitor voltage decreases, and the phase shift angle decreases when the input capacitor voltage increases. In this manner, regardless of the output load (RL) stipulated by the next stage, or environmental conditions for the PV module, the input voltage can be regulated to remove, reduce, or minimize oscillations from the input port and to substantially avoid deviation from the maximum power point.

The embodiment of Figures 2 and 3(a) will now be described in detail. The inverter output current controller forces the current to be in-phase with the grid voltage. Since the input power extraction from the power generator is DC, the power oscillation will be drawn from the bus capacitor Cb _US and the resonant tank passive elements. As a result the passive components may be small. With some simplification, bus voltage oscillation may be derived as follows: ? ¹ p i.n

This equation shows that for a given DC bus voltage and input power, if the bus voltage variation is large a smaller bus capacitor may be selected. For example, in a conventional approach the bus voltage variation may be 1 volt, whereas in the embodiments described herein, the voltage variation may be 20 volts, so the capacitor may be 20 times smaller. The bus voltage can be estimated as:

As a result the equivalent resistance of the embodiment may be calculated as:

The maximum and minimum of this equation are used in the following discussion to estimate maximum and minimum of RL.

Various techniques may be used to control the switches of the chopper. For example, a modified pulse-width-modulation (PWM) technique may be used. For example, in the embodiment of Figure 3(a), an asymmetrical pulse-width-modulation (APWM) technique is shown. A PV cell may be connected directly to the input of the resonant converter, as shown in Figure 3(a). A challenge in using PV cells is presented by their nonlinear current-voltage (I-V) characteristics, which result in a unique maximum power point (MPP) on the power- voltage (P-V) curve, as shown in Figure 4. Therefore, to achieve maximum power point tracking ability, the resonant converter can control its input voltage, so that it can operate at any point of the PV cell I-V curve, or at least at points near the maximum power point (MPP). The output voltage of the resonant converter may be regulated by a grid-connected inverter. An embodiment of an APWM series resonant converter control scheme is shown in

Figure 3(a). This embodiment includes a MPPT block 310 that receives the input current I _g and voltage V _g information from voltage and current sensor blocks H and R, respectively, a subtracter 320, a duty cycle controller 330, and an APWM modulator 340 that produces the gate signals for switches Mi and M ₂ . In one embodiment the duty cycle controller may include a proportional integral-derivative (PID) compensator. The MPPT block generates a reference voltage (V _ref ) which is compared with V _g and the result fed to the PID compensator. The PID compensator forces the input voltage to follow the voltage reference generated by the MPPT block. Operation of the series resonant converter of Figure 3(a) will now be described with reference to the plot of typical steady state operating waveforms shown in Figure 5, and the equivalent circuits shown in Figure 6. The equivalent circuits show the inherent output capacitances C _m i and C _m2 of Mi and M ₂ , respectively. Interval (to-ti): At to, the resonant current i is crossing zero. During this interval, M| is on and M ₂ is off. The output voltage VAB of the chopper is V _g . Current i _sc flows through the diode D ₂ . The voltage on the primary side of the transformer T is nV ₀ , neglecting the voltage drop of the diode. At ¾, M| is turned off for voltage regulation.

Interval (ti-t ₂ ): At the beginning of this interval, Mi is turned off. The positive current flowing through the resonant branch forces the charging of the output capacitance C _m i of main switch Mi and the discharging of the output capacitance C _m2 of M ₂ . Once C _m i has fully charged and C _m2 has fully discharged, the current forces the conduction of body diode D _M 2 of M ₂ . At t ₃ , M ₂ is turned on to achieve zero voltage switching (ZVS).

Interval (t ₂ -t ₃ ): During this interval, Mi is off and M ₂ is on. Di is off and D ₂ is still on. The output voltage VAB of the chopper is zero. The voltage on the primary side of

transformer is nV ₀ . At t ₃ , the resonant current is crossing zero.

Interval (t ₃ -t4): The resonant current changes its direction at t ₃ . It forces conduction of the diode Di. During this interval, M ₂ and Di are on, and Mi and D ₂ are off. The output voltage VAB of the chopper is zero. The voltage on the primary side of transformer also changes its direction, -nV ₀ . At t ₄ , M ₂ is turned off.

Interval (t ₄ -t ₅ ): At the beginning of this interval, M ₂ is turned off. The negative current flowing through the resonant branch forces the discharging of the output capacitance Cmi of main switch Mi and the charging of the output capacitance C _m2 of M ₂ . Once C _m i has fully discharged and C _m2 has fully charged. The current forces the conduction of body diode D _m i of Mi. At t ₅ , Mi is turned on to achieve zero voltage switching (ZVS).

Interval (t ₅ -to): During this interval, Mi is on and M ₂ is off. D ₂ is off and Di is on. The output voltage VAB of the chopper is V _g . The voltage on the primary side of transformer is -nV ₀ . At to, the resonant current i is crossing zero. The current through the diode Ό _\ goes to zero. Another operating cycle begins. Steady-state analysis

The AC equivalent resistance is given

The voltage VAB on the input of the resonant tank: also can be given using Fourier series:

Where ω ₀ is the switching frequency and D is the duty cycle. The following qualities are defined as:

ft)

ω

The kth harmonic equivalent circuit is shown in Figure 7. The resonant current ir may be expressed as:

i=l kn Z ink (7) o achieve ZVS for Mi, the current at the turn off of M ₂ must be negative, as given by (8):

o achieve ZVS for M ₂ , the current at the turn off of Mi must be positive, as given by (9):

As an example, Figures 8(a) and (b) illustrate zero voltage switching of both switches, obtained from a simulation using PSIM™ software version 9 (Powersim Inc., Woburn, MA).

Embodiments have been described herein using a voltage input. The embodiments can also be configured to work with a current input.

Modified Pulse Width Modulation

There may be input power oscillation from the power generator. In such case, maximum power is only achievable at the peak of oscillation, which translates into less average power extraction than the available maximum power. This is a power loss that reduces the efficiency of PV systems, and substantially the same problem exists in systems with wind turbines or fuel cells as the power generators, for single phase systems or unbalanced three phase systems. Therefore, power oscillation is a key problem in such systems and the converter should decouple the output power oscillation from the input DC power generation to maximize efficiency. Power decoupling is conventionally performed passively by using large electrolytic capacitors to minimize the effect of the output power oscillation on the input operating point. However, use of large capacitors increases the size and weight of the converter, and lowers reliability, resulting in high maintenance costs. The decoupling control method used in the converter described herein removes double frequency oscillations from the input power generator source. As shown in Figure 9, this oscillation is displaced to the middle bus voltage (V _bUS ) that oscillates around a DC value at twice the grid frequency. A bus voltage controller regulates the DC value of the bus voltage so that the lowest value of the bus voltage is higher than the peak of the grid voltage, otherwise it results in a discontinuous mode of operation where the output current becomes distorted.

As noted above, a modified pulse width modulation (PWM) technique is used to generate the output current in the presence of bus voltage ripples. If a conventional PWM technique was used in the presence of ripples, this double frequency harmonic would be multiplied by the fundamental harmonic of the carrier, creating first and third harmonics at the output current. Such a detrimental low frequency harmonic should be avoided.

This problem is avoided by the controller described herein. In the modified PWM technique, an active compensation factor is generated as shown in Figure 9. When the oscillatory input DC voltage increases, the compensation decreases the modulation index proportionally. As a result, an increase in the DC current value is compensated by a reduction in the modulation pulse width, and vice versa. Therefore, the inverter output voltage average over one switching cycle will be equal to the case where the bus was a constant voltage. This compensation prevents the oscillatory harmonics from appearing at the output current because the energy transfer to the output will be equivalent to the case where the bus voltage was a constant DC with no oscillation. Figure 10 shows two cases; at the beginning the compensation factor is not applied and the output current harmonics appear. In the second half of the waveform the modified PWM is applied and the output current has been fully compensated and contains no low order harmonic component.

The following non-limiting example is provided to further illustrate the invention. Working Example

A simulation of a converter based on the schematic of Figure 3(a) was carried out using PSIM software version 9 (Powersim Inc., Woburn, MA). Figures 11 and 12 show experimental results. Figure 11 shows zero voltage switching of the converter switches. Figure 12 shows that the bus voltage and the resonant tank current and voltage all have low frequency oscillations, without affecting the input operating point. Figure 12 also shows that the oscillatory power is being supplied from all of these passive components and this is the reason that the components are optimized.

The contents of all references, pending patent applications, and published patents cited throughout this application are hereby expressly incorporated by reference.

Equivalents

Those skilled in the art will recognize or be able to ascertain variants of the embodiments described herein. Such variants are within the scope of the invention and are covered by the appended claims.

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