Background Nearly all commercially available power amplifiers for fre- quencies in the above range are of the linear analog type: class A, AB, and B. Since the output transistors of such amplifiers operate in the linear region, they have a low efficiency and dissipate a considerable amount of heat. The basic pulse modulation ("digital") switching class D power amplifier method in theory provides a much higher effi- ciency, which lowers the amplifier volume and heat develop- ment. Despite this efficiency advantage, prior art in the field has not provided solutions with an acceptable audio quality, that could make them generally useful and direct replacements for analog amplifiers. Accordingly, the use of digital power amplification has been limited to applica- tions where the demands for output quality are low. The reasons for this will be presented in the following, when the general prior art principles are discussed.

Prior Art There is a fundamental requirement for efficient control systems to eliminate errors generated in the various blocks of the digital switching power amplifier. In the basic digital power amplifier the input signal is modulated into a pulse modulated signal. A switching power stage performs amplification of the signal, and a low pass filter regen- erates the modulated but now amplified signal. The basic method form the basis for a range of prior art arrange-

ments. However, there are numerous non-ideal features with this method - Any modulator errors are fed directly to the load.

- Any power stage errors are fed directly to the load.

- There is no rejection of power supply perturbations.

Since the power stage output is largely proportional to the supply voltage, any supply ripple will intermodulate with the audio signal.

- The post filter errors will introduce further distortion, since magnetic core materials are not ideal.

- The total output impedance is high, especially at high frequencies, due to the filter.

- The sensitivity to load variations is high due to the passive post filter. Accordingly, changes in load impedance will distort the frequency response of the amplifier.

- The sensitivity to temperature drift, component toler- ances, and aging effects is high. The non-controlled digi- tal switching amplifier is therefore not robust and reli- able.

Compensation for these major problems areas is absolutely vital, if very high fidelity is to be obtained. Prior art methods are based on two different control methods, charac- terized by having a single feedback loop.

One basic principle is to feed back the amplifier output.

However, the poles of the low pass filter cause a large phase shift which puts strong restrictions on loop design.

Consequently, reasonable loop bandwidths requires high switching frequencies. This causes several problems, such as lower efficiency, and poor performance due to the first two errors mentioned above.

Another basic principle of feedback in prior art, is single loop feedback before the filter network, whereby the phase lag of the post filter is avoided. However, the high fre- quency content at the power stage output makes the feedback

source potentially very noisy. Furthermore, a number of the above errors are not (or only partially) compensated The fundamental problem of both methods are the conflicting desires for a low carrier frequency, high gain-bandwidth product and good stability characteristics in all situa- tions. A further problem with prior art arrangements is the apparent complexity of the system, partially caused by in- efficient control.

The following publications are relevant as background mate- rial and illustration of the methods and the problematic issues of three prior art arrangements: [1] Suzuki, T. Pulse Width Modulated Signal Amplifier. US Patent no. 4021745 (1977) [2] Yokoyama, K. Pulse-Width Modulation Circuit. US Pat- ent no. 4531096 (1986) [3] Attwood, B.E. Design Parameters Important for the Optimization of Very High-Fidelity PWM (Class D) Audio Am- plifiers , Journal of the AES, Nov. 1983. p. 842-853.

[4] Taylor, W.E. Digital Audio Amplifier . US Patent.

No. 4724396 (1988).

[5] Hancook, J. A class D Amplifier Using MosFET's with Reduced MinorityCarrier Lifetime , 89th Convention of the AES. Los Angeles. CA. September 21-25. 1991.

[6] Solomon, E.E : Digital Power Amplifier. US Patent No.

5126684 (1992).

[7] McCorkle, D.P. Class D amplifier. European Patent.

Publ. No. 557032A2 (1993).

[8] Nakajima, Y. Pulse-Width Modulation amplifier. European Patent. Publ. No.503571A1 (1993).

[9] Leigh, S.P et al. Distortion analysis and reduction in a completely digital PWM class D power amplifier, Interna- tional Journal of Modeling & Simulation, Vol. 14, No. 2, 1994.

OBJECTIVES According to the above stated problems that exist with prior art arrangements, the primary objective of the pres- ent invention is to provide a pulse modulation amplifier which can deliver very high power outputs, and still pro- vide ultra low distortion (less than 0.01%) and noise (less than 100WV RMS), and yet a very high efficiency (90-95%) and low idle losses.

Another objective of the invention is to maintain low com- plexity by avoiding the use of advanced but complex and hence potentially unreliable circuitry, and also to elimi- nate the requirement for tuning in production.

Another important objective of the invention is to elimi- nate the need for a stabilized supply, meaning that a sim- ple non-regulated bridge rectifier with a stabilizing ca- pacitor is sufficient. In this way, a minimal complexity and maximal efficiency is secured from the mains input to the amplifier output terminals.

The final objective of the invention is to obtain minimal sensitivity to load variations, and furthermore to provide robustness and reliability.

SUMMARY OF THE INVENTION The above objectives are obtained with the present inven- tion. In the first preferred embodiment, an amplifier ac- cording to the invention is particular in that negative feedback is introduced from the switching power stage out- put to one or several loops feeding into one or several pre-amplifier stages preceding the modulator. This offers a range of advantages that are new to the art, in terms of performance and stability control. A further embodiment of the invention is particular in that the local feedback has a enhanced cascaded structure with a single feedback path, that comprises a filter with a phase characteristic such that a pole in the demodulation filter is compensated.

Another embodiment of the invention is particular in that further feedback is established from the output of the de- modulation filter to one or several pre-amplifier stages, so that the pulse modulation ("digital") switching power amplifier circuit elements are enclosed by an enhanced cas- cade structure of feedback loops, providing further im- proved performance and stability control.

A further embodiment of the invention is particular in that the pulse modulator is a controlled self-oscillating modu- lator comprising a non-hysteresis comparator for pulse modulation, and a higher order oscillating loop realized by means of two poles, preferably a pole in the first (local) forward path and feedback path. This provides en extremely simple and stable configuration, obviating the need for a separate carrier frequency generator.

A further embodiment of the invention is particular in that the pulse modulator is a carrier based modulator. This means that well-known design techniques for carrier based modulators may be used in the configuration according to the invention.

A further advantageous embodiment of the invention when carrier based modulation is used, is particular in that a notch filter is provided in the single feedback path block between the amplifier and the loops. Thereby it is possible to simplify the filtering out of the carrier frequency in- fluences on the system performance in carrier based sys- tems. A similar result is obtained in a further advanta- geous embodiment of the invention which is particular in that a structure creating a high frequency pole is provided in the feedback path block.

BRIEF DESCRIPTION OF THE DRAWINGS The invention will be further described with reference to the drawings, in which,

Fig. 1 shows the principle in prior art conventional pulse modulation ("digital") switching power amplifiers Fig. 2 shows a prior art method with single loop feedback based on the overall amplifier output as the feedback source Fig. 3 shows a prior art single loop feedback with power stage output as the feedback source Fig. 4 shows the a general block diagram of a first embodi- ment of the present invention, a digital switching ampli- fier improved by an Multivariable Enhanced Cascade Control- ler (MECC) based on single feedback from the switching power stage output.

Fig. 5 shows a general block diagram of a second embodiment of the present invention, Fig. 6a and 6b exemplify general recursive loop synthesis methods for the first (Fig. 6a) and second (Fig. 6b) em- bodiments of the invention.

Fig. 7 shows a detail of a preferred embodiment of single feedback MECC.

Fig. 8 shows a detail of a preferred example of the second embodiment of the invention, the dual feedback Multivari- able Enhanced Cascade Controller (MECC).

Fig. 9a and 9b show the loop characteristics for the two preferred examples of the first two embodiments of the pre- sent invention, shown in Fig. 7 and Fig. 8.

Fig. 10 illustrates the third embodiment of the present in- vention, the controlled self-oscillating pulse modulator characterized by a non-hysteresis comparator and additional

poles in the first local loop to secure self-oscillating conditions Fig. 11 shows essential signals of the third embodiment of the invention, the controlled oscillating pulse modulator.

Fig. 12 illustrates the principle of a fourth embodiment of the invention, an alternative three-level pulse width modu- lator for implementation with a 4 transistor bridge power stage. The signals are from top to bottom : Modulating sig- nals, normalized voltage on each phase in a bridge power stage (A,B), and finally both the normalized differential and common mode output signals (A-B and A+B). The differ- ence signal is fed to the load.

Fig. 13 illustrates the spectral characteristics of the al- ternative three-level modulator. The output amplitude spec- trum is shown OdB and -60dB relative output levels.

Fig. 14 (a)-(c) illustrate various performance specifica- tions for an implemented example embodiment of the inven- tion with a 250W maximal power handling capability. Fig. 13 (a) shows power efficiency. At 250kHz switching frequency the efficiency approaches 92%, and with 50kHz switching frequency the efficiency approaches 96%.

Fig. 14 (b) shows the very low measured Total harmonic dis- tortion + Noise (THD+N) vs. output power at 100Hz, 1KHz and 10KHz (top curve).

Fig. 14c (c) shows the measured amplifier noise. The noise floor at -150dB corresponds to an RMS noise level of only 70we.

DETAILED DESCRIPTION The basic "digital" power amplifier is shown in Fig. 1. The input signal is modulated (11) into a pulse modulated sig- nal. A switching power stage (12) performs amplification of

the modulated signal, and a low pass filter (13) regener- ates the audio waveform.

A first category of prior art control principle is shown in Fig. 2. This displays the basic problems: that a reasonable gain- bandwidth product requires very high switching fre- quencies due to post filter phase lag and the lack of local correction schemes. Reasonable gain-bandwidth product at all frequencies within the amplifier bandwidth is impossi- ble to obtain.

Fig. 3 shows a second category of prior art control princi- ples which displays the following basic problems: noisy feedback source, no post filter error correction, load sen- sitivity and limited loop gain-bandwidth product caused by single loop control.

Fig. 4 illustrates the first embodiment of the invention.

The basic pulse modulation ("digital") switching power am- plifier circuit elements are enclosed by an enhanced cas- cade (or nested) structure of feedback loops. This first embodiment is termed a (single feedback) Multivariable En- hanced Cascade Controlled (MECC) digital power amplifier.

The control structure is characterized first by having a single feedback source and second by a single feedback path A (7) having a lowpass characteristic. The control struc- ture is simple in implementation since it consists of a single feedback path A and a set of forward path blocks Bi.

Using preferred loop design procedures, each loop has a very stable 1st order characteristic and the forward path blocks are preferably simple. MECC offers a range of advan- tages that are new to the art: - A higher order control system combined with a high level of stability and robustness, since each loop considered in- dividually is very stable.

- The enhanced cascade control method provides hitherto un-

known freedom in loop design and optimization possibili- ties. Thus, the equivalent loop gain bandwidth at any fre- quency can as such be increased infinitely relative to sin- gle loop systems without compromising stability.

- The lowpass characteristic of the feedback path provides a closed loop zero-pole lead characteristic that can be de- signed to cancel one of the filter poles. The result is a much improved system for global feedback.

- Each loop individually reduces the sensitivity to power stage errors and improves system performance (distortion, noise...) by a factor corresponding to the loop gain. MECC provides an equivalent feedback corresponding to the prod- uct of contributions of each loop.

- Successive improvement by multiple loops is more effi- cient than in a one loop realization.

- Frequencies of unity loop gain in each loop can be re- duced compared to single loop system, while still providing improved performance. Thus, the switching frequency can be reduced.

- MECC has low requirements for dynamic range of the indi- vidual compensator blocks A and Bi. Using preferred design procedures, the signal levels throughout the control system will have a level similar to the input signal.

The second embodiment of the invention involves the exten- sion of the first embodiment to a dual feedback Multivari- able Enhanced Cascade Control (MECC)structure, were the two cascades are closely connected. A general block diagram is illustrated in Fig. 5. The system relies on the zero-pole lead characteristic caused by the single lowpass feedback path of the local enhanced cascade. Dual feedback MECC pro- vides further improvements to the system. The system band- width is increased beyond the demodulation filter bandwidth

limit. This minimizes phase and amplitude distortion within the audio band. Furthermore, the transient response is im- proved. Post filter errors are corrected, meaning that the requirements for filter inductor linearity is low. Further- more, the output impedance is reduced considerably, and so is the sensitivity to load variations.

A third embodiment of the invention is a forced self- oscillating pulse width modulator, characterized by first a non-hysteresis comparator as a modulator and secondly by a higher order oscillating loop realized with both forward path B1 and feedback path A to determine self-oscillating conditions.

A fourth embodiment of the invention is an alternative car- rier based three-level pulse width modulator with attrac- tive characteristics in combination with MECC.

Further embodiments of the invention include - Means for obtaining improved elimination of noise from the noisy feedback by using a notch filter and high fre- quency poles in the local feedback or alternatively forward path. This improves distortion when carrier based pulse width modulation is used.

- Means for compensating for large scale power supply regu- lation, in order to obtain improved stability and effi- ciency improvements at all output levels. This is relevant with carrier based modulation, where the gain of modulator and power stage is dependent on the power supply rail level, meaning that the power supply perturbation may in- fluence stability unless such precautions are taken.

DESCRIPTION OF THE FIRST EMBODIMENT The enhanced cascade control method new to the art relies on the fact that the modulator and amplifier can be repre- sented by a constant gain Kp over a wide bandwidth. One

general recursive approach to design the local cascade is illustrated in Fig. 6a. The feedback path has a lowpass characteristic: 1 1 K rl S +1 (1) The lowpass characteristic of the feedback path is benefi- cial in several aspects. It causes a closed loop zero-pole phase lead characteristic which can be very useful for re- alizing a cancellation of one of the poles of the demodula- tion filter. Furthermore, the pole causes an important fil- tering of the high frequency switching noise from the power stage output, which is essential when carrier based modula- tion methods are used (the fourth embodiment of the inven- tion). In this simple examplary embodiment the initial for- ward block is a simple gain, with a gain that leads to a constant open loop and closed loop gain of K in the fre- quency band of interest: Whereas the i'th block has a pole-zero characteristic to compensate for the zero-pole characteristic of the preced- ing loop: By this realization the open loop gain of the single loop configuration can be written by the following 1. order ex- pression:

HL1,1 (s) = Kp A(s)B(s) rl <BR> <BR> <BR> + 1 r2 It is easy to show that for all loops: The closed-loop transfer function can be approximated to: By this specified simple loop synthesis procedure, each loop will exhibit identical and stable 1st order behaviour as illustrated in Fig. 6a.

Numerous alternative approaches to loop synthesis can be devised. The recursive design procedure specified above provides a frequency independent loop gain at all frequen- cies within the target bandwidth, irrespective of the num- ber of cascaded loops and the loop unity gain frequency.

Instead of a constant loop gain, each loop can realize e.g.

an integrator. Alternatively, each loop can be designed us- ing two pole compensation. Both methods lead to a consid- erably higher but frequency dependent loop gain within the frequency band of interest.

Alternative variants of single feedback MECC power ampli- fier include the use of alternative different feedback paths A, which are optimized in respect to noise attenua- tion which is essential in systems where carrier based modulation is used. If HF components related to the switch- ing frequency were to reach the modulator, it would be im-

possible to obtain a pulse width according to the instanta- neous amplitude of the audio signal. By introduction of a notch filter, the first critical component and intermodula- tion components can be eliminated. The specific placement of the notch filter in the feedback path effectively elimi- nates the switching fundamental in all loops. By further addition of a high frequency pole in block A, a 2nd order attenuation characteristic of the higher order harmonics of the switching frequency in the feedback path is realized, and the noise attenuation is thereby further improved. A further advantage of placing the noise attenuating cir- cuitry in the feedback path is, that it will only have a local effect for each loop, if the unity loop gain fre- quency and switching frequency are properly chosen. The noise elimination circuitry operates well above unity loop gain, and does therefore not influence the closed loop be- havior of any of the loops.

DESCRIPTION OF THE SECOND PREFERRED EMBODIMENT The second embodiment of the invention involves the exten- sion of the first embodiment to a dual feedback Multivari- able Enhanced Cascade Control (MECC)structure, were the two cascades are closely connected. A general block diagram is illustrated in Fig. 5. The system relies on the zero-pole lead characteristic caused by the single lowpass feedback path of the local enhanced cascade. Due to the use of two feedback sources, this embodiment is termed Dual feedback Multivariable Enhanced Cascade Control. The second cascade has the same special characteristics as the first cascade with only one single feedback path C and a set of forward path blocks Di. One preferred approach of dual feedback MECC design is based on the local enhanced cascade in Fig.

6a and illustrated in Fig. 6b. The main reconstruction fil- ter F(s) is assumed to be 2. order. The feedback path has a constant gain characteristic:

C(s)= K (7) The initial block forward block D2 is a simple gain block with a gain in the frequency band of interest <BR> <BR> <BR> <BR> <BR> <BR> <BR> (8) <BR> <BR> <BR> 23 (8) <BR> #3 Whereas the i'th forward block has a pole-zero characteris- tic By this simple recursive design procedure, each loop will exhibit identical behavior, which is illustrated by both the open loop HGi,oL(s) and closed loop characteristics HGicL(s) in Fig. 6b: The closed-loop transfer function can be approximated to <BR> <BR> <BR> <BR> <BR> <BR> K<BR> HG <BR> (r3s+t)(r2s+1) (11) It should be emphasized, that the given design approach is mainly illustrative, and that there are numerous possible extensions which will be apparent to the skilled person, e.g. a first order characteristic in each loop to improve correction for errors at lower frequencies.

To maximize the robustness of the dual feedback MECC Digi- tal power amplifier, the number of global loops should be minimized and preferably only one global loop should be

used. Tuning towards the desired distortion and noise char- acteristics should be carried out by adjusting the number of local loops, since the dominating errors are introduced in the switching power stage block, and should therefore be corrected locally.

A fundamental advantage of the powerful Multivariable En- hanced Cascade Control method in both embodiments is, that design of modulator, power stage, post filter and power supply can be relaxed considerably without compromising audio performance. These fundamental elements can be imple- mented with simple standard components and thereby have a low cost and complexity.

The control method comprising first and embodiment of this invention functions independently of the pulse modulator and power stage realization. The only requirement for the- ses two blocks is that they realize an amplification of the analog modulator reference input over a frequency range that is wider that the desired bandwidth. Thus, the modula- tion may be single sided or double sided, two level or mul- tilevel pulse width modulation, or even alternative modula- tion schemes such as Sigma-Delta modulation.

THE THIRD EMBODIMENT OF THE INVENTION A further preferred embodiment regarding modulator imple- mentation for the MECC digital power amplifier is a Con- trolled self-Oscillating Pulse Modulator, new to the art.

An example of an embodiment realizing this method is shown in Fig. 10. The preferred method is characterized by having a non-hysteresis comparator as a modulator and by modifying the first local loop to a have higher order characteristic, by an additional pole in both forward path block B1 and feedback path A. This secures controlled and stable self- oscillating conditions. The desired pulse modulation effect is then obtained by superposing the oscillating signal with the signal input (Vi). Fig. 11 shows an example of the sig-

nal characteristics at the reference point for the modula- tor where the oscillating signal is superposed with the in- put signal. Furthermore, the pulse modulating effect is shown.

Advantages of the above described Controlled self- Oscillating Pulse Modulator over constant frequency carrier based methods are several. First, the modulator is ex- tremely simple to implement since no carrier generator is needed. Secondly, the bandwidth of the unity gain frequency of the first local loop is the oscillation frequency lead- ing to wide bandwidth control even with modest switching frequencies. Third, power supply rail no longer determines the equivalent gain of the modulator/power stage meaning that large scale power supply perturbation is automatically cancelled, and does no longer influence stability in other loops in the enhanced cascade structure. The unity gain frequency of the local loop is inherently determined by the frequency of positive feedback, i.e. the control loop band- width is considerably wider than in traditional carrier based systems, where a rule of thumb is a factor of three between unity gain frequency and carrier frequency.

OTHER EMBODIMENTS Other embodiments of the of invention include the use of pulse width modulation, preferably with three discrete am- plitude levels. To aid in this understanding this alterna- tive embodiment regarding modulator implementation, Fig. 12 illustrates the essential time domain waveforms and Fig. 13 the frequency domain spectral amplitude characteristics.

From Fig. 12 it is obvious that by using three-level PWM the effective sampling frequency is doubled since there are two samples pr. switch cycle. This is obtained without in- creasing the power losses, since each transistor operates at a rate equal to the switching frequency. Accordingly, three-level PWM allows the loop bandwidths to be increased or alternatively the switching frequency to be decreased.

A further embodiment of the invention relates to the case where MECC is combined with constant frequency carrier based modulation, and is characterized by the design change described hereinafter. Improved means to compensate for any large scale variations of the power supply voltage is ob- tained by intelligent automatic gain adjustments in the first local loop. The equivalent gain of the modulator and power stage, Kp, is largely proportional to the power sup- ply voltage. Accordingly, the gain of B1 should be regu- lated inversely proportional to the supply voltage varia- tion: <BR> <BR> <BR> <BR> <BR> <BR> <BR> K K r1 <BR> <BR> Kp (Kp 22 = K t (12) This design change will have a further stabilizing effect on the digital power amplifier based on MECC since the characteristics of the first loop will be independent power supply perturbations. Large perturbations of the power sup- ply voltage will not influence stability. The adaptive ad- justment of the power supply allows an intelligent control of the power rail voltage e.g. controlled by the volume control without compromising stability. This can be util- ized to secure optimal efficiency at all output levels.

Further embodiments include additional secondary filtering circuitry on the amplifier output for further elimination of switching components, and the addition of an input fil- ter to shape the overall amplifier response in frequency and time.

EXAMPLES OF THE FIRST AND SECOND EMBODIMENT Fig. 7a and 7b show more specific double loop examples of both the single and dual feedback MECC embodiments. The de- sign procedure of each loop is given in Fig. 9a and 9b re- spectively. Compared with the general design procedure, the

LF performance has been optimized such that the equivalent loop gain in both loops is significantly increased at lower frequencies compared to the general approach. Table 1 gives example parameter values relative to the amplifier band- width.

Table 1 Parameter Frequency (rel. to bandwidth) 2 l/r2 10 l/rLF 1/5 The enhanced cascade provides a significant improvement over a the single loop approaches known in the art, with only a marginal increase in system complexity in terms of the added forward block B2. Provided that the reconstruc- tion filter is reasonably linear, the local feedback MECC embodiment can realize high end specifications. Further im- provement may be realized by adding blocks equal to B2. Al- ternative use of this first embodiment of the invention is to drive loudspeakers directly without reconstruction fil- ter.

Fig. 7b provides an example of an embodiment of dual feed- back Multivariable Enhanced Cascade Control (MECC) with one local and one global loop, both closely connected. Compared to the general design procedure specified in Fig. 6 both loops have been LF optimized to significantly increase the loop gain in both loops at lower frequencies. Table 2 gives examples of parameter values relative to the amplifier bandwidth.

Table 2 Parameter Frequency (rel. to bandwidth) 1/#1 2 1/#2 10 1/viz 4 1ITLF 1/5 X In order to clarify the significant advantages of the dual feedback MECC digital power amplifier over prior art, the embodiment in Fig. 8 has been implemented in two higher power examples, one to cover the full audio bandwidth of 20kHz utilizing a 250kHz switching frequency, and one exam- ple covering a reduced bandwidth of 4kHz, utilizing a 50kHz switching frequency.

It should be emphasized that the chosen parameters are only illustrative, and that the MECC digital power amplifier is well performing on a much wider range of output powers and bandwidths. Figs. 14 (a) - (c) illustrate various key specifications for the given example, and the results ob- tained are summarized in Table 3. If further improvements are desired, one preferred approach is to use a double lo- cal loop by simple addition of a B forward block.

Table 3 Specification Measurement Max. power 250W Bandwidth 20kHz / 4kHz THD+N (1KHz, 1W) <0.01% THD+N (20Hz-20KHz) <0.05 % Intermodulation dis- <0.01% tortion (IMD) Idle noise (RMS) 70>V within bandwidth Dynamic range 115dB Power Stage Efficiency 92% / 96% (250kHz / 50 kHz In general no prior art approach yields a comparable combi- nation of fidelity, efficiency, and low complexity.