* * *

Field of the invention

This disclosure relates to overload protection of resonant converters.

The disclosure was developed with attention paid to its possible use in connection with resonant converters as used to power supply e.g. high flux LED lighting sources.

Description of the related art

Overload conditions in resonant converters are usually detected by means of a current sense coupled to the output stage of the power converter. The signal produced thereby is compared with a reference to produce an overload signal which leads to power transfer being discontinued in the presence of an overload condition.

Object and summary of the invention

An intrinsic drawback of conventional arrangements as discussed in the foregoing lies in that any failure in the overload detection loop makes it impossible to effectively discontinue power transfer in the presence of an overload condition. The inventors have noted that this drawback is particularly felt in providing overload protection as requested by UL {Underwriters Laboratories, a trusted source worldwide for product compliance) standards in Class 2 power units for the NAFTA market. Exemplary of such a standard is the reference norm UL1310 paragraph 39.6.

Specifically, if any kind of short or open circuit condition arises at one of the components, Class 2 operation has to be guaranteed for the power supply, by- ensuring that the output voltage and energy is limited even in the presence of a fault condition in any overload protection circuitry already existing. This means that an independent protection is needed. The object of the invention is thus to satisfactorily deal with these drawbacks and needs.

According to the present invention, such an object is achieved by means of a circuit having the features set forth in the claims that follow. The invention also relates to a corresponding method.

The claims are an integral part of the disclosure of the invention provided herein.

In an embodiment of the arrangement described herein, the overload condition is recognized unambiguously on the primary side of the converter independently of any overload protection circuit already implemented in the power converter.

Compliance with standards such as UL 1310 is thus ensured with a simple and cost-effective solution.

Brief description of the annexed drawings

The invention will now be described, by way of example only, with reference to the enclosed figures of drawing, wherein:

Figure 1 is a exemplary block diagram representative of a power converter including a protection circuit as described herein, - Figure 2 is a diagram helpful in understanding operation of the arrangement described herein, and

Figure 3 is an exemplary detailed circuit diagram of one of the blocks illustrated in figure 1.

Detailed description of preferred embodiments

In the following description, numerous specific details are given to provide a thorough understanding of embodiments. The embodiments can be practiced without one or more of the specific details, or with other methods, components, materials, etc. In other instances, well-known structures, materials, or operations are not shown or described in detail to avoid obscuring aspects of the embodiments.

Reference throughout this specification to "one embodiment" or "an embodiment" means that a particular feature, structure, or characteristic described in connection with the embodiment is included in at least one embodiment. Thus, the appearances of the phrases "in one embodiment" or "in an embodiment" in various places throughout this specification are not necessarily all referring to the same embodiment. Furthermore, the particular features, structures, or characteristics may be combined in any suitable manner in one or more embodiments.

The headings provided herein are for convenience only and do not interpret the scope or meaning of the embodiments . The block diagram of figure 1 is exemplary of a resonant power converter of the half-bridge, anti- parallel diode type.

Such a resonant converter can be used to provide electric power to a load L, isolated from the high voltage DC bus (400V) . In an embodiment, the output voltage of the converter is 10 or 24 volts, which may be used for supplying power to a load constituted e.g. by a high flux LED lighting source.

The converter of figure 1 includes a driver (e.g. IC 6598) providing two drive signals DRV hs (high side) and DRV Is (low side) - with a frequency of e.g. 100kHz - to alternatively switch on and off two power mosfets Ql, Q2 interposed between a high voltage (e.g. mains voltage) line HV bus with an amplitude equal to E volts and ground (0 volts) .

The two power mosfets Ql, Q2 thus represent first and second drive switches cascaded between a feed voltage (e.g. HV bus) and ground with the primary winding of the insulation transformer T connected therebetween. The driver D is configured for providing first and second drive signals (DRV hs; DRV Is) for the drive switches Ql, Q2 to produce a positive alternate drive signal to apply to the primary winding of the insulation transformer T. At the intermediate point between the power mosfets Ql, Q2 a square wave will thus be present with a frequency dictated by the switching frequency of the mosfets Ql, Q2 and an amplitude between E and 0 volts. This square wave is used to drive the primary winding of an insulation transformer T, and the voltage across the primary side of the insulation transformer T will be a square wave with amplitude between +E/2 and -E/2.

The secondary winding of the transformer T feeds a rectifier R that supplies power to the load L via a low-pass (e.g. LC) filter designated LP as a whole.

Cl, Dl and C2, D2 denote the parallel connection of a capacitor (Cl, C2) and a diode (Dl, D2 ) . Each such parallel connection is in turn connected in series with the primary winding of the transformer T. The capacitors Cl, C2 form a resonant circuit with the magnetizing inductance of the primary side of the transformer T. Specifically, each of the first capacitor Cl and the second capacitor C2 is coupled to the primary winding of the transformer to form with the magnetizing inductance thereof a first and a second branch of the resonant circuit.

The switches Ql, Q2 as driven by the driver D apply to the primary winding of the transformer T a positive alternate drive signal, whereby the first branch of the resonant circuit is traversed by a current to produce energy transfer to the secondary winding of the transformer T.

Depending on the frequency of the square wave generated by the power mosfets Ql, Q2, the converter will exhibit a gain value G varying as a function of frequency according to a resonance "bell" as illustrated in figure 2. For instance, on the right- hand side of the resonance bell reducing the frequency will result in a higher gain.

By changing the frequency of the square wave generated by the power mosfets Ql, Q2 it is thus possible to manage the effective impedance of the LCC stage (i.e. the resonant circuit including the magnetizing inductance L of the transformer T and the two resonant capacitors Cl and C2) and control the transfer of energy from the primary to the secondary side of the transformer and thus control the output power generated therefrom via rectification (in the rectifier R) and filtering (by the low-pas filter LP) .

The arrangement described is thus capable of managing increases in the power needed by correspondingly increasing (by causing the driver D to vary the frequency of the square wave generated by the power mosfets Ql, Q2 ) the amplitude of the resonant oscillations in the intermediate point between the resonant capacitors Cl and C2.

Unless otherwise indicated in the following

(primarily in . connection with the protection circuit 10) , the general arrangement illustrated in figure 1 is conventional in the art, thus making it unnecessary to provide a more detailed description herein.

This also applies to the possible presence of an overload protection circuit adapted to detect overload (e.g. by means of a current sense coupled to the output stage of the power converter) and to turn off the converter in the presence of an overload condition. Such a possible, already existing overload protection circuit may be of any known type and is not explicitly shown in the drawing for the sake of clarity. The instant disclosure in fact focuses on an improved overload protection circuit which acts on the primary side of the transformer T and will possibly ensure protection of the converter even in case of a failure in such an already existing overload protection circuit .

The inventors have noted that during operation at the rated working conditions (i.e. during regular operation of the converter as illustrated herein) the low side branch of the resonant converter (i.e. D2, C2 ) will be practically inoperative, i.e. the current flowing through it will be zero. Conversely, if an overload condition sets in, i.e. the power needed at the output of the converter becomes very high, the voltage at the intermediate point between the resonant capacitors Cl and C2 reaches the OV level and then falls also below this voltage. In this way, the diode D2 (whose cathode is connected to the intermediate point between the resonant capacitors Cl and C2) becomes conductive and keeps the cathode voltage at the forward voltage of the diode under the OV (e.g. 0.7V) . Current will then flow through D2 until no energy is finally stored into the capacitors Cl and C2. By monitoring the forward voltage of the diode D2 it is thus possible to recognize unambiguously the presence of an overload condition. This result is achieved by acting on the primary side of the transformer T and independently of any other overload protection circuit (e.g. a current sense coupled to the output stage of the power converter) possibly already implemented in the circuit.

Any such signal thus derived from the diode D2 can thus be used in order to at least partly switch off the circuit that provides the output power, or otherwise reducing or blocking the flow of power from the input to the output side of the converter.

The diagram of figure 3 is exemplary of a circuit 10 for performing such an overload protection action.

The circuit 10 includes a first input 12 for connection to the driver D. The input 12 is related via an input resistor Rl to the anodes of two diodes D3. The cathodes of the two diodes D3 are in turn related to respective parallel connections of resistors R2, R3 and capacitors ClO, C20 to derive constant bias voltages from the DRV Is output of the driver D of the resonant stage.

In the embodiment illustrated, the time constant of the first RC branch R1-C20 is larger than the time constant of the branch Rl-ClO, so that the voltage on the capacitor C20 will be slower to increase than the voltage on the capacitor ClO.

The circuit 10 includes a second input 14 for connection to the intermediate point of the diodes Dl and D2. The input 14 is related to the base of a bipolar transistor (BJT) Q4 which is also connected to the resistor R2. The collector of the BJT Q3 is in turn connected to the resistor R3 as well as to the anode of a diode D4.

The cathode of the diode D4 drives the base of a further bipolar transistor (BJT) Q3 via a RC low pass network R5 - C3. The collector of the transistor Q4 drives the gate of a mosfet Q5 whose drain-source line is connected between a fixed voltage (e.g. 15 V) and ground to load a capacitor C4 interposed between the source of the mosfet Q5 and ground.

The voltage across the capacitor C4 is fed to an output line 16 of the circuit 10 to drive the Vin input of the driver D with the capability of turning off the driver D (and the converter as a whole) in the presence of overload conditions as better detailed in the following.

The mosfet Q5 thus represents an output switch of the overload protection circuit which can be switched between an active condition (e.g. closed) to feed the driver D with an activation signal and an overload protection condition (e.g. open) to remove the activation signal from the driver D and thereby turn off the converter. R4 and C30 denote a bias resistor and capacitor for the mosfet Q5. The transistors Q3 and Q4 represent intermediate switches cascaded between the input 14 and the output switch (i.e. the power mosfet Q5) of the protection circuit 10. coupled to the input of the intermediate switch Q4 is a lowpass filter stage including a capacitor C3 and a resistor R5 to filter out pulsed signals applied thereto.

The first and second intermediate switches Q3 and Q4 are coupled to the first and second bias circuits Rl, R2, ClO and Rl, R3, C20 having first and second time constants, respectively, with the second time constant larger than first time constant .

The BJT Q4 will thus be prevented from switching on (thus turning off the driver D - and the converter) before Q3 is turned on as a result of the second branch of the resonant circuit (i.e. D2 ) being traversed by a current .

As indicated, during regular operation of the converter the low side branch of the resonant converter will be practically inoperative, with D2 behaving essentially as an open circuit. The (npn) transistor Q3 will thus be polarized (saturated) by the voltage across ClO with the anode of the diode D4 short- circuited to ground, so that transistor Q4 will be open (i.e. cut-off) and the mosfet Q5 closed: the 15V voltage will thus be fed via the line 16 to the driver supply input by keeping it operative.

As indicated, if an overload condition occurs, the diode D2 will become conductive and the forward voltage Vf across the diode D2 will be imposed between the emitter and base of the transistor Q3, causing it to turn off.

In this case the transistor Q4 will be turned on by the voltage of the capacitor C2 every time the oscillations at the intermediate point between the diodes Dl and D2 reach the value OV, so that and the gate of Q5 is short circuited to ground. This will result in the output line 16 being disconnected from the d.c. (e.g. 15V) voltage; the driver D will thus be caused to shut down (i.e. turn off) the converter.

The RC network including the capacitor C3 filters all the voltage pulses on the base of Q4, providing a constant "on" time of QA during the overload condition. The transistor Q4 will thus be kept in a steady condition, unaffected by these pulses, when the transistor Q4 drives the output switch Q5 in the overload protection condition.

If the overload condition persists, any attempt to restart the converter will result in the same circuit behaviour as described in the foregoing: i.e. the overload protection circuit will once again turn off the power stage, whereby operation will be restored only if the overload condition disappears.

Without prejudice to the underlying principles of the invention, the details and the embodiments may vary, even appreciably, with respect to what has been described by way of example only, without departing from the scope of the invention as defined by the annexed claims. For instance, those of skill in the art will appreciate that the same behaviour as demonstrated herein in respect of the diode D2 could be achieved by using a sense resistor in the place of the diode D2, which will however involve a higher power dissipation. Also, such a diode or resistor acting as current sensing elements may be replaced by cascade diodes/resistors with the input for the circuit 10 derived form an intermediate point in the cascade.