FIELD OF THE INVENTION

This invention relates to control circuitry for inductive ignition systems for internal combustion engines, and more particularly for simplifying the driver circuitry of the ignition coil power switches for coil energizing and spark firing, and for partially temperature compensating simple current sensor means used to set and control the peak coil charging current and power converter current.

BACKGROUND OF THE INVENTION AND PRIOR ART

There is a move in the automotive industry to distributorless ignition systems of one coil per spark plug, and particularly towards plug-mounted coils. Motivations for this are more compact ignition, elimination of electromagnetic interference, and higher ignition efficiency (no distributor or spark plug wires), as well as other reasons.

Such distributorless ignition systems require one power switch per ignition coil, typically from four to eight for four to eight cylinder engines. Each switch must have a driver circuit to turn the switch on and off to build up the required primary current and energy in the ignition coil and deliver it as an ignition spark at the spark plug in the engine cylinder. Such multiple driver circuits can be more complex and costly than required. Furthermore, the ignition triggering and cylinder phasing (sequencing) circuits may also be complex, especially for retrofit applications where a more universal but simple form of triggering and phasing is required. Therefore, circuits that can simplify the muhiple driver aspects of such distributorless ignitions, or more generally ignition circuits with multiple coils and power switches, and simplify the trigger and phase input requirements, are desirable. Partially temperature compensating simple forms of current sensing means associated with the power switches, and power supply switching if used, is also desirable.

SUMMARY OF THE INVENTION

The invention comprises a simple form of ignition power switch driver circuit based on the principle of keeping the gates of the ignition power switches Si (assumed to be Insulated Gate Bipolar Transistors, IGBTs) in the low or disabled condition by means of one driver transistor per power switch, or one driver diode per power switch with only one driver control transistor shared among the multiple power switches and driver diodes, and firing the ignition system by energizing transformer coil Ti by disabling the driver transistor (or transistors) and applying a signal to the gate of the appropriate switch Si to turn it on, which is then turned off when the coil set peak primary current Ipo has been attained by turning on the driver transistor(s) which disable, i.e. pull to a low or ground level, the gates of the power switches, to turn off all the power switches including the one that was turned on irrespective of whether it still has a drive signal being applied to its gate. A comparator is preferably used with the driver circuit to define the on-time (dwell-time) of the power switch by turning off the disabling driver switch during the coil energizing time (dwell-time).

A preferred automotive driver circuit for a 4-cylinder engine with one coil and power switch per cylinder, the reference engine and ignition for the present disclosure, has four diodes, one driver switch which pulls all four gates of the power switches to ground, or sufficiently close to ground to disable them when the coils are not being energized, and a coil charging duration controller which is preferably a comparator which senses the coil peak primary current Ipo and switches states when Ipo has reached a set point or reference voltage level. The gates of switches Si are driven by any of a number of ways, one way employing an octal counter for an engine with up to eight ignition coils switches Si whose outputs are connected to the gates of the power switches Si through resistors. For the preferred simple forms of current sensor means using the base- emitter voltage Vbe of a transistor as the sense voltage, simple diode means are disclosed to partially compensate the negative temperature coefficient of Vbe.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a partial circuit drawing of an embodiment of the invention, in which one driver switch QDi is used per power switch Si of a transformer ignition coil Ti of a multi-cylinder engine using a conventional inductive ignition system with current sensors at the emitters of the power transistors Si, and also including a simplified ignition trigger input circuit.

FIG. 2 is a partial block diagram, partial circuit drawing of a preferred embodiment of the invention, applied to an advanced form of inductive ignition system, called Hybrid Inductive Ignition, HBl, disclosed elsewhere, wherein one driver switch QD is used for multiple ignition coils Ti and gate diodes Di, and wherein both a preferred simplified trigger input circuit and phase circuit are shown for use in conjunction with an octal counter for operating an engine with up to eight ignition coil switches Si.

FIG. 3 depicts a preferred form of HBl ignition circuit which uses diodes in place of, or in addition to, resistors in the current sensing circuits to partially compensate the negative temperature variation of the base-emitter voltage Vbe of the current sense transistor.

FIGS. 4a and 4b are graphs of typical diode current versus diode voltage Vd with temperature as a parameter showing the diode's negative temperature variation of voltage Vd which is used to partially compensate the negative temperature variation of the base-emitter voltage Vbe of the sense transistor.

FIG. 5 depicts the use of diodes for partial temperature compensation of base-emitter voltages Vbe of two sense transistors where a single sense point is used, as is preferred with the boost converter powered HBl type ignition system.

DESCRIPTION OF PREFERRED EMBODIMENTS

FIG. 1 is a partial circuit drawing on an embodiment of the invention applied to a standard automotive inductive ignition system including an input triggering and disabling circuit. The ignition assumes operation from a standard 12 volt car battery 1 (voltage Vb) with two ignition coils 2a and 2b of several possible shown (also designated as TI, T2, or more generally Ti, where the subscript "i" designates the "ith" transformer coil). Each coil has primary winding 3 of turns Np and inductance Lp and coil primary leakage inductance Lpe, and secondary windings 4 of inductance Ls. The coils each have a main or coil power switch 5a, 5b, ... , (SI and S2 shown), and spark gap 6. All the switches have a common emitter connection (Insulated Gate Bipolar Transistors, IBGT's, shown for switches Si) with current sensor resistor 7 to ground.

In operation, switch Si is turned on by a controller and switch driver for the "dwell" period Tdw to build up a prescribed peak or "break" current Ipo developing a voltage Vsense across sensor resistor 7, and then opened to deliver the energy El stored in the magnetic core to the secondary coil circuit, where:

El = l/2»Lp«Ipo ² , where "•" denotes multiplication.

An embodiment of the controller and switch driver comprises an optional gate resistor Ri (8a, 8b for switches 5a, 5b) of switch Si, drive NPN transistors QDi (9a/QDl, 9b/QD2 for switches 5a, 5b) or other switch type with collectors connected to the resistors Ri and their emitters grounded. Drivers QDi have their bases connected through optional current limiting resistor 11 to open collector output of comparator 10 which is normally high through pull-up resistor 12 connected to switched battery voltage Vcc. The non-inverting input of comparator 10 is at a reference voltage V'ref, e.g. 2.5 volts, and the inverting input is connected to the output point 10a of a trigger and control circuit, which is normally low, setting the comparator 10 output high, all the drive switches

DQi on, and all the coil power switches Si off.

The gate of each switch Si is connected to trigger points Tri (Tri and Tr2 for gate switches SI, S2) through resistors 13a, 13b, ... , (for switches SI , S2 respectively) which are sequentially turned on to energize each coil Ti in turn. In this design, it is assumed that one of the trigger points Tri is high at all times (and the others are held low) designating which switch Si should be operating, and comparator 10 provides the dwell time duration control by turning off all the drive switches QDI, QD2 QDn, which enables all the power switches SI, S2,

.... Sn, allowing switch Si to be turned on from trigger Tri. Once coil Ti is energized, i.e. set or peak current Ipo flows through sensor resistor 7, comparator 10 inverting input is pulled low (by transistor 14), its output goes high, all driver switched QDI, QD2, ..., QDn, are turned on, thereby turning off all the power switches including switch Sj which was on.

In this trigger input design, turn-off of the power switches is accom¬ plished by sensor NPN transistor 14 (or other switch or circuitry) whose base is connected to the sense point 7a of current sensor 7 and its emitter to ground. The trigger input circuit shown comprises differentiating input circuit made of up of capacitor 15 in series with the trigger input Tr and shunt resistor 16, of values, for example, 0.05 microfarads (uF) and 22 kilohms (22K) respectively, followed by series time delay resistor 17, e.g. 2.2K value, shunted by voltage setting zener 18, e.g. 6.2 volt zener, and series isolating diode 19 through which timing capacitor 20 (of value, e.g. 0.010 uF) is charged, across which is override discharge resistor 21. Turn-off transistor 14 is across the capacitor with its collector connected to the inverting input 10a of the comparator 10 (representing the control point 10a) and its emitter grounded. When a positive trigger signal is received at the trigger input Tr it is differentiated (if it is a square wave versus pulse) and clipped by zener 18, rapidly charging dwell-time capacitor 20 with a time delay due to (time delay) resistor 17, making control node point 10a high and enabling all the power switches. Capacitor 20 remains charged (slowly discharging through discharge

resistor 21) and is discharged when control transistor 14 is turned on, to in turn disable all the power switches (including switch Si which was on). The value of the discharge resistor 21 is preferably selected to discharge dwell-time capacitor 20 to the reference voltage V'ref in a time longer than the normal dwell time but not much longer to provide a protective override function.

A simplification of the driver circuit is shown in FIG. 2 wherein one driver switch QD, 22, is used in place of the multiple driver switches QDI, QD2, ..., QDn, with associated isolating diodes Dl , D2, ..., Dn (22a, 22b for switches 5a, 5b). Like numerals represent like parts with respect to FIG. 1. In this figure is also shown a preferred embodiment power converter and isolation circuit for raising the battery voltage to a higher voltage Vc, which is disclosed elsewhere and shown here as block 23. At its output it includes energy storage capacitor 24 with diode 25 across it which is charged to a voltage Vc from the power converter 23, and includes the current sensor resistor 7 between it and ground and the sensor transistor 14a whose base is grounded in this case and emitter connected to the sense point 7b since the sense voltage is negative in this case (versus positive in the case of FIG. 1) and equal to transistors 14/14a base-emitter voltage (of typically approximately 0.6 volts).

In this drawing is also shown optional saturating inductor 26 with optional diode 26a across it to reduce the coil output voltage on power switch Si closure as disclosed elsewhere. Also shown is snubber circuit comprised of isolation diodes 27a, 27b, connected to each collector of the IGBT power switches Si with their cathodes interconnected to a snubber capacitor 28 with series resistors 29a and 29b shunted across it to allow capacitor 28 to discharge and to provide low control voltage at node point 30. Node point 30 is connected to the base of trigger input disabling transistor 31 (which in FIG. 1 played the role of the control transistor 14). Snubber circuit plays the usual role of storing and dissipating the high frequency energy associated with both the coil leakage inductance Lpe and optionally saturating inductor 26 inductance.

The input trigger circuit is the same as that as shown in FIG. 1 except that transistor 31 across the dwell time capacitor 20 is a disabling switch which maintains capacitor 20 grounded for a period of time following power switch Si opening (versus switch 14a which actually controls and turns off switch Si). Also shown is an alternate trigger input, designated TrQ, which is a pull-to-ground trigger (usually an open collector connection point). This trigger input has a pull- up resistor 32 to Vcc and an optional isolating diode 33 in series to the trigger input TrQ. To resistor 32 is connected debounce and isolating circuit made up of paralleled resistor 34 and isolation diode 35, in turn connected to isolation capacitor 36 which blocks Vcc from interfering with the trigger input.

Operation of trigger point TrQ occurs by it being pulled low by an open collector or ignition points and then allowed to go high. When it goes high, a signal passes through isolation diode 35 and capacitor 36 to provide a pulse at input of resistor 17 as with the other trigger circuit Tr, to charge up dwell-time capacitor 20 to enable the power switches Si, as already described, with turn-off as already described.

In this preferred embodiment is shown a circuit for sequencing the coil firings (versus in FIG. 1 where it was assumed that properly sequenced signals Tri, Tr2, ..., Trn, were already provided). This circuit requires a coil firing phasing input and the use of a sequencer, such as an octal counter 37 which is the preferred device shown.

The phasing circuit is modelled after the trigger circuit so that components that play similar roles as in the trigger circuit are given the same numerals with the suffix "a". The PhsQ input is a pull-down phase input and employs pull-up resistor 32a, paralleled resistor 34a and diode 35a and isolating capacitor 36a. The positive signal phase input Phs uses differentiating capacitor 15a and resistor 16a. However, beyond that point the circuit differs from the trigger circuit in that it uses an emitter-follower NPN transistor 38 to provide a high impedance to the two phase inputs, with its base connected to input base

resistor 17a, its collector connected to a reference voltage, e.g. 5 volts, and its emitter to capacitor 20a and discharge resistor 21a. The base-emitter diode of the transistor plays the isolating role of diode 19, and the reference voltage Vref provides the limiting reference voltage for the non-inverting input of comparator 10a (so diode 18a can be a simple diode versus a zener in the case of diode 18).

In this case (versus for the case of the trigger circuit), comparator output is normally low, with its inverting input connected to a reference voltage V'ref well below Vref, e.g. 2.5 volts. This voltage (already proposed) is attainable by making a resistive divider with two resistors 39 and 40 and Vref. The output of the comparator 10a has pull-up resistor 12a to a voltage Vx, preferably higher than 12 volts to be able to drive industrial type IGBT's which require higher gate drive than more conventional clamped ignition IGBTs. Likewise, supply of comparator 10, clock (CLK) input and VCC input of octal counter 37 are connected to Vx. By connecting output of trigger comparator 10 to the enable (ENA) input, and output of phase comparator 10a to the reset (RST) input, as disclosed in U.S. patent # 5,558,071, proper phasing and actuation of the octal counter 37 outputs connected to the trigger resistors 13a, 13b can be obtained.

That is, with the clock (CLK) input kept high, the outputs of the octal counter will shift when sequential low signals (GO) are received at the enable (ENA) input. Operation of the circuit is known to those versed in the art.

While in the examples of applications of the switch driver it is assumed that one switch is activated at each ignition firing, a number of switches can be simultaneously activated, e.g. two switches at a time as in an ignition with two spark plugs with per cylinder and one coil and switch per spark plug, and so on. Moreover, while a preferred embodiment of the switch driver circuit is the use of one diode Di per switch, one control switch QD, and one comparator, other combinations are possible, including using a different comparator or other kind of duration control device or scheme, the main feature of the simplified switch driver circuit being the enabling and disabling of all the switches during

the ignition coil energizing, on-time, and off-time respectively, and then separately selecting the appropriate switch to be turned on, making for a very simple and effective design. It is also noted that the resistors 13a, 13b ... , through which is conducted the turn-on current for the gates of the power switches Si, can be large, e.g. in the kilo-ohm range, while the turn-off resistors 8a, 8b, ..., FIG. 1, will be much lower (or absent as shown in FIG. 2, although a small resistor may be included in series with switch 22 (QD)).

A feature of the present invention is simplicity and low cost of the control circuitry. In the embodiment of FIG. 2 the current sensing circuit is very simple, comprised of one transistor 14a and resistor 7, as is also disclosed in U.S. patent 5,558,071. In that patent is disclosed use of a negative temperature coefficient resistor (with a resistor in series with it) across the sense resistor to compensate, over a temperature range, the negative temperature coefficient of the base-emitter voltage Vbe (approximately minus 2 millivolts per °C) of the sense transistor. However, in the present case sense resistor 7 is about one order of magnitude smaller so the compensation may be more difficult and expensive.

FIG. 3 depicts a simple and low cost way of partially compensating the reduction of the sense transistor base-emitter voltage Vbe with temperature, by paralleling the resistors in the current sensing circuits with diodes which carry some or essentially all of the current to be sensed. Only partial compensation is required since it is desirable to have lower peak currents at higher temperatures, and vice versa. Diode 41 (Dc) is placed across sense resistor 7 to partially compensate the temperature variation of Vbe of sense transistor 14a. A second current sense circuit for the flyback version of power converter 23 (FIG. 2) is shown, comprising sense transistor 14b and resistor 42 with diode 43 (Ds) across it for controlling the current in a flyback converter. Like numerals represent like parts with respect to the earlier figures. The flyback converter, disclosed elsewhere, has a transformer 44, FET switch 45, and output diode 46 to raise battery voltage to a preferred voltage Vc of 30 to 60 volts.

10

In the case of the power converter sensor, one can select a diode 43, as depicted in FIG. 4a, to have a significant diode voltage Vd reduction with temperature at the sense current Is,, shown as 6 amps at 25 °C (for Vsense of 0.6 volts), so that at 125°C the sense current Is ₂ is higher than it might otherwise be, i.e. Isj (shown as 4 amps) had the diode 43 not been used. Resistor 42 is included to allow for flexibility of design and for holding the sense point 7b close to ground for shorting out any reverse current. Diode 43 is preferably a Schottky diode to provide low forward drop (although a standard diode may be used if higher sense voltage is used, i.e. by making transistor 14b a Darlington. In this application, the sense circuit is used in conjunction with a timing resistor 47 as disclosed elsewhere.

The current sense circuit for sensing the peak ignition current Ipo works similarly, with curves of FIG. 4b depicting the characteristics of the diode 41 (Dc) which must handle much higher currents. Two or more diodes may be used in parallel, and the sense voltage Vsense may be made larger as described, i.e. by making transistor 14a a Darlington, or using other means to increase the sense voltage (values are not indicated in FIG. 4b to indicate this variability). In this case, as shown in FIG. 4b, the peak current Ip, is lowered from 30 amps to a peak current Ip ₂ of 26 amps, versus an overly low current Ip ₃ of 21 amps attained without the partial compensating effect of the diode 41.

A particularly simple case of the use of compensating sense diodes is when the same sense point is used, as shown in FIG. 5, which depicts for power supply 23 a boost converter with inductor 50, switch 45, and output diode 46. Like numerals represent like parts with respect to the earlier figures. In this case, sense point 7c is the same point for both current sense circuits, although sense transistor 14b has its base connected to the sense point 7c and emitter to ground since it is sensing a power converter positive, versus negative, current. Note that in this figure saturating inductor 26 is shown, which is not essential in some applications.

In the simplest form of a common sense point 7c (for two sense circuits), resistor 7 can be used as the sense resistor for transistor 14b (not requiring sense diode 43) since the sense current is much smaller, e.g. 3 to 15 amps versus about 30 amps for the peak coil charging current. Resistor 7 is typically between 0.04 and 0.2 ohms, depending on the power output of the power converter and type and design, i.e. boost or flyback. Resistor 7 preferably has negative temperature coefficient as is achievable with carbon resistors, or special design resistors, or paralleled resistors as already discussed, to provide some temperature compensation. Since typically ferrite core material is used for the cores of the magnetic components 44, 50 of the power converters, and core saturation flux density drops by about 30% for a 100 °C increase in core temperature, only small temperature compensation may be required for the power converter sense circuit. Therefore, a simple sense circuit for common sense point sensing is achievable with a total of only four main components, two sense transistors, one diode, and one resistor (with parallel resistor components as discussed). If one uses sense transistors with differing base-emitter voltages Vbe, e.g. 0.6 volts for 14b and 1.2 volts for 14a, and the two sense currents differ by a factor of only two to three, then the sense resistor can carry the dominant current even for the higher current sense circuit (transistor 14a) and the single sense diode may be either eliminated or be a lower cost, lower current diode. Clearly, with common point sensing, both currents cannot be flowing simultaneously, i.e. the power converter must be turned off during coil charging, as is the preferred case.

By using the various simplifying control features, circuits, and components disclosed herein, one achieves the benefits of an ignition system with outstanding spark characteristics and acceptable cost and relatively simple design. Since certain changes may be made in the above circuits and method without departing from the scope of the invention herein disclosed, it is intended that all matter contained in the above description, or shown in the accompanying drawings, shall be interpreted in an illustrative and not limiting sense.