Transmitter, power amplifier and filtering method

Field

The invention relates to filtering in transmitters and power amplifiers in transceivers, especially in RF transmitters and RF power amplifiers.

Background

In multiradio concepts, where the number of different radio systems is increasing all the time, the interoperability of different radio systems is challenging. Different radio systems operating on different frequency bands are required to operate properly without disturbing each other, even if they are op- erating at the same time. This sets strict requirements for both receiver and transmitter chains especially in transceivers when a transmitter of a transceiver is having high power levels at the same time when a receiver of a transceiver is receiving a weak signal.

In the transmitter, a power amplifier is used to amplify the signal to be transmitted to the required power level. However, power amplifiers have usually a broad bandwidth. Therefore, they amplify the signal to be transmitted not only on the desired transmitter band but also outside the desired band. In such a case, an unwanted receiver band may be amplified with the same gain. After the power amplifier, there has to be a tight filter that filters these un- wanted receiver band signals away before transmitting the power from the antenna. The requirements of such filters are strict causing losses also to the transmitter path. This, in turn, has to be compensated for with increased output power from the power amplifier. This decreases the total transmitter efficiency and leads to increased power consumption and increased heat in portable transmitters such as mobile phones.

Furthermore, the noise requirements for the transmitter path before power amplifier are very strict so as to guarantee that the noise level before the power amplifier will not be too high. This is required to ensure that the filters after the power amplifier can reduce the receiver band noise level to be low enough. If no filtering is performed before the power amplifier, the capability of the filters after the power amplifier sets a limit to the maximum gain of the power amplifier and can increase the required output power of the transceiver block before the power amplifier.

Traditionally, most of the unwanted noise filtering is performed after the power amplifier in a front-end module of the transmitter, which consists of

switches and filters. Since the gain of the power amplifier is constant in both transmitter and receiver bands, it sets strict requirements for the filter after the power amplifier to decrease the signal level in receiver bands.

Figure 1 illustrates a section of a traditional front end of a transmit- ter. The front end comprises a power amplifier 100 and a band pass filter 102 connected to the output of the power amplifier. The power amplifier comprises multiple amplifier stages 104, 106 and a matching circuit 108 after the stages. Figure 2A illustrates an RF signal and noise strengths at the input of the power amplifier. Frequency is shown on the X-axis and signal strength is shown on the y-axis. A TX-arrow 200 denotes the signal strength on the desired transmission frequency. In addition, noise level 202 is shown. RX denotes the receiver band. Figure 2B illustrates the RF signal strength at the output of the power amplifier. The RF-signal has been amplified on all frequency bands, as the power amplifier is a broadband amplifier. Both the desired transmission 202 and the noise signal 202 have been amplified. Figure 2C illustrates the RF signal strength at the output of the band pass filter 102. The noise signal strength 202 on the RX band has been somewhat reduced.

Brief description of the invention

An object of the invention is to provide an improved solution for fil- tering in a transmitter and a power amplifier. According to an aspect of the invention, there is provided a transmitter comprising a power amplifier amplifying an RF signal and having multiple stages, and a local oscillator, the power amplifier comprising between at least two stages of the power amplifier an impedance circuitry for forming an impedance at a frequency related to the frequency of the local oscillator, and a switch for switching the impedance of the impedance circuitry to RF frequency.

According to another aspect of the invention, there is provided a power amplifier in a transmitter comprising a local oscillator, the power amplifier amplifying an RF signal and having multiple stages, the power amplifier comprising between at least two stages of the power amplifier impedance circuitry means for forming an impedance at a frequency related to the frequency of the local oscillator, and switching means for switching the impedance of the impedance circuitry means to the RF frequency.

According to another aspect of the invention, there is provided a fil- tering method, the method comprising: amplifying an RF signal in multiple amplifying stages of a power amplifier, forming an impedance in an impedance

circuitry between at least two stages at a frequency related to the frequency of the local oscillator of the receiver, and switching the formed impedance to the RF frequency.

According to another aspect of the invention, there is provided a fil- tering method in a transmitter, the method comprising: amplifying an RF signal in multiple amplifying stages of a power amplifier of the transmitter, forming an impedance in an impedance circuitry between at least two stages at a frequency related to the frequency of the local oscillator of the receiver, and switching with a switch arrangement the created impedance to the RF fre- quency.

According to yet another aspect of the invention, there is provided a transceiver comprising a transmitter with a power amplifier amplifying an RF signal and having multiple stages, and a local oscillator, the power amplifier comprising between at least two stages of the power amplifier impedance cir- cuitry means for forming an impedance at a frequency related to the frequency of the local oscillator, and switching means for switching the impedance of the impedance circuitry means to the RF frequency.

The embodiments of the invention provide several advantages. The filtering requirements after the power amplifier can be relaxed. In transceivers, receiver-band filtering requirements of the duplex filters can be relaxed. This relaxed requirement for attenuation decreases also the transmitter band losses in the duplex filter which in turn increases the total transmitter chain efficiency. Approximately an increase of 1% in power amplifier efficiency can be reached if the losses at the transmitter chain after the power amplifier decrease 0.1dB. Thus, already a saving of 0.5dB in losses may increase the power amplifier efficiency with 5% units. This decreases the heat that the power amplifier generates and, therefore, also the reliability of the power amplifier is increased. Furthermore, the gain of the power amplifier can be increased without adding extra filter at the input of the power amplifier. This means that the required out- put power from a radio frequency integrated circuit (RFIC) may be reduced.

The design of the proposed filtering arrangement is simple and it may be configured to be used on different frequency bands with minimal changes. The change of the frequency band used may be performed by software.

List of drawings

In the following, the invention will be described in greater detail with reference to the embodiments and the accompanying drawings, in which

Figure 1 illustrates an example of the front end of a prior art trans- mitter;

Figures 2A to 2C illustrate examples of signal strength on different parts of the prior art transmitter;

Figure 3 illustrates an example of a telecommunication system in which embodiments of the invention are applicable; Figure 4 illustrates an example of the front end of a transceiver in which embodiments of the invention are applicable;

Figure 5 illustrates an example of the front end of a transmitter of an embodiment of the invention;

Figures 6A to 6C illustrate examples of signal strength on different parts of the transmitter of an embodiment of the invention;

Figure 7 illustrates an example of a tunable band pass filter of an embodiment of the invention;

Figure 8 illustrates another example of a power amplifier of the front end of a transmitter according to an embodiment of the invention; Figures 9A to 9C illustrate another examples of a band pass filter;

Figure 10 illustrates another example of a band pass filter;

Figure 1 1 illustrates another example of a tunable band pass filter of an embodiment of the invention;

Figure 12 illustrates the simulated transfer function of the filter of Figure 11 ; and

Figure 13 illustrates yet another example of a tunable band pass filter.

Description of embodiments With reference to Figure 3, let us examine an example of a telecommunication system in which embodiments of the invention are applicable. Figure 3 shows a base station 300 which is in connection with terminal equipment 302, 304, 306 and 308. The terminal equipment 302 and 308 may also be in contact with another base station 310. The base station 300 and the ter- minal equipment 302, 304, 306 and 308 comprise an RF transceiver. Embodi-

merits of the invention may be applied both in base stations and in terminal equipment.

Different multiple access methods may be used in the telecommunication system in which embodiments of the invention are applicable. The sys- tern may utilize CDMA (Code Division Multiple Access) WCDMA (Wide CDMA) or TDMA (Time Division Multiple Access). The access method used is not relevant regarding the embodiments of the invention.

Embodiments of the invention are not limited to transmitters, transceivers or power amplifiers of transmitters of telecommunication systems, but they may be applied to any transmitter, transceiver and power amplifier of a transmitter, especially to an RF transceiver, an RF transmitter and an RF power amplifier.

Figure 4 illustrates an example of the front end of a transceiver in which embodiments of the invention are applicable. The transceiver comprises an antenna 400 connected to a transmitter 402 and a receiver 404. The front end of the transmitter 402 comprises a power amplifier 406 and an external filter 408 between the antenna and the amplifier. The filter may be a SAW or a BAW filter, which blocks the signal received by the receiver 404 to reach the power amplifier 406 of the transmitter 402 and filters the signal amplified by the power amplifier. Also other filter arrangements may be used. The front end of the receiver 404 comprises a pass band filter 410 and a low noise amplifier 414 placed in series.

Figure 5 illustrates a section of the front end of a transmitter according to an embodiment of the invention. The front end comprises a power ampli- fier 500 and a band pass filter 502 connected to the output of the power amplifier. Also, a controller unit 514 of the transmitter is shown in Figure 5. The power amplifier 500 comprises multiple amplifier stages 504, 506 and a matching circuit 508 connected to the output of the last stage. The power amplifier further comprises a band pass filter 510 between at least two amplifying stages of the power amplifier 500. In this embodiment, filtering is performed inside the power amplifier stages so that the filtering requirements of the filter after the power amplifier may be decreased. In addition, the output power and the noise requirements of the transmitter path before power amplifier are decreased.

The band pass filter 510 may be tunable so that the same power amplifier may be used for different bands and modes, such as GSM1800, GSM 1900, WCDMA1900 and WCDMA2100. The filter 510 may be tunable

with frequency and bandwidth. The filter may be controlled by a control signal 512 originated from a controller unit 514 of the transmitter, for example.

Figure 6A illustrates the RF signal strength and noise strengths at the input of the power amplifier 500. Frequency is shown on the X-axis and signal strength is shown on the y-axis. A TX-arrow 600 denotes the signal strength on the desired transmission frequency. In addition, a noise level 602 is shown. RX denotes a receiver band. Figure 6B illustrates the RF signal strength at the output of the power amplifier 500. The RF-signal has been amplified at all frequency bands, as the power amplifier is a broadband amplifier. However, due to the band pass filter 510, the noise signal 602 at the RX receiver band frequencies has been attenuated considerably in comparison with the prior art solution of Figure 2B. Figure 6C illustrates the RF signal strength at the output of the band pass filter 502. The noise signal 602 strength at the RX band has been reduced even more and the attenuation is better in com- parison with the prior art solution of Figure 2C.

Figure 7 illustrates an example of a band pass filter 510. The filter comprises a resistor 700 having a resistance of R and four capacitors 702, 704, 706 and 708 placed in parallel. The capacitors have capacitances C1 , C2, C3 and C4, respectively. Each capacitor is placed behind a switch 710, 712, 714, and 716. The switches are controlled to switch the four parallel capacitors alternately so that each of them is on 25% of the time cycle. The switching frequency of the capacitor switches 710, 712, 714, and 716 is related to the local oscillator frequency. If the input RF frequency differs from the switching frequency of the capacitor switches 710, 712, 714, and 716, the capacitors are charged with the frequency difference and create a band pass filter response with a corner frequency of

1 τ = •

2πRC where C=C1 +C2+C3+C4.

Figure 8 illustrates a power amplifier 800 of the front end of a transmitter according to an embodiment of the invention. Here, the band pass filter of Figure 7 is used. The power amplifier comprises multiple amplifier stages 504, 506 and a matching circuit 508 connected to the output of the last stage. The power amplifier further comprises a band pass filter 802 connected between at least two amplifying stages 504, 506 of the power amplifier. The filter comprises capacitors and switches as described in connection with Figure

7. The power amplifier further comprises matching circuits 804, 806 before and after the band pass filter 802.

The filter 802 may be tuned to different frequency bands by adjusting the frequency of the signal 808 which controls the switches. The frequency may be derived from local oscillator 810 of the transmitter and it may be controlled by the controller unit 514 of the transmitter, for example.

The operation of the band pass filter 510 between at least two amplifying stages of the power amplifier is further described in Figures 9A, 9B and 9C which are examples among others of a simplified schematic view of the filter 510. The embodiments of Figures 9A, 9B and 9C use MOSFETs (metal- oxide-semiconductor field-effect transistors) as switches.

In an embodiment of the invention shown in Figure 9A, the filter comprises MOSFET switches 900, which are switched with signals 904, 906 between on and off states. The frequency of the signals 904, 906 is related to LO (local oscillator) signal. The filter further comprises capacitors C 902 connected to the switches 900. As the MOSFETs 900 are switched between on and off states the capacitors are then switched between RF-P and RF-M ports which act as input to the MOSFETs. Referring to Figure 8, the ports RF-P and RF-M receive a signal from the matching circuit 804 having resistance R. It should be noted here that the resistance may be a general impedance of the form: Z= a+bj ohms.

In an embodiment of the invention, the frequency of the signals 904 and 906 is not exactly the same as the frequency of a local oscillator signal but derived from it. If the frequency of the incoming RF signals in ports RF-P and RF-M differ from the frequency of the signals 904, 906, then the capacitors C 902 will be charged with a signal the frequency of which is the difference of the RF and signals 904, 906. The driving impedance is the impedance R of the matching circuit 804. Therefore the result is impedance filtering at frequency F _L O+FRC, where F _L o is the LO-signal frequency and F _RC is the corner frequency of the resistance R and the capacitor C 902 (i.e., 1/2πRC).

This means that the filter 510 is a band pass filter with pass band corner frequencies (also called -3 dB frequencies or half-power frequencies) FLO+FRC and F _L O-FRC, respectively. The shape of the filter 510 is very steep, since the attenuation increases as a function of the RC constant corresponding to the low frequencies.

Let us study an example. If the LO frequency is 2GHz and an RC time constant is equivalent to 2MHz, then the signal of frequency 2.002GHz attenuates 3dB. If we had a standard RC -3dB point at that frequency, 2OdB attenuation would be reached at the frequency of about 20.002GHz (i.e. one decade away). With the transferred-impedance filter 510, the 2OdB attenuation will be reached at 2.020GHz (i.e. one decade away from the RC frequency 2MHz). Thus the low frequency (defined by the RC constant) is transferred to the RF frequencies. This is a significant improvement over the possible prior art solutions. Thus, in an embodiment of the invention, the filter comprises means for forming impedance at a frequency derived from the frequency of the local oscillator and switches for switching the impedance to the RF frequency.

It is noted that other impedances can be transferred to higher frequency filtering using the methodology described in the present invention. In the embodiment of Figure 9A, capacitors 902 were used as impedance in the filter 510. However, any impedance Z may replace the capacitors. The capacitors 902 in Figure 9A can be replaced with an LC-resonator or with a combination of capacitors and an amplifier, for example. Figures 9B and 9C demonstrate LC resonator options. In the embodiment of Figure 9B, inductors L 908 are added in series with the capacitors C 902 (compared to Figure 9A) and the center frequency of the filter (or a reference frequency) is given by F _L O-FLC or F _L O+FLC, wherein F _L o is the local oscillator frequency 904, 906 provided to the filter 510 and F _L c is an LC resonant frequency given by F ₁₀ = l/2π yflc . F _L c can be made as low as 900 kHz, for example. In this case, the resultant center frequency of the filter could be F _LO -900kHz or F _LO +900kHz.

Moreover, according to an embodiment shown in Figure 9C, an inductor L 910 is added in parallel with the capacitors C 902 (compared to Figure 4A) with an LC resonant frequency F _L c given by FL _C = l/2π 4∑C . It is noted that for the resonant curve with the center frequencies F _L o +Fι_c and F _L o -Fι_c _> the corner frequencies (-3dB frequencies) of the pass band depends on the inductor L 910 (in addition to being a function of the resistance R and the capacitors C 902). Thus, if the inductor L 910 and the capacitor C 902 are placed in parallel, then there are narrow pass bands around the resonant frequency at FLO+FLC and F _L O-FLC where F ₁₀ = l/2π yflc .

The inductors 908 or 910 can be generated, e.g. from capacitors with operational amplifiers (which imitate inductors) or by making a second (or higher) order filter by generating an impedance with a magnitude degrading as a second order filter response thus providing a low area, high performance filter systems.

There are a lot of variations of the above-presented structure of the filter 510. It is noted that the NMOS switches used in examples of Figures 9A, 9B and 9C can be of other types.

Also, it is clearly understood that the technology described in the in- vention can provide a broad range of LC resonant frequencies and impedances transferred to filtering of radio frequencies, according to the present invention. Furthermore, the examples presented in the above-described Figures use differential (i.e., both positive and negative) signals but the method of the present invention can be also used in single-ended systems with only one sig- nal line.

The frequency of the signals 904, 906 is related to LO (local oscillator) signal. The frequency may be derived from the frequency of the local oscillator signal or it may be locked to the frequency of the local oscillator signal. The signals may be generated in the local oscillator or in a separate oscillator. Figure 10 illustrates a more complete example of a band pass filter

510. In the example of Figure 10, the filter comprises separate I- and Q- branches 1000, 1002. As input there are signals RF-P and RF-M as in the example of Figure 9A. In this embodiment, there are four signals derived from local oscillator signal. On the l-branch 1000 of the filter there are F _L O-IP 904A and FLO-IM 906A. On the Q-branch 1002 of the filter there are F _L O-QP 904B and FLO-QM 906B. The phase difference of F _L O-IP and F _L O-QP is 90 degrees, and the phase difference of F _L O-IM and F _L O-QM is like wise 90 degrees. The phase difference of FLO-IP and F _L O-IM is 180 degrees and the phase difference of F _L O-QP and FLO-QM is like wise 90 degrees. Figure 1 1 illustrates another example of a band pass filter 510 where a series LC circuit is utilized. In this embodiment, the filter comprises an inductance arrangement 1 100 connected in series with switches 1 110, 1 1 12, 1 1 14, and 1 1 16 and four capacitors 1 102, 1 104, 1 106, and 1 108. The capacitors have capacitances C1 , C2, C3 and C4, respectively. The switching fre- quency of the capacitor switches 1 1 10, 1 1 12, 1 1 14, and 1 116 is related to the local oscillator frequency. The inductance arrangement 1 100 is realized with

one or more inductances. Z _in and Z _0Ut represent matched input and output impedances.

In this embodiment, the inductance arrangement 1100 having an inductance L1 is in RF frequency and the capacitors 1102, 1104, 1106, and 1108 are in local oscillator frequency and transferred to RF frequency with the four switches. As the inductance arrangement 1100 is in RF frequency the value of L1 is small and it is easy to realize in an integrated circuit.

Figure 12 illustrates the simulated transfer function of the circuit of Figure 11. In this example, the values used were L1=2.1 nH, Cn=330pF, n=1...4, Zin=Z _O u _t =50W, and the resistance of the switches R _SW =2W. The switches are being switched with a local oscillator signal between on and off states, the local oscillator frequency being in this example 900MHz. In this example, the attenuation of the unwanted signal is greatest at 10MHz distance from the carrier frequency but it decreases further away from the carrier fre- quency. As the circuitry of Figure 7, this realization creates a mirror response into the other side of the local oscillator signal, creating a band-pass response with notches.

The distance of the notch can be further tuned by using variable capacitors instead of discrete capacitors with discrete values. This is useful es- pecially in systems where the duplex distance is not constant but can be variable (for example, WCDMA or 3.9G systems may utilize variable duplex distance).

Figure 13 illustrates yet another example of a tunable band pass filter. Frequency (in MHz) is on x-axis and power (in dBc) relative to the power of the transmitted signal is on y-axis. In this example, the filter comprises a second switch arrangement 1300, 1302 connected between the inductor arrangement 1100 and the switches 1110, 1112, 1114, and 1116 and four capacitors 1102, 1104, 1106, and 1108. The switches 1300, 1302 are switched with the local oscillator signal whereas switches 1110, 1112, 1114, and 1116 are switched with a double frequency.

When utilizing the embodiments of Figures 11 and 13 in a transceiver, the RX-band filtering requirements from duplexers of the transceiver can be relaxed. This relaxed requirement for attenuation decreases also the TX-band losses in a duplexer increasing the total transmitter chain efficiency. Approximately 1% increase in power amplifier efficiency can be reached if the losses at the TX chain after the power amplifier decrease 0.1 dB. Thus, already

the saving of 0.5 dB in losses can increase power amplifier efficiency 5% units by decreasing the heat that the power amplifier generates and through that increasing also the reliability of the power amplifier. In addition, the gain of the power amplifier can be increased without adding extra filter at the input of the power amplifier. This means that less output power is required from a radio frequency integrated circuit (RFIC).

Also, due to the inductance of the inductor arrangement the impedance levels for the switches 1110, 1112, 1114, and 1116 are increased. Thus, the size of the switches can be reduced and the drivers for these switches do not need to drive such large currents leading to decreased power consumption.

The embodiments of Figures 11 and 13 may be utilized in various apparatuses in addition to power amplifiers which are used here merely as an illustrative example. In an embodiment, the invention is applied to a multiband transceiver which supports several frequency bands. The transceiver may comprise more than one local oscillator. When the transceiver is transmitting and receiving on a given frequency band, the local oscillator of the given band is used and switched to the filter 510. The switching may be performed under control of the controller unit 514.

Even though the invention has been described above with reference to an example according to the accompanying drawings, it is clear that the invention is not restricted thereto but it can be modified in various ways within the scope of the appended claims.