Technical Field

The present application relates to a voltage detector and a voltage detector system. Background to the Invention

Sensors such as those used in medical devices can be powered by wireless power transfer or energy harvesting sources. In the case of sensors that are worn on the human body, these sources provide intermittent power due to the body's movement. For example, if the wearer of the device rests in a position such that a power receiving antenna is poorly oriented relative to the transmitter, then an input power lull occurs. This problem of intermittency occurs with the majority of energy generation and wireless power transfer techniques, and is only exacerbated when some of the system is situated on a moving body.

Figure 1 provides a schematic illustration of a sensor system that may be powered by intermittent sources. The intermittent source feeds an input rail VIN. A power management circuit converts VIN into a useful stable internal rail voltage VDD for the sensor sub-systems. The power management circuit requires power for its own internal control circuits, which is supplied from the VDD rail. In order to stop this quiescent power from unnecessarily draining power from the VDD rail, the power management is powered off during input power lulls. A voltage detector, the device under consideration here, senses the input voltage VIN, and power-gates the power management circuit via a PMOS transistor, which is switched by the voltage detector's open-drain output Three-terminal voltage detectors are used that do not require connection to a supply rail, and therefore, in principle provide always-awake input monitoring, even when the rest of the sensor system is powered off. Alternatively, as shown in Figure 1 , the voltage detector could be used to switch on a power supply, for example from a battery, thus minimising use of stored power to when there is a detection signal.

In practice, however, existing commercially-available voltage detectors have a significant input quiescent current, especially once activated. This quiescent current significantly reduces the source's output voltage, and thus prevents the use of wireless rectifying antennas (rectennas) with multiple voltage multiplying stages, and wearable energy harvesters such as ferroelectret textiles, because these have high source impedances of up to tens of ΜΩ. For these sources, input monitoring and input-dependent power gating is therefore not currently possible.

This problem is illustrated in Figure 2, which shows the operation of a commercially available voltage detector being used in the circuit of Figure 1. Figure 2 shows the source voltage VIN when the source is unloaded (dashed line), and also when loaded by the voltage detector (solid line). On reaching the detector's threshold, the quiescent current IIN increases sharply, dragging the input voltage down, thus switching the detector off again. The quiescent input current drops and the process begins again, resulting in a pulsing output VOD rather than an output pulse that lasts the duration of the incoming power, as required for the power-gating of sensor sub-systems. More importantly, the source voltage VIN has been affected in such a way that the supply to the power management circuit shown in Figure 1 is significantly reduced.

Another potential solution is the Power-On-Reset circuit (POR), which monitors an input without the need for a stable voltage rail. Figure 3 illustrates the operation of a basic POR pulse generator circuit. However, the output is a short reset-pulse, which does not provide information on the duration of the incoming source power. Figure 4 shows the desired behaviour of an ideal voltage detector, where the incoming voltage is preserved, the quiescent current is zero most of the time, and the output lasts the duration of the incoming power.

Many different types of low-power integrated circuits for monitoring voltage rails have been proposed, including those previously referred to. In one such circuit (as described in "A 330nA energy -harvesting charger with battery management for solar and thermoelectric energy harvesting" by K. Kadivrl et at) a voltage detector consists of a comparator and a band-gap reference to determine the input condition. However, the quiescent current of this circuit is 180 nA, mostly due to continuous flow of bias current through reference- voltage generating circuitry and the comparator. The voltage detection threshold of this circuit is 2.1V and the maximum operating voltage is 4V. Additional protection circuits could be used to protect the circuit from potentially damaging over voltage conditions, but by clamping the input voltage at 4V, a significant percentage of the source power would be lost.

A monitoring circuit consuming sub-nA current has also been proposed (Toshishige Shimamura, Mamoru Ugajin, Kenji Suzuki, Kazuyoshi Ono,Norio Sato, Kei Kuwabara, Hiroki Morimura, Shin'ichiro Mutoh: "Nano-Watt Power Management and Vibration Sensing on a Dust-Size Batteryless Sensor Node for Ambient Intelligence"). It operates without the need for a bandgap voltage reference for low power, but triggers only at a high input voltage.

Detectors which do not need comparators have been proposed, (e.g. by P. Chen, et al: "Startup Techniques for 95 mV Step-Up Converter by Capacitor Pass-On Scheme and VTH- Tuned Oscillator with Fixed Charge Programming" and X. Zhang et al: "A 0.6 V Input CCM/DCM Operating Digital Buck Converter in 40 nm CMOS"). Their quiescent current is therefore reduced to few nA. The technology used permits detection thresholds of a few hundred mV, but this also reduces the maximum allowed input voltage. In "Design and Realization of a Voltage Detector Based on Current Comparison in a 40nm Technology" by S. Y. Wu, W. B. Chen, N. Ning, J. Li, Y. Liu, and Q. Yu, a current comparison voltage detector is reported, which uses current comparison instead of a voltage comparator. The current reference is programmed by a logic control circuit, and provides the detector with three selectable thresholds. The power, however, is high at μ\ν and the voltage detection threshold is 1.1 V.

A battery supervisor circuit consuming 635 pW at 3.6 V is reported in "A 635pW Battery Voltage Supervisory Circuit for Miniature Sensor Nodes" by I. Lee, S. Bang, Y. Lee, Y. Kim, G. Kim, D. Sylvester, and D. Blaauw. This circuit achieves low power consumption through a 57 pA voltage reference and a low power comparator. The detection threshold, however, is 3.6 V, which is too high for many applications. In "248pW, 0.1 lmV/°C Glitch- Free Programmable Voltage Detector With Multiple Voltage Duplicator for Energy Harvesting" by Teruki Somey, Hiroshi Fuketa, Kenichi Matsunaga, Hiroki Morimura, Takayasu Sakurai, and Makoto Takamiya, a 248 pW voltage detector for energy harvesting is demonstrated. However, its maximum input voltage is 0.1 V, which is too low for many applications.

As will be appreciated from the discussion above, existing commercial discrete voltage detectors and integrated detection circuits are unsuitable for the continuous, non-invasive monitoring of high-impedance source voltages. A need therefore exists for a voltage detector which combines a high (greater than 10V) maximum input voltage, a low (lower than IV) detection threshold and a low (lower than 248pW) quiescent power consumption.

It is an object of embodiments of the present invention to provide a voltage detector and a voltage detector system that address the problems of the prior art.

Summary of Invention According to a first aspect of the invention there is provided a voltage detector comprising: a first voltage reference generator for generating a first voltage reference signal; a second voltage reference generator for generating a second voltage reference signal, wherein the second voltage reference signal is higher than the first voltage reference signal; a trigger, powered by an input signal to the voltage detector, and having an input for receiving either the first or second voltage reference signal and an output for generating a detection signal; and a switch for selectively connecting the input of the trigger to the first or second voltage reference signal, wherein: the switch is operative to connect the input of the trigger to the first voltage reference signal when the detection signal output by the voltage detector is low and is operative to connect the input of the trigger to the second voltage reference signal when the detection signal output by the voltage detector is high.

The first and second voltage reference generators of the voltage detector may each comprise a first transistor and a second transistor connected in series between the input signal and a common reference, wherein the first transistor has a different gate threshold than the second transistor.

The trigger may comprise a PMOS transistor and an NMOS transistor connected so as to form an inverter.

The first and second reference voltage signals produced by the first and second voltage reference circuits may be low enough to ensure that the inverter is in subthreshold mode when it switches.

The common reference may be ground.

The voltage detector may further comprise a third voltage reference generator for generating a third voltage reference signal, wherein the third voltage reference generator is configured to output the third voltage reference signal to the first and second voltage reference generators. According to a second aspect of the invention there is provided a voltage detector system for monitoring an input signal and outputting a detection signal when the voltage of the input signal meets a threshold, the voltage detector system comprising: a first voltage detector having an input for receiving the input signal and an output for outputting a detection signal, wherein the first voltage detector is configured to output the detection signal when the voltage of the input signal meets a first rising input voltage threshold; a second voltage detector having an input for receiving the input signal and an output for outputting a detection signal, wherein the second voltage detector is configured to output the detection signal when the voltage of the input signal meets a second rising input voltage threshold which is higher than the first threshold, wherein the first and/or second voltage detector is according to the first aspect of the invention, and the output of the second voltage detector controls a connection between the input signal and the input of the first voltage detector such that when the voltage of the input signal meets the second rising input voltage threshold the connection between the input signal and the input of the first voltage detector is restricted or disconnected.

The voltage detector system may further comprise: a third voltage detector having an input for receiving the input signal and an output for outputting a detection signal, wherein the third voltage detector is configured to output a detection signal when the voltage of the input signal meets a third rising input voltage threshold which is higher than both the first threshold and the second threshold, wherein the output of the third voltage detector controls a connection between the input signal and the input of the second voltage detector such that when the voltage of the input signal meets the third rising input voltage threshold the connection between the input signal and the input of the second voltage detector is restricted or disconnected.

The third voltage detector may be according to the first aspect of the invention.

The first voltage detector may be configured to cease outputting a detection signal when the voltage of the input signal meets a first falling input voltage threshold, wherein the first falling input voltage threshold is lower than the first rising input voltage threshold. The second voltage detector may be configured to cease outputting a detection signal when the voltage of the input signal meets a second falling input voltage threshold, wherein the second falling input voltage threshold is lower than the second rising input voltage threshold.

The third voltage detector may be configured to cease outputting a detection signal when the voltage of the input signal meets a third falling input voltage threshold, wherein the third falling input voltage threshold is lower than the third rising input voltage threshold.

The voltage detection system may further comprise an open drain output stage, the output stage comprising a plurality of MOSFET devices, each MOSFET device having: a gate terminal connected to an output of a respective one of the voltage detectors; a source terminal connected to the common reference; and a drain terminal connected to an output terminal of the output stage.

A gate threshold of each MOSFET device may be lower than the rising input threshold and the falling input threshold of the respective voltage detector to which the MOSFET device is connected.

Brief Description of the Drawings

Embodiments of the invention will now be described, strictly by way of example only, with reference to the accompanying drawings, of which:

Figure 1 is a schematic representation of a sensor system that is powered by intermittent sources;

Figure 2 illustrates the operation of a commercially available voltage detector as used in the circuit of Figure 1 ; Figure 3 illustrates the operation of a basic power on reset (POR) pulse generator circuit; Figure 4 illustrates the ideal behaviour of a POR circuit;

Figure 5 is a schematic representation of a system architecture of a system incorporating a plurality of voltage detectors;

Figure 6 is a timing diagram illustrating the operation of the system of Figure 5;

Figure 7 is a schematic representation of a voltage detector for use in the system of Figure 5;

Figure 8 is a timing diagram illustrating the operation of the voltage detector of Figure 6;

Figure 9 is a schematic representation of a circuit for implementing voltage reference circuits in the voltage detector circuit of Figure 6;

Figure 10 is a graph illustrating start-up behaviour of the circuit of Figure 6;

Figure 11 is a schematic representation of conventional low power body biasing schemes;

Figure 12 is a schematic representation of the circuit of Figure 9, with an additional body- bias generator;

Figure 13 is a schematic diagram showing a body-bias circuit; and

Figure 14 is a schematic representation of a power-gating circuit suitable for use in the architecture of Figure 5.

Description of the Embodiments Figure 5 illustrates a voltage detection system which combines all the desired properties of voltage detectors, namely high maximum input voltage, a low voltage detection threshold and low quiescent power consumption. The system, shown generally at 100 in Figure 1 , includes a power gating stage 120, a voltage detection stage 140 and an output stage 160.

The voltage detection stage 140 includes a first voltage detector 142 which is configured to detect input voltages in a high voltage range (e.g. in the range 2.8 volts to 20 volts), a second voltage detector 144 which is configured to detect input voltages in a medium voltage range (e.g. 0.6 volts to 2.8 volts) and a third voltage detector 146 which is configured to detect input voltages in a low voltage range (e.g. 0.45 volts to 0.6 volts). This arrangement of three voltage detectors ensures that the voltage detection system 100 is able to operate in a wide input voltage range, therefore facilitating the capture of energy from pulses with a wide power range.

The power gating stage 120 is configured to prevent potentially damaging input voltages from reaching the second and third voltage detectors 144, 146, and is arranged such that the first voltage detector 142 gates the second and third voltage detectors 144, 146 and the second voltage detector 144 gates the third voltage detector 146. As can be seen in Figure 5, the voltage input VEST of the system 100 is connected to an input VIN(H) of the first voltage detector 142. An output VOUT(H) of the first voltage detector 142 is connected to the gate terminal of a first P-channel MOSFET 122. The source terminal of the first P-channel MOSFET is connected to the voltage input VEST and the drain terminal of the first P-channel MOSFET 122 is connected to an input VIN(M) of the second voltage detector 144, and, via a second P-channel MOSFET 124, to an input VIN(L) of the third voltage detector 146.

When the first voltage detector 142 detects an input voltage in the high voltage range, its output VouT(H) goes high, causing the first P-channel MOSFET 122 to switch off, thereby restricting or preventing the input voltage from reaching the second voltage detector 144 or the third voltage detector 146. Similarly, the gate terminal of the second P-channel MOSFET 124 is connected to an output VouT(m) of the second voltage detector 144, such that when the second voltage detector 144 detects an input voltage in the medium voltage range, its output VOUT(M) goes high, causing the second P-channel MOSFET 124 to switch off, thereby restricting or preventing the input voltage from reaching the third voltage detector 146.

The output stage 160 of the voltage detection system 100 includes (in the illustrated example) first, second and third N-channel MOSFETS 162, 164, 166. The gate terminal of the first N-channel MOSFET 162 is connected to the output VOUT(H) of the first voltage detector 142, whilst the gate terminal of the second N-channel MOSFET 164 is connected to the output VouT(M) of the second voltage detector 144 and the output of the third N- channel MOSFET 166 is connected to the output VOUT(L) of the third voltage detector 146. The drain terminals of the first, second and third N-channel MOSFETs 162, 164, 166 are all connected to an open-drain output terminal VOD of the output stage 160, whilst the source terminals of the first, second and third N-channel MOSFETs 162, 164, 166 are all connected to ground. Accordingly, if any one of the first, second or third voltage detectors 142, 144, 146 is triggered, the open drain output VOD of the output stage 160 will be activated.

Figure 6 is a timing diagram illustrating the operation of the circuit of Figure 5. The input voltage VIN is shown as a trapezoidal pulse. As the rising edge of this pulse reaches the voltage detection threshold of the third (low voltage range) voltage detector 146, the third voltage detector 146 turns the third N-channel MOSFET 166 on, activating the system's output VOD. When the input voltage VIN reaches the voltage detection threshold of the second (medium voltage range) voltage detector 144, the second voltage detector 144 switches on the second N-channel MOSFET 164, and power-gates off the third voltage detector 146, thus protecting it from a further rise in VIN, and causing the third N-channel MOSFET 166 to be switched off. As VIN reaches the voltage detection threshold of the first (high voltage range) voltage detector 142, the first N-channel MOSFET 162 is switched on, and both the second and third voltage detectors 144, 146 are disconnected from the input (or at least the connection restricted), resulting in the turning off of the second N-channel MOSFET 164. The first voltage detector 142 remains active until the input voltage drops below its voltage detection threshold, which leads to the second voltage detector 144 being powered on. A further fall in the input voltage activates the third voltage detector 146.

The thresholds at which the voltage detectors 142, 144, 146 switch on in response to a rising input voltage may be different from (higher than) the thresholds at which the voltage detectors 142, 144, 146 switch off in response to a falling input voltage, in order to provide hysteresis and thus avoid system oscillation. The overall result is a continuous activation of the open-drain output VOD for the duration of the input pulse. To ensure seamless operation of the voltage detection system 100, the threshold voltage of each MOSFET device 162, 164, 166 should be lower than the detection thresholds at which the respective voltage detectors 142, 144, 146 switch on in response to a rising input voltage and the thresholds at which the respective voltage detectors 142, 144, 146 switch off in response to a falling input voltage.

This illustrates the reason for using an open-drain output. When activated, the voltage outputs VouT(H), VouT(M) and VOUT(L) of the voltage detectors 142, 144, 146 are at the same potential as their corresponding inputs VIN(H), VIN(M) and VIN(L). Since the voltage detector system 100 will interface to other CMOS devices with much lower maximum allowable voltages, VOUT(H) and VOUT(M) cannotbe used as the output of the overall system. The open drain output stage allows the voltage detector system 100 to output a signal that is usable by, and not damaging to, an external device.

Figure 6 also illustrates the profile of the total quiescent current IIN of the voltage detector system 100. IIN is zero when the input voltage VEST is zero, and on a rising input voltage the current increases from 42 pA at 0.2V (below the voltage detection threshold of the third voltage detector 146) to 83 pA at VEST = 1 V (above the detection threshold of the second voltage detector 144), and to around 236 pA at VEST = 5V (above the detection threshold of the first voltage detector 142), with current surges occurring at the activation points of each of the three voltage detectors 142, 144, 146. The measured total energy loss per transition caused by static current when an input voltage rises from 0 to 2.8V, a voltage slightly higher than the high detection threshold (2.8V) with two typical gradients of lV/s and lOOV/s respectively are 788pJ and 7.9pJ respectively. This is lower than prior art voltage detection circuits, which contain circuits that, for a given rail voltage, continuously consume quiescent current. An alternative version of the voltage detector system 100 in which the third voltage detector 144 is disabled draws 1.2pA at 0.2V, around 6pA at IV and l l lpA at 5V. Its total energy consumption at lV/s and lOOV/s transition from 0 to 2.8 V of the input voltage is 229pJ and 23pJ respectively.

Figure 7 is a schematic diagram illustrating a voltage detector for use in the voltage detector system 100 shown in Figure 5, to form the first, second and/or third voltage detector 142, 144, 146 used in the voltage detector system 100.

The voltage detector, shown generally at 300 in Figure 7, provides a programmable trigger threshold that is not fixed by the process technology chosen. The detection threshold and hysteresis can be set, at the time of circuit design, to a desired value. This is desirable for under-voltage-lockout circuits that activate other circuits in a specific input voltage band.

The voltage detector 300 has a trigger stage 310 which is powered by the input signal VEST. The trigger stage 310 comprises an inverter formed by a transistor pair consisting of a PMOS transistor 312 (labelled MPl in Figure 7) and an NMOS transistor 314 (labelled MN1 in Figure 7), followed by two inverter buffer stages 320, 330, which drive an open- drain output switch 340. The voltage detector 300 also includes a first ultra-low-power voltage reference circuit 350 and a second ultra-low-power voltage reference circuit 360, which generate, respectively, a fixed higher reference voltage V _re f_H and a lower voltage V _re f_L. The body terminal voltages V _re f_BB of the first and second ultra-low-power voltage reference circuits 350, 360 are shown in Figure 7 to be controlled by another voltage reference. This other voltage reference may be, for example, a further reference generator block 370 with output V _re f_BB . Alternatively, two reference generator blocks could be provided, one for each of the first and second ultra-low-power voltage reference circuits 350, 360. As a further alternative, the V _re f_BB inputs to the high and low voltage reference blocks could be provided by nodes internal to these blocks themselves (self-biased blocks). The self-biased version will now be discussed.

Depending on the rise and fall of the input voltage VEST, V _re f_H and V _re f_L are multiplexed by a switch 380 to a single reference input V _re f to the gate of the trigger stage 3 10. As shown in Figure 7, when the output signal VOUT of the voltage detector 300 is low, the switch 380 connects the output ofthe first ultra-low-power voltage reference circuit 350 (i.e. the higher reference voltage V _re f_H) to the input of the trigger stage 3 10, whilst when the output signal VOUT of the voltage detector 300 is high, the switch 380 connects the output of the second ultra-low-power voltage reference circuit 360 (i.e. the lower reference voltage Vref L) to the input of the trigger stage 310.

The two reference voltages V _re f_H and Vref L, along with the relative sizing of the PMOS and NMOS transistors 312, 314 which make up the trigger stage 3 10, determine the different rise and fall detection thresholds (Vth rise and Vthjaii) of the voltage detector 300, to provide detection hysteresis. The detection (trigger) thresholds are determined by equating the current logger through the PMOS transistor 312 and through the NMOS transistor 314, when the input VEST rises to Vth nse and falls to Vthjaii (given that the two transistors 312 and 314 operate in subthreshold and in saturation):

I trigger (rise) = K _p (π ^~ (1 )

I trigger (fall) = Kp (fl ^~ (2) where VTH _p and VTHII are respectively the gate thresholds of the PMOS transistor 3 12 and the NMOS transistor 314 ofthe trigger 310 in Figure 7. K _p = μ _ρ C _ox ^■ (Wi/Li) and K _n = μ _η Cox ^■ (W2/L2) are their gain factors (μ is mobility, C _ox is gate oxide capacitance per unit area, and Wis transistor width and L is transistor length); n is the subthreshold slope factor (n = 1 + Cd/Cox where Cd is depletion capacitance, and assuming the n difference between PMOS and NMOS is small); VT is thermal voltage (~ 26 mV). The detection hysteresis is

Vth_rise ^_ Vth fall = 2* (Vref_H - Vref_L).

From (1 ) and (2), analytical solutions for V _t h_nse and Vthjaii are

Vthj-ise = ^~ WVT ln(K _p /K„) + 2 V _re /_H (3) Vth jail = ^~ WVT ln(K _p /K„) + 2 V _ref _ _L (4)

By selecting width (W) and length (L) of the PMOS transistor 312 and the NMOS transistor 314 appropriately, the first and second terms in (3) and (4) can be set to cancel out and achieve Vth_rise = 2V _re f_H and Vth rise = 2V _re f L. Therefore, controlling the detector thresholds and hysteresis only requires the changing of the two reference voltages V _re f_H and V-ef L.

The operation of the voltage detector 300 is illustrated in Figure 8. Initially, the two reference voltages Vref H and V _re f_L and the output V _re f_BB of a third reference generator (which, as explained above, may be the further reference generator block 370) rise as VEST increases, and then settle to three different steady-state levels.

During the settling time (to), VOUT stays low so that V _re f follows Vref H via the multiplexer 380. When VEST rises to Vth rise (= 2V _re f_H) at ti, the PMOS transistor 312 is turned on, and VOUT is thus pulled up to VEST. Meanwhile, the multiplexer 380 switches V _re f from V-ef H to V ref L until VEST drops to Vthjaii Then VOUT goes low and V _re f switches back to Vref_H for the next incoming input pulse.

Figure 9 is a schematic representation of a circuit for implementing the first and second voltage reference circuits 350, 360 of Figure 8. The circuit, shown generally at 400 consists of two stacked transistors 410, 420 with different gate thresholds VTH. A first one 410 of the stacked transistors has a lower VTH and zero gate-source voltage, whilst a second one 420 of the stacked transistors has higher VTH and is configured as a diode-connected transistor. At steady-state the circuit consumes only pico-amperes of current. A shortcoming of the circuit 400 is that it is relatively slow to reach a steady-state output during a rising input voltage VEST. For fast rises of VEST, this leads to the reference voltages not having been established by the time the input voltage reaches the desired detection threshold voltage. The circuit 400 therefore triggers too soon. The reason for the long start-up time is the circuit's extremely low current that is charging parasitic capacitors.

It is possible to increase the speed with which the circuit 400 reaches a steady state output. This involves increasing the current through the low VTH transistor 410 relative to that flowing in the high VTH transistor 420 of Figure 9. The difference in the currents, the output current (shown as Ιουτ in Figure 9), determines how fast the output parasitic capacitor is charged. However, this is done by changing transistor sizes, which, in turn, changes the output parasitic capacitance (affecting the start-up) and generated reference voltage. This makes it very difficult to achieve both desired fast start-up and reference voltages at the same time.

It is possible largely to decouple the setting of reference voltages and speed-up of the reference voltage generators 350, 360, and this leads to a wider allowable range of input voltage gradients.

Returning to Figure 1 5, consider the system 300 the reference generator 370, which provides a reference output V _re f_BB, which sets the body bias of the low VTH transistor 410. The reference generator 370 speeds up the reference generation, as illustrated in the graph of Figure 1 0, thus setting up the detection threshold sooner and allowing higher input voltage gradients.

This requires a body-biasing method. Again, commonly-used bandgap reference circuits use too much power. Known lower power body bias schemes are shown in Figure 1 1. For standard CMOS process technology, the body terminal ('Β') of NMOS transistors is connected to the lowest potential in the circuits, e.g. VCOM, a common reference (which may be ground), as shown in the left-hand circuit of Figure 1 1. This results in a negative body-source voltage i.e. VBS = - V _re f, which strongly constrains the current level through the IOW-VTH transistor and hence that charging the output parasitic capacitor. Flexibility is provided in isolated CMOS process technology, which allows shorting body and source terminals, as shown in the right-hand circuit of Figure 1 1 .

Figure 12 is a schematic representation of the circuit of Figure 9, with an additional body- bias generator, whilst Figure 13 is a schematic diagram showing a body -bias circuit 500 which introduces an additional body-bias generator.

The body-bias circuit 500 of Figure 13 sets the parameter V _re f_BB to be higher than V _re f so that VBS becomes positive. This further increases the charging current and makes the current controllable, which in rum speeds up the start-up. The V _re f_BB value can be customised for different VEST gradients to ensure that V _re f has settled before VEST reaches the desired detection threshold. For instance, setting a higher V _re f_BB allows for a steeper VEST gradient.

The transistor sizes of the body-bias generator should be chosen in such a way that V _re f_BB has a steeper gradient than V _re f when VEST rises. This ensures a positive body-source voltage during the start-up.

Figure 14 is a schematic representation of a power- gating circuit 120 suitable for use in the voltage detection system 100 of Figure 5. The power-gating transistor 122 (labelled MP2 in Figures 5 and 22) should be a medium-gate-oxide (5.5 V breakdown) PMOS transistor because its source-gate voltage can be as high as 2.8 V (corresponding to the threshold of the first voltage detector 142) before its gate-drive signal goes high. By contrast, the power- gating transistor 124 (labelled MP1 in Figures 5 and 22) can be a thin-gate-oxide PMOS transistor because its source-gate voltage does not exceed 0.6 V (corresponding to the threshold of the second voltage detector 144) and its low gate-threshold voltage helps the third (low voltage range) detector 146 start to operate at a low input voltage.

Ideally, the input voltage VIN(L) of the third detector 146 will drop after the transistor 122 (MP2) is switched off for protecting this detector. However, in some thin-gate-oxide CMOS process technologies, the leakage current through the transistor 124 (MPl) (when 'OFF') can be sufficiently high so that VIN(L) can continue to rise and follow the input voltage VIN(M) of the second (medium voltage range) detector 144. This may cause overvoltage damage to the third detector 146 especially when VIN(M) has a slow voltage gradient. A simple but efficient protection solution is to add a route to ground (or a common reference) for the leakage current. In the power gating circuit 120 of Figure 14, this is implemented with a diode-connected NMOS transistor 126 (labelled MNO), which is a thick-gate-oxide (high gate-threshold) transistor. When the second voltage detector 144 output VouT(M) goes high, which switches off the transistor 124 (MPl) (zero gate-source voltage), the two transistors 124 (MPl) and 126 (MNO) are configured as the ultra-low- power two-transistor voltage reference illustrated in Figure 9. The reference circuit clamps ViN(L) to a fixed voltage while consuming pico-amperes current only, and by adjusting the W/L ratios of transistors 124 and 126 (MPl and MNO), VIN(L) can be set at a safe voltage for the third voltage detector 146. Before transistor 124 (MPl) is switched off and while the third voltage detector 146 is operating normally, transistor 126 (MNO) draws negligible current from the input VIN(L) because its gate-threshold is much higher than VIN(L) and thus this transistor operates in its deep subthreshold region.

The voltage detector system described herein combines ultra-low power consumption, low detection threshold and wide operating range. It is useful for a wide variety of applications including high- and low-side signal monitoring and power-gating, but also for low power control components such as oscillators, gate-drives, and switching devices in low-power converters. For example, due to the voltage detector system's low quiescent input current, capacitive or resistive divider circuits using 100-1000 ΜΩ resistors can be used to adjust the detection threshold, for example to operate a load only over a desired rail voltage band, commonly referred to as Under- Voltage Lockout. Similarly, high value ΜΩ pull-up resistors can be used to convert the output into a 2-level output, for example for use in ring oscillators, timers, clocks, wake-up circuits, and pulse generating circuits. This ability to use high-impedance (capacitive or resistive) peripheral components leads to control circuits that use only a few nA of current, which is important for the miniaturisation of wireless sensor nodes, wearable medical health sensors, and internet of things devices. The above embodiments are described by way of example only. Many variations are possible without departing from the invention as defined by the appended claims.