Description of Related Art Radio receivers requiring simultaneous reception of multiple radio channels require the extraction of a number of radio signals from a single wideband signal. Such receivers may include macro base stations, micro base stations, pico base stations and others.

Presently, radio channels extracted from a wideband signal using a filter band channelizer have a sampling frequency fs that is restricted to an integer multiple of the channel spacing. In many applications, such as cellular radio, it is highly desirable for the sampling frequency to be an integer multiple of the symbol rate (i. e., not an integer multiple of the channel spacing).

One solution to this problem is to feed each output of a channelizer to a digital sample rate converter to convert the sample rate of the output to an integer multiple of the symbol rate. However, having a digital sample rate converter on each output channel of a channelizer increases the cost, complexity and power consumption of a receiver. A method for directly providing a sampling frequency that is an integer multiple of the symbol rate without increasing cost, complexity and power consumption of the receiver would be highly desirable.

SUMMARY OF THE INVENTION The present invention overcomes the foregoing and other problems with a method and apparatus enabling computationally efficient channelization of a wideband signal wherein the extracted narrowband signals have a sampling frequency that is not limited to an integer multiple of the channel spacing, and have a sampling rate that is an integer multiple of the symbol rate. A wideband signal is input to the channelizer wherein a plurality of M polyphase filters having decameters on the input sides thereof extract a first sequence of signals from the received wideband signal.

The extracted first sequence of signals is processed by an M-point inverse discrete fourier transform (IDFT) to calculate the IDFT coefficients of the first sequence of signals. The calculated IDFT coefficients are modulated with a carrier signal sequence to shift the desired narrowband signals to baseband. The channelizer is valid for any combination of down sampling factors and number of narrowband channels.

BRIEF DESCRIPTION OF THE DRAWINGS For a more complete understanding of the present invention, reference is made to the following detailed description taken in conjunction with the accompanying drawings wherein: FIGURE 1 is a block diagram of a generic wideband receiver; FIGURE 2 is a functional diagram of one branch of a DFT-channelizer having an input sample rate of Fs and output sample rate of fs; FIGURE 3 illustrates a DFT-channelizer valid for any M and N having an input sample rate of f s and output sample rate of fs ; FIGURES 4a and 4b illustrates a novel implementation of the i-th polyphase branch valid for any M and N;

FIGURE 5 illustrates a modified and computationally efficient DFT-channelizer valid for any M and N having an input sample rate of F, and output sample rate of fs; and FIGURE 6 is a table illustrating the associated parameters for several values of Fs within a channelizer designed according to the present invention.

DETAILED DESCRIPTION OF THE INVENTION Referring now to the drawings, and more particularly to FIGURE 1, there is illustrated a block diagram of a generic wideband receiver. A transmitted wideband signal is received at an antenna 5. Through several stages of mixing and filtering (shown generally at 10), the signal is processed to a desired frequency band, and is then mixed down at a mixer 15 to a baseband signal x (t) with relatively wide bandwidth for input to a wideband analog- to-digital converter 20. The analog-to-digital converter 20 converts the analog baseband signal x (t) to a digital wideband signal x (n) which is processed by a digital channelizer 25 to extract various radio channels 30. The sampling rate of the A/D converter 20 is Fs Existing DFT- channelizers 25 provide a method manner for extracting every channel within the wideband signal from the digital wideband signal x (n).

Referring now to FIGURE 2, there is illustrated a functional diagram of a branch of a DFT-channelizer. ho (n) defines a real, lowpass FIR filter 105. Every other filter within a filter bank of a DFT channelizer is a modulated version of this lowpass prototype FIR filter (i. e., Hi (w) = Ho (w-M i) M O<i<M-1 the total number of narrowband channels in the wideband signal x (n)). In the branch of FIGURE 2, Hi (w) represents a bandpass filter centered on the discrete time frequency-i (where i is the branch of the filter bank), or equivalently centered on the continuous time frequency F'<BR> <BR> <BR> Si (where F equals the symbol rate of the wideband<BR> <BR> M signal x (t)). In other words there are exactly M equal

bandwidth filters in the filter bank, and these filters are centered exactly at integer multiples of the channel spacing cs m mathematically, the relationship between Fs, M and f cs is Fs cl filter bank in FIGURE 2 can be implemented using an inverse discrete Fourier transform (IDFT) and the polyphase decomposition of the lowpass prototype filter hi (n). This implementation is typically referred to as a DFT-channelizer.

Referring now to FIGURE 3, there is illustrated a block diagram of a DFT-channelizer, where the i-th branch of the DFT-channelizer implements the system of FIGURE 2.

In FIGURE 3, Ei (zk) represents the i-th polyphase component of filter response Ho (z), where Ho (w) = Ho (z), Z = eiW ; Ei (Z) is the Z-transform of ei (n); and Ho (z) is the Z- transform of ho (n).

For computational efficiency, in the implementation of the DFT-channelizer depicted in FIGURE 4, it would be highly desirable to move the decameters 6 to the left of the polyphase filters 105, so that the polyphase filters would operate at the lowest sampling rate in the system.

However, this move can only be made if M is an integer multiple of N (i. e., M = N x K; where K is same positive integer). Unfortunately, with M = N x K, the sampling rate of output channel fs is always an integer multiple of the channel spacing (i. e., fs-K x fes)-As explained before, in many communications applications, it is highly desirable to have fs which is a small integer multiple of the symbol rate which is not necessarily an integer multiple of the channel spacing. Requiring fg to be an integer multiple of the channel spacing and an integer

multiple of the symbol rate, typically forces fsto be a very large multiple of the symbol rate. For example, in a D-AMPS cellular mobile communication standard, the symbol rate is 24.3 KHZ and the channel spacing is 30KHZ.

Thus, the smallest 5that is an integer multiple of both these rates would be 2.43 MHZ, i. e. 100 times the symbol rate.

By modifying the traditional implementation of the DFT-channelizer illustrated in FIGURE 3, the decameter 6 can be moved to the left of the polyphase filters 105 for any value of M and N. A computationally efficient channelizer may be constructed this way in which the sampling frequency fs of the output is a small integer multiple of, for instance, the symbol rate.

Assuming, for illustration purposes, that each polyphase filter of FIGURE 3 has two non-zero coefficients, the i-th polyphase filter may be expressed by the equation Ei (z M) = Ei, o + z-M Ei, l (where Ei, o equals ho [i], Eil equals _ho [M+i]. However, it should be realized that this process can be generalized to polyphase filters with any number of non-zero coefficients. The key step is moving the decameters 6 to the left of the polyphase filters 105.

The method for moving the decameter 6 to the left of the polyphase filters 105 when the number of channels M is not an integer multiple of the downsampling factor N is accomplished in the following manner. The i-th branch of the polyphase filters 105 in FIGURE 3 may be expressed as shown in FIGURE 4a. FIGURE 4a illustrates that decameters 6 may be moved to the left of gains (which combined are equal to Ei (z m) in accordance with Ei (z M) = Ei, o + z-M Ei, 1), even if M is not an integer multiple of N since Eio and Eil are pure gains. This provides the structure illustrated in FIGURE 4b. By applying the identity of FIGURE 4b to each branch of the filter bank illustrated in FIGURE 3, a modified DFT- channelizer valid for any M and N as shown in FIGURE 5 is

created. The channelizer of FIGURE 5 is valid for any M and N and not only for M = N x K.

Given a desired channel spacing f and a desired sampling frequency fs of the output channels, the constraint that the bandwidth of the wideband signal F F = M x fCs combined with fs = Ns implies that the sampling rate of the A/D Converter Fs is an integer multiple of both the output sampling frequency f and channel spacing fcs. The smallest such Fs is the least common multiple of fs and fcs (denoted by LCM (fs/fcs)) Any integer multiple of LCM (fs, fcs) is also a valid symbol rate Fs. For example, in the D-AMPS standard for cellular mobile communication, fs = 24.3 kHz andf = 30 kHz, the smallest valid symbol rate F is 2.43 MHz. However, the symbol rate Fs must be large enough to avoid aliasing in sampling x (t). For a 10 MHz wide x (t), any symbol rate Fs larger than 20 MHz would avoid aliasing. The table in FIGURE 6 summarizes the parameters of the modified DFT-channelizer for several valid values of sampling rate of the A/D converter F of the wideband signal x (t) and several down sampling factors N.

The modified DFT-channelizer of FIGURE 5 receives the digitized wideband signal x (n) from the analog-to-digital converter 100 and extracts various channels using polyphase filters 105 consisting of decameters 6 and gains 11. The filtered channels are combined to yield a first signal sequence si (n). The first signal sequence si (n) is then provided to a M-point inverse discrete fourier transform (IDFT) 90 to generate a plurality of IDFT coefficients. The IDFT coefficients are modulated by a carrier signal e to provide the output channels Ci (n), where the sampling rate of Ci (n) in fs.

The M-point IDFT 90 in the modified DFT-channelizer can be computed using any known fast algorithm for computing DFT/IDFT. These algorithms include the radix-2 FFT (fast fourier transform) algorithm, the Cooley-Tukey FFT algorithm, the Wionogard prime-length FFT algorithm,

and the prime-factor FFT algorithm. Depending on the exact value of M, a particular algorithm for computation of the IDFT might be more efficient. Hence, the free parameters of the subsampled DFT-channelizer (e. g., F and M) can be chosen such that the resulting IDFT can be computed more efficiently using a particular FFT/IFFT (inverse fast fourier transform) algorithm. In other words, these parameters can be chosen to get an IDFT size that can be computed efficiently.

For example, if M is a highly composite number, the Cooley-Tukey FFT algorithm can be used to efficiently compute the resulting IDFT. On the other hand, if M is a prime number, the Winograd prime-length FFT algorithm can be used to efficiently compute the resulting IDFT.

Finally, if M has powers of four, the radix-4 FFT algorithm can be used to efficiently compute the resulting IDFT.

The architecture for a subsampled DFT-channelizer described in commonly assigned, Co-Pending Application, U. S. Serial No., (Attorney Docket No. 27951-160), entitled Wideband Channelization Using Subsampled Discrete -Fourier Transformers, which is incorporated herein by reference, can easily be used with polyphase filters of any definite length. With polyphase filters of length.

There will be vertical sections between the output of the analog-to-digital converter 100 and the M-point IDFT 90.

Each of these vertical sections is similar to the section below the dotted line in FIGURE 6.

Although a preferred embodiment of the method and apparatus of the present invention has been illustrated in the accompanying Drawings and described in the foregoing Detailed Description, it is understood that the invention is not limited to the embodiment disclosed, but is capable of numerous rearrangements, modifications, and substitutions without departing from the spirit of the invention as set forth and defined by the following claims.