The present invention relates to the wireless transmission of energy, in particular electric power from a generator to one or more spatially separated consumer loads.

Wireless transmission of electric energy is generally known in the art. Known systems for wireless energy transmission are based on the provision of coupled inductors, usually in the form of coils, wherein the alternating current provided by a generator is supplied to one of the coils and thus, in turn, induces an electric voltage in the other coil(s) which then may be used to generate the desired net power at a resistive load, i.e., a consumer. Depending on the desired use of the wireless energy transmission, the spatial separation of the generator and the consumer may vary between some centimeters up to a few meters. For example, the article of J. L. Villa et al., "Design of a high frequency inductively coupled power transfer system for electric vehicle battery charger", Applied Energy, Vol. 86, pp. 355-363, 2009, mentions various applications of ICPT systems, such as contact-less power supplies for professional tools, contact-less battery charging across large air gaps for electric vehicles, compact electronic devices, mobile phones and public transport systems. Furthermore, in A. Kurs et al., "Wireless power transfer via strongly coupled magnetic resonances", Science, Vol. 317, 6 ^th July 2007, it is specifically focused to mid-range transmission of energy and reported on the transfer of 60 watts over distances in excess of 2 meters. Since inductive coupling between a generator and a consumer strongly decreases with increasing distance, the current through the generator coil must be increased accordingly in order to maintain the energy transfer. In order to improve the operating distance and the efficiency, both the primary generator coil and the secondary consumer coil usually is expanded with a capacitor to an oscillating circuit having a resonant frequency that depends on the inductance L of coil and the capacitance C of the capacitor. Such a system usually is operated near the resonant frequency which should be identical in both oscillating circuits at the generator and the consumer side. However, when operating systems of this kind with a resonant coupling between the generator and the consumer, stability problems may occur. For example, C.-S. Wang et al. report in "Power transfer capability and bifurcation phenomena of loosely coupled inductive power transfer systems", IEEE Transactions on Industrial Electronics, Vol. 51, No. 1, February 2004 on unstable operating conditions (bifurcations) in systems having two resonant oscillating circuits. Similarly, in J. T. Boys et al., "Stability and control of inductively coupled power transfer systems", IEE Proc. Electr. Power AppL, Vol. 147, No. 1, January 2000, describes that a system having a plurality of resonant consumer circuits that are strongly coupled to a common generator circuit, becomes unstable if the number of consumers exceeds a certain critical value.

In order to overcome these problems, it is suggested in the document of A. Kurs et al. referred to above, to operate a mid-range system of two coupled coils at their parasitic resonance frequencies of about 5 to 10 MHz, rather than an operating frequency of 10 kHz to 100 kHz that is usually used in short-range. In the system described in this document, the capacitors of the oscillating circuit are formed by the parasitic capacities of the windings of the coils. Furthermore, it is described in this article that a system of two weakly coupled resonators has two eigenmodes with frequencies f ₀ - f and f ₀ + f , wherein f ₀ is the (common) resonance frequency of the individual resonators. Due to the high operating frequencies, the coils however have substantial radiation losses that cause a decrease in efficiency and lead to an unwanted exposure of the surroundings with electromagnetic radiation. It is therefore suggested in WO 2009/140506 to operate a system for mid-range energy transmission having two coupled resonators at a specific frequency that leads to a destructive interference of the electromagnetic radiation in order to minimize the radiation losses. However, there is still a need to provide an improved system for wireless transmission of electric energy, in particular, wireless transmission from one or more generators to one or more spatially separated consumers. Preferably, the system and method of the present invention should be able to transfer the energy more efficiently and/or should reduce the exposure of the surroundings with electromagnetic radiation. Further, preferably the stability of energy transmission should be increased, specifically in view of possible fluctuations in the operating parameters. The invention provides a device and method for wireless transmission of energy. The device according to the invention comprises, among other things, a resonator system comprising a plurality of resonators. The resonators are weakly coupled, i.e., the magnetic flux shared between adjacent resonators is small compared to the flux generated by each of the resonators itself. Such a system of coupled resonators has a defined number of eigenmodes with different eigenfrequencies. The device further comprises at least one generator coupled to one of the resonators, the generator being configured to operate the resonator system at least at one, and preferably at one of the plurality of eigenmodes. By operating the resonator system at an eigenmode with a specific eigenfrequency, a spatially coherent magnetic field is generated that may be used to transmit energy to an appliance. Correspondingly, the method according to the present invention comprises, among other things, the step of driving at least one generator coupled to one of the resonators in the resonator system to operate the resonator system at one of a plurality of eigenmodes.

The generator operating the resonator system may be an AC generator providing an alternating current to the resonator coupled to the generator that, in turn, is coupled to the remaining resonators of the system. Alternatively, the resonator system may make use of frequency selective feedback to allow the resonator system to autonomously oscillate at the desired eigenfrequency as an inherently stable oscillator, i.e. without requiring an external AC generator. The stable oscillator system precisely oscillates at the desired eigenfrequency, thereby safely tracking temporal changes of the eigenfrequency which may be induced by fluctuations in operating parameters, for example, coupling factors, values of the LC components, temperature etc.. In an embodiment, the oscillator system comprises an amplifier and a feedback circuitry for refeeding a part of the excitation signal to the input of the amplifier in such a way, that exactly one of the plurality of eigenmodes is excited.

Preferably, the generator excites the resonator system coupled thereto at the eigenfrequency of the one of the pluralities of eigenmodes or at a sub-harmonic thereof. If two or more generators are used, they may all work at the same (eigen)frequency or at different eigenfrequencies of the resonator system. More preferably, the resonator system operates at the eigenmode having the highest eigenfrequency. The resonators may be provided in the form of oscillating circuits, i.e., circuits comprising a capacitor, an inductor and a resistive load. Usually, the inductor is formed of a coil made of round wires, litz wires , or possibly round tubes as cylindrical or planar coils. Also coils made of metal strips may generally be advantageously used in oscillating circuits for resonator systems with coupled resonators. Coils made from metal strips may be reproducibly produced with defined eigenresonance frequencies. Accordingly, coils of metal strips may also be advantageously used in the resonator system of the device according to the present invention.

The device and method according to the present invention may be used in various applications which involve the need to provide energy to different kinds of appliances. One field of application is the provision of energy to an appliance that is situated in areas that do not allow a wired energy connection. That may be, for example, transmission of energy to sensors situated in hazardous areas or medical equipment in a sterile environment. Another example is contactless energy transmission to safety systems. Another field of application is wireless charging of storage batteries, for example storage batteries in areas exposed to explosion hazards, in electric vehicles or in mobile IT equipment, such as laptops, PDAs, printers or mobile phones. The device according to the present invention including coupled resonators may further be used for sensing and wirelessly transmitting quantities to be measured. The invention further provides a receiving device, e.g. a rectifier or any kind of AC-to-DC- converter configured to receive energy from the transmission device according to the present invention. Again, when the storage battery of an electric vehicle is to be charged, the receiving device may be coupled to the drive system of the electric vehicle, providing for a less complex structure of the electric vehicle since the driving system that is anyway present in the electric vehicle is concurrently used also for providing energy to the storage battery.

The present invention will be explained in the following with reference to the appended figures, wherein Fig. 1 schematically shows a plurality of weakly coupled oscillating circuits forming a resonator system, coupled to two generators, according to an embodiment of the present invention,

Fig. 2 schematically illustrates the eigenmodes of a system of four coupled resonators, Fig. 3 illustrates a calculation of the currents through the coils of the system shown in Fig. 2 as a function of the frequency, Fig. 4 exemplarily shows the calculated frequency responses of the real and imaginary part of the transfer functions describing the system of Fig. 2,

Fig. 5 illustrates different possible arrangements of the resonators in the resonator system, Fig. 6 schematically shows an embodiment of an inherently stable oscillator system according to an embodiment of the present invention,

Fig. 7 illustrates the effect of a frequency-selective feedback, Fig. 8 schematically shows two embodiments of the inherently stable oscillator system,

Fig. 9 illustrates a simulation model of a system according to an embodiment of the present invention including an inherently stable oscillator system coupled to four oscillating circuits, Figs. 10a to 10c schematically illustrate embodiments of the combination of a wireless power transmission receiver with the power electronics of an electric vehicle,

Figs. 11a to 11c show an inherently stable oscillator system that can be operated at any subharmonic of an eigenfrequency of the resonator system, and the corresponding voltage and current waveforms,

Fig. 12 shows a PFC power supply with an output stage and transmitting resonator according to a further embodiment of the invention, and Fig. 13 illustrates a method for limiting voltage and current in the resonant elements of the resonator system according to an embodiment of the invention. The present invention is based in part on the idea to generate a quasistatical electromagnetic field that may be used for transmitting energy using a group of resonators that are weakly coupled among each other and thus form a resonator system having a plurality of eigenmodes. Specifically, an arrangement of N spatially separated but weakly coupled resonators form a resonator system having N eigenmodes. The physical properties of these eigenmodes may advantageously be used in a device for wireless transmission of energy.

Fig. 1 schematically illustrates a device for wireless transmission comprising N spatially separated resonators (N=14 in the example of Fig. 1) in the form of LC oscillating circuits that preferably have a high Q factor (Q _n = ^ _n /C _n / R » 1). S of the resonators (S=2 in the example of Fig. 1) are respectively coupled to AC current generators. The remaining resonators

M=N-S include a resistive load R. The individual resonance frequency /„ = of each

2^L _n C„ one of the resonators, i.e., the resonance frequency of the non-coupled resonators, preferably is identical for all N resonators. The values of the inductance Lj, and the capacitance C _n for each resonator may be different, however.

The inductive coupling among the resonators depends, among other things, on their spatial distance. Specifically, the distance between the resonators should be chosen such that a weak coupling is achieved, that is, the coupling factor k _n>m should be less than one: k _nm = M _nm /-JL _D L _m « 1 . In this formula, is the mutual inductance between the n ^th and the m ^th resonator, Ln and L _m are the inductances of the resonators, respectively. In order to ensure a weak coupling, knm should be less than 0.5, and more preferably less than 0.1.

The mutual coupling of the N resonators in the system shown in Fig. 1 causes the system to have N well-defined eigenmodes. The N eigenmodes of the system each represent a stable temporal and spatial oscillation, wherein all N resonators oscillate synchronously at a specific eigenfrequency at a fixed mutual phasing. Due to a fixed temporal phase relation of the coil currents in each of the N resonators, a spatially coherent magnetic field is formed in the surrounding of the resonator system by the superposition of the three-dimensional field distribution of each single coil in each resonator. A specific example of four coupled resonators is shown in Fig. 2. The four resonators are arranged at positions z _ls z , z ₃ and z ₄ and each comprises identical components L, C, R arranged in a series. The inductive coupling factor k _n,m = k between the n ^th and the m ^th resonator circuits depends on the distance between the circuits and is small, i.e., k«l. Each resonator circuit has the (identical) resonance frequency f _n =

Fig. 2 additionally shows a diagram of the coil currents in each one of the resonator circuits and the resulting magnetic field for the four possible eigenmodes as a function of the position (in one dimension only). The fundamental mode (mode 1) has the eigenfrequency f _x = f ₀ - 3Af . In this mode, all coil currents are in phase so that the magnetic fields of all individual resonators positively add up to the resultant magnetic field. In mode 2 having the eigenfrequency f ₂ — f _g — _y , two adjacent resonators oscillate in phase, whereas the two pairs oscillate out of phase, so that the magnetic field distribution has a root. Accordingly, mode 3 leads to a magnetic field distribution having two roots. Finally, in mode 4, all resonators oscillate out-of-phase, so that the resulting field distribution shows three roots, i.e., three phase reversals. As it is obvious from the field distribution illustrated in Fig. 2, the highest order mode 4 is the mode with the lowest magnetic energy density stored between the resonators. Even though Fig. 2 only shows the field distribution in one dimension, the same is true for the three-dimensional representation of the magnetic field distribution and the resultant energy density. Accordingly, as can be seen from Fig. 2, the highest order mode is least prone to interact with other magnetically sensitive objects nearby.

In Fig. 3, the calculated coil currents for the resonator system shown in Fig. 2 are plotted as a function of the frequency. In the simulation of Fig. 3, the resonance frequency of the individual resonators is f ₀ =100 kHz, the coupling factor is k=0,035. As can be seen from Fig. 3, the four coil currents all have maxima at the eigenfrequencies fi to f ₄ of the resonator system. The mode- specific distribution of the amplitudes and phases can be seen at the different eigenfrequencies f _n , the highest mode 4 has a frequency f ₄ close to 103 kHz. As indicated above, it is preferred to operate the system comprising N coupled resonators exactly at the frequency f ^* N of the highest order eigenmode since in this mode, the energy density in the volume of the resonator system is as low as possible. Accordingly, unwanted electromagnetic interaction may be avoided and the system works as efficient as possible. However, it is equally possible to operate the system at a different eigenmode by applying a signal to the system at the respective eigenfrequency. Also, it is possible to additionally apply different frequencies to the system if more than one generator is used, as shown, for example, in Fig. 1. The operating frequency of the system may be chosen by adapting the dimensions of the resonators. It is preferred to operate the system at a low frequency in order to avoid radiation losses caused by electromagnetic waves. Therefore, the typical dimension of a resonator preferably is smaller as 1/20 of the wavelength l ₀ - cl f _Q . The inductive coils are an important component in the individual resonators of the resonator system used according to the present invention. Specifically, it is important to be able to provide coils having well defined characteristics, in particular a well defined inductance. While resonator coils usually are made of round wires, litz wires, or round tubes in the form of cylindrical or planar coils, it may be advantageous to use coils made of metal strips in the resonators of the device according to the present invention. The metal strip should be arranged in parallel to the coil axis and thus parallel to the magnetic field of the coil. By selecting a suitable thickness and suitable dielectric properties of the insulation material between the metal strips forming the coil, the distance between the individual layers of the metal strip can be controlled and thus, the capacity of the individual windings and the resulting losses due to the proximity effect may be defined. Thus, the resonance frequency of the coil can be controlled when manufacturing the coil. Furthermore, the use of a metal strip instead of a round wire reduces the skin effect on the windings which leads to an improved quality, i.e., Q factor, of the coil compared to a coil made of a round wire having equal weight. Furthermore, when using a metal strip as a conductive material of the coil, the ratio of the surface to the cross section can be increased, allowing higher current densities, and thus improved thermal characteristics of the coil.

In order to be able to efficiently transfer power from the generator with a real internal resistance RG coupled to one or more consumers, such as the resonators as shown in Fig. 2, each one of the consumers must fulfill the known complex matching conditions. That means that the imaginary part of the transfer function from the generator to each of the consumers must be zero (reactive- power compensation) and, vice versa, the real parts of all consumers have to be equal to the resistances seen towards the generator. Fig. 4 shows the real and imaginary parts of the transfer functions of the consumers, i.e., resonators, in a system as schematically shown in Fig. 2, wherein the first resonator is coupled to an AC current generator. That is, the exemplary system includes one generator circuit (S=l) and three consumer circuits (N=3) and thus has N=M+S=4 eigenmodes. In case of the serial resonator circuits shown in Fig. 2, the transfer functions from the generator to each one of the consumers, that describe the system, are given by the complex transadmittances Υ _η (ω) - I _Ln I U^ , wherein Y„(co) is the admittance seen by the generator towards the n ^th resonator , as a function of the angular frequency ω, _n is the current through the n resonator coil and Uj is the voltage at the output of the generator. Specifically, Yi(co) is the input admittance of the complete resonator system as seen by the generator .

As can be seen from the calculated frequency responses of the real and imaginary part of the four above-noted transmittances shown in Fig. 4, the imaginary part of all transmittances is zero and the real part has a mode specific value R _n at the four eigenfrequencies of the system. That is, at these frequencies, an optimal energy transfer from the generator to the consumers is ensured, as determined by the coupling conditions. Adapting a load resistor having a certain value to a generator may be done using an impedance converter stage that transfers the load resistance to the resistance R _n needed in the system, for example a transformer having the necessary winding ratio or an electronic impedance converter. Accordingly, operating the system at one of the eigenfrequencies of the system ensures simultaneous power matching of all consumers and thus maximizes the transfer efficiency of the system. Further, the real input admittance enables the so called "zero-current switching" operation of the generator, which is a prerequisite for a high DC- to-AC efficiency of the power source. As a consequence, operating the power transfer system according to the present invention also optimizes its overall efficiency, which is the product of the generator efficiency and the transfer efficiency.

If the resonators are formed by standard LC circuits, various arrangements of the resonator circuits are possible to achieve coupling between the circuits. For example, a linear, circular or cubic arrangement may be chosen. It is noted in this context that not only the distance between the circuits controls the mutual coupling of the resonators. The coupling may also be chosen by arranging the coil axis of two adjacent circuits at a certain angle, wherein orthogonal coils are not coupled at all. The components of the LC circuits, i.e., the inductance L, the capacitance C and the resistor R, may be arranged in series or parallel. When using a series arrangement, it is important for the stability of the power transmission system that the generator possesses the characteristics of a voltage source having a low impedance RQ. This ensures that the current amplitude becomes maximum in case of a resonance and turns to zero in case of strong deviations from the resonance frequency (open circuit stability). When using a parallel oscillating circuit, the suitable generator is a current source with a high impedance RG, SO that the voltage is maximum at the resonance frequency and tends to zero at high and low frequencies (short circuit stability). Both generator types may be transformed into the other one using an impedance converter stage, as discussed above. In order to operate the resonator arrangement at one of the eigenfrequencies as described above, the generator should be tuned to the desired eigenfrequency as precisely as possible. Under practical operating conditions, however, a temporal fluctuation of the eigenfrequency may be caused by load changes or fluctuations of the characteristics of the system, e.g., coupling factors, values of the LC components, temperature etc. According to an embodiment of the invention, the at least one generator comprises a frequency selective feedback path adapted to select any one of the eigenfrequencies. The feedback path may comprise a power amplifier, a branch circuit, an isolation amplifier and an electrically tunable band-pass filter. The power amplifier included in the feedback path should be adapted for zero- voltage and/or zero-power switching. To precisely control the frequency supplied by the generator, an inherently resonant oscillator system such as shown in Fig. 6 may be used. The oscillator system according to Fig. 6 comprises the following components: a power amplifier with a high efficiency (for example, a class C- amplifier, class D-amplifier, class E-amplifier, or bridge amplifiers with e. g. insulated gate bipolar transistors IGBTs or MOSFETs) including the necessary driver electronics, a multi- resonant, complex load impedance or admittance, respectively, formed by the input impedance/admittance of a coupled resonator system as shown, for example, in Fig. 1, a branch circuitry that outcouples a small portion of the output signal A (current I _L or voltage UL, respectively) to a feedback path, an isolation amplifier with frequency independent gain and high reverse isolation, and a band-pass filter having an electronically tunable center frequency. The positive feedback needed for actuating and driving the oscillation is achieved by refeeding a small portion of the output signal to the input of the amplifier in phase. The small feedback signal first is amplified by the isolation amplifier in a frequency-independent way. The amplified feedback signal is then fed to an electrically tunable narrow-band filter. Thereby, the isolation amplifier causes a decoupling of the band-pass filter from the multi-resonant load. Thus, the occurrence of unwanted additional eigenfrequencies is avoided. The output signal of the bandpass filter that has been amplified in a frequency-selective way, then forms the input signal for the power amplifier. By means of this frequency-selective feedback mechanism (FSFB = frequency-selective feedback), it is achived that the oscillator stably oscillates at the desired eigenfrequency f„ of the resonator system that is chosen by the control voltage U _contr oi. The amplitude-dependent gain G(A) of the power amplifier thereby ensures that the oscillation amplitude finally sets to a stable value. The effect of the frequency-selective feedback is illustrated in Fig. 7: the possible relevant frequencies of an oscillator are determined by the maxima, i.e., resonances, of the gain of the open feedback loop. Since the loop gain G (CO) is proportional to the complex load impedance or load admittance, by suitably choosing the feedback mode (current feedback or voltage feedback), the loop gain has the same resonances as the multi-resonant load. Fig. 7a shows the situation without frequency-selective feedback: after a transient oscillation at multiple frequencies, the oscillator will operate at that eigenfrequency f _n having the highest loop gain (modal gain). Since, however, the adjacent eigenfrequencies may have similar gains, depending on the coupling arrangement, there is a risk of mode hopping, i.e., frequent change of the eigenfrequencies due to load changes and variations of the coupling factors. This effect is very efficiently prevented by substantially increasing the gain of one of the eigenfrequencies using the electronically tunable narrow-band filter shown in Fig. 6. This mode-selective effect of the frequency-selective feedback is illustrated in Fig. 7b. The bandwidth of the band-pass filter must be smaller than the distance between the resonances (that is typically 100 Hz to some kHz). A further advantage of the frequency-selective feedback is that no dedicated feedback path is needed for stabilizing the system on the consumer side to which energy is to be transferred using the system.

It turns out that not every combination of amplifier type, complex load and feedback type leads to a stable oscillation of the eigenfrequency. Therefore, it is important to carefully choose the arrangements of the intrinsically stable oscillator system for the energy transfer system using frequency-selective feedback. One possible arrangement which ensures, that the oscillator system with a multi-resonant load is stable against load changes and frequency fluctuations, is shown in Fig. 8a: A voltage amplifier - i.e. an amplifier whose output has the characteristic of a voltage source - works with series resonances and is fed back with a signal that is proportional to the load current (open circuit stability).. Alternatively, in the arrangement shown in Fig. 8b, a current amplifier -i.e. an amplifier whose output has the characteristic of a current source- works on a load impedance with parallel resonances and is fed back with a signal that is proportional to the load voltage (short circuit stability). It should be noted, that a voltage amplifier can be realized by a class D-amplifier, whereas a current amplifier is obtained using a class E-amplifier.

In Fig. 9, a simulation model is shown that was used to simulate a fourth order system with frequency-selective feedback (S=l , M=3, N=M+S=4), including transient oscillation, mode selection and stability against parameter fluctuations.

Because of the band-pass filter characteristic of the resonant load it is not necessary that the output waveform of the generator is sinusoidal. It is possible to operate e. g. a class D-amplifier with a switching frequency equal to the desired eigenfrequency, thus generating a rectangular output voltage. In this case the generator is operated under a zero current switching condition with minimized switching losses (see fig. 1 lb).

Rather than operating the generator at exactly one of the eigenfrequencies of the system, it may be advantageous to operate the generator at a frequency that corresponds to a subharmonic of the eigenfrequency of the resonator system. This may be done, for example, with the arrangement shown in Fig. 11a: here, the frequency f„ of the feedback signal C ^' IL is divided by a factor F = 2,3,4,..., so that the oscillator operates at a subharmonic frequency f _su b = f _n /F. The shape of the oscillation generated by the oscillator should be optimized so, that the desired harmonic is present in the output signal in a well pronounced manner as can be seen in fig 11c. By exciting the resonator system with a subharmonic of one of its eigenfrequencies, switching losses in the oscillator may be reduced which allows the use of cheaper semiconductors and reduces the necessary cooling. It is also possible, to use semiconductors devices with higher current or voltage ratings, since the permissible switching frequency of high power devices is generally lower than that of low power devices. Furthermore, given a specific maximum switching frequency f _SW itch ⁼ f _S ub, the resonator circuits can be operated at a higher harmonic frequency f _n = F ^' f _su b, where the quality factors of the resonators have their maximum values. Due to the operation of the system of the present invention at one of its resonance frequencies it is necessary to control the transmitted power and to protect the components of the resonant circuits against excessive voltage at approximately open circuit operation of a parallel resonant circuit or approximate short circuit of a series resonant circuit. This is done by controlling the voltage at the input of the resonator coupled to the generator. Furthermore, the transistors must switch at zero crossing of voltage or current (zero voltage switching ZVS or zero current switching ZCS) in order to achieve low switching losses. Following statutory regulations according to the European standards, circuits that are connected to the mains, must not exceed the prescribed limits for harmonic currents. Above a certain power the use of Power-Factor-Correction (PFC) power supplies is mandatory. Such PFC power supplies typically have output voltages that are higher than the peak value of the mains voltage. For controlling the transmitted power, also the voltages that are smaller than the peak value of mains voltage must be adjustable.

The simplest way for controlling the power transmission would be to waive the condition of zero voltage or current switching and either to operate the generator at a frequency different from the resonance frequency or to switch the transistors with a changed duty cycle. However, both possibilities lead to increased switching losses in the transistors.

Another way to control the power transmission would be to use a DC/DC converter that is inserted between the PFC power supply and the output stage. Its task is to lower the output voltage of the PFC power supply to the value required for the transmission. This would, however, create additional efforts for the DC/DC converter.

According to a further aspect of the present invention it is possible to control the power transmission without needing an additional DC/DC converter inserted between the PFC power supply and the output stage of the generator. This maintains the benefits of zero current or voltage switching as well as the operation at one of the resonance frequencies of the system. According to this aspect of the invention a setting of the voltage is initially done in wide steps by selecting an appropriate subharmonic and applying it to the generator. The fine adjustment of the voltage is done by setting an appropriate target value on the output voltage of the PFC power supply.

Figure 12 shows a possible implementation of this additional aspect of the invention. The coil LI, the transistor Ml and the diode D5 form a PFC power supply, together with the rectifier diodes D1-D4. The optional elements resistor 1 (Rl) and the relay contact SI can be used to limit the charge current of the capacitor CI. The output stage is formed by the transistors M2 and M3. The resonance circuit coupled to the generator consists of the coil L2 and the capacitor C2. Other switching devices may be used instead of the illustrated MOSFETs such as IGBTs for example.

The following table shows the adjustable amplitudes of the fundamental frequency at the input of the resonance circuit at various subharmonics depending on the voltage V_DC at capacitor Cl.V_DC_min is the lowest DC voltage adjustable. Theoretically, for the maximum voltage V DC max also higher values may be chosen. However, it is advantageous to take into account the allowable voltages of the transistors. At the subharmonic selected, V_DC_min results in V_out_min and accordingly V_DC_max results in V out max. Subharmonic 2 in the table means that only every second positive half cycle is generated and the input to the resonators, for subharmonic 3, correspondingly, every third positive half cycle is generated, etc.

To control the output stage of the generator, feedback signals are necessary to capture the phasing of voltage and current in the resonant circuit as well as the amplitude at the resonant components. As a feedback signal for the phasing of the current, the voltage signal of a measuring coil or a tapping of the coil that is connected to the generator may be used. Furthermore the coil current may be measured and evaluated by a current transformer. It is also possible to monitor the voltage at the resonant capacitor. If several capacitors are connected in series, it is sufficient to detect the voltage at one of these capacitors. In order to synchronize the output stage with the resonator it is necessary to shift the zero crossings of the voltage at these reactive components by 90° in the control unit. This results in a phase shift of 0° between the voltage and the current in the resonant circuit. During a measurement of the current, the 90° shift is not needed. Zero crossings can be detected by using comparators.

If delay times in the signal transmission between the controller and switching transistors are to be compensated, the actual switching times of the output stage transistors can be monitored by a comparator and provided to the controller as an additional signal. In case of a short circuit of a receiver that is operated as a series resonator or in the absence of a load at a receiver that is operated as a parallel circuit, and when the transmitter is operated without a receiver being present, very high voltages of several kilovolts at the resonant elements can occur. The control unit of the transmitter must therefore monitor the voltage at the resonant elements. If the voltage exceeds a limit, the fundamental frequency of the input voltage of the transmission resonant circuit must be reduced by choosing an appropriate subharmonic and/or by reducing V DC.

The limits for voltage and current of the resonant elements of the transmitter are monitored with the help of a window comparator which uses the amplitude of the feedback signal. These limits are sent to the control unit as a digital signal (see Fig 13). The sampling of the threshold information is done at a 90° shift with respect to the zero crossings of the feedback signal, i.e. when the voltage and current are at a maximum. If the predetermined threshold is exceeded or underrun, the control unit selects a suitable subharmonic in conjunction with an appropriate voltage target value for the PFC power supply. The load applied to the resonant components thus stays within the limits. Thereby, voltage and current may not necessarily be measured separately as analog signals, but it is possible to digitally derive all necessary information from the feedback signals via the comparators. In order to be able to load the storage battery of an electric vehicle and generally loading storage batteries of all kinds of appliances, it is necessary to transfer the alternating-current (AC) voltage provided by the energy transmission system to a direct-current (DC) voltage. However, electric vehicles include power electronics that transform the DC voltage of the battery into AC voltage for the motor when driving, and vice versa when recuperating AC voltage of the motor to charge the battery. These power electronics of electric vehicles may thus be used to re-charge the battery when loading the battery with the wireless transfer unit according to the present invention. By using the existing electronic, an additional circuitry for adjusting the receiver side of the wireless power transmission system to the requirements of the battery charge may be avoided.

In accordance with an embodiment of the present invention, a receiver for the wireless energy transfer system may be incorporated in the driving system of the electric vehicle in two ways. The inverter of the driving system may comprise switching transistors Tr and free-wheeling diodes D. Fig. 10a shows the use of the free-wheeling diodes of the inverter as diodes for rectifying the AC voltage from the wireless energy transmission system. Thereby, the amplitude of the AC voltage must be adjusted so that optimal charging of the battery may be achieved. Accordingly, details of the required voltage to charge the battery must be made available to the energy transmission system. During charge, switch SI is closed and switch S2 is opened in order to separate the motor, which is indicated by windings LI, L2 and L3, from the inverter.

In the second embodiment shown in Fig. 10b, the inverter, consisting of transistors Tr and freewheeling diodes D, is used as a boost-converter while loading the battery. The output voltage of the wireless energy transmission system is rectified using a rectifier. In addition to the rectifier, a filter capacitor may be used. When loading the battery, switch SI is closed. The negative terminal of the wireless power transmission system is connected to the negative terminal of the battery, the positive terminal is connected via switch S 1 to the neutral point of the motor, which again is illustrated by windings LI to L3. The inverter can now be operated as a boost converter, the inductance of the motor windings being used as storage chokes. The boost converter sets the rectified output voltage of the wireless energy transmitting system to a value that is adapted to the level of the battery voltage. As three half-bridges are available for the boost converter, it is also possible to operate the boost converter as a multi-phase boost converter. In this way, the power loss at all half-bridges of the inverter is evenly distributed and ripples of the voltage at the filter capacitor on the output side of the rectifier of the wireless energy transmitting system is reduced. If the neutral point of the engine is not accessible, the arrangement shown in Fig. 10c may be used. According to Fig. 10c, an additional storage choke L is inserted and the wireless energy transmitting system is connected via switch SI with the motor supply lines. By synchronizing one or all of the lower transistors of the inverter, the inverter works as a boost converter which sets the wireless energy transmitting system to a desired value.

By combining the driving system and the circuitry for coupling to the wireless energy transmitting system, the packaging in the vehicle is made simpler since the system needs less space. Furthermore, costs may be saved and the reliability may be improved since less components are needed. For back-fitting the wireless energy transmitting system to existing driving systems, the arrangement shown in Fig. 10c is particularly suitable.

An additional application of the present invention is the use of the claimed device as a sensor and for wireless transmission of the metered values. Various systems for wirelessly transmitting quantities to be measured are known, including radiocommunication or RFID systems.

When studying systems including coupled resonators for wireless energy transmission, it has been found that these devices are very sensitive to minute changes of several parameters, like the resonance frequency, the quality factor and the coupling between transmitter and receiver. The sensitivity of the device of the present invention may be used to gain information on the quantity to be measured. Specifically, a change of the resonance frequency of the receiver, the quality factor of the receiver circuit and/or the coupling between the receiver and the transmitter can be detected on the transmitter side as changes of the resonance frequency, the input power or a phase shift between voltage and current of the transmitter circuit. Thereby, information on the quantity to be measured can be obtained.

As an example, parameters such as humidity, temperature or a deformation of the capacitors can lead to a change in the capacitance. This leads to a change of the resonance frequency of the receiver which can be detected by measuring the resonance frequency of the transmitter circuit. Other parameters that can be detected using a change of the resonance frequency of the receiver circuit are for example the deformation of the coils or the change of the relative permeability through a ferromagnetic material which results in a change of the inductivity. In case of a change of the temperature, humidity or strain the quality factor varies. Furthermore, a change in the distance of the transmitter and the receiver or in their spatial orientation can be detected by determining a change of the coupling factor. A combination of several systems is also possible, for example to build a wireless resolver for measuring an angle of rotation, which does not include ferromagnetic materials. When using the system according to the present invention, various physical quantities can be measured and transmitted wirelessly and contactlessly. Since no ferromagnetic materials are needed in these systems, sensors can be realized with low mass and at low costs. The sensors are not influenced by magnetic or electric fields due to the frequency selective method of operation. With the present invention, wireless power transmission to a large number of consumers in a three-dimensional space may be achieved. Furthermore, the overall efficiency is improved by operating the system at one of its eigenfrequencies. When using the eigenmode with the highest eigenfrequency, the least possible amount of electromagnetic energy is stored in the environment. By means of frequency-selective feedback to a power amplifier, a self-stabilizing oscillator circuit precisely operating at an user-selectable eigenfrequency is realized.