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Title:
INBAND RIPPLE COMPENSATION
Document Type and Number:
WIPO Patent Application WO/2019/145042
Kind Code:
A1
Abstract:
It is provided a method, comprising weighting each of plural input signals with a respective weight; performing an inverse Fourier transform on the plural weighted input signals to obtain a complex intermediate signal; filtering the complex intermediate signal by a filter function having a passband in frequency domain to obtain an output signal, wherein a frequency response of the filter function in the passband has a ripple relative to an ideal frequency response in the passband; each of the weights is an inverse of a respective frequency response with respect to the ideal frequency response in the passband.

Inventors:
WOLFF GUNTER (DE)
KRATZERT ANDREAS (DE)
NADER CHARLES (DE)
Application Number:
EP2018/051946
Publication Date:
August 01, 2019
Filing Date:
January 26, 2018
Export Citation:
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Assignee:
NOKIA SOLUTIONS & NETWORKS OY (FI)
International Classes:
H04B1/12; H04L25/03; H04L27/26; H04L27/36
Domestic Patent References:
WO2014124930A12014-08-21
Foreign References:
EP2159979A12010-03-03
Other References:
Y. C. LIM; Y. LIAN: "The optimal design of one- and two- dimensional FIR filters using the frequency response masking technique", IEEE TRANS. CIRCUITS SYST. II, vol. 40, February 1993 (1993-02-01), pages 88 - 95, XP000322105, DOI: doi:10.1109/82.219838
3GPP TS36.104 V15.0.0, Retrieved from the Internet
3GPP TS38.104 V15.0.0, Retrieved from the Internet
3GPP TS38.211 V15.0.0, Retrieved from the Internet
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Claims:
Claims:

1 . Apparatus, comprising

weighting unit configured to weight each of plural input signals with a respective weight;

transformer configured to perform an inverse Fourier transform on the plural weighted input signals to obtain a complex intermediate signal;

channel filter configured to filter the complex intermediate signal by a filter function having a passband in frequency domain to obtain an output signal, wherein a frequency response of the filter function in the passband has a ripple relative to an ideal frequency response in the passband;

each of the weights is an inverse of a respective frequency response with respect to the ideal frequency response in the passband. 2. The apparatus according to claim 1 , wherein

each of the input signals is a subcarrier;

at least a subset of the subcarriers belongs to a numerology;

each of the subcarriers of the numerology is orthogonal to each of the respective other subcarriers of the numerology.

3. The apparatus according to any of claims 1 to 2, further comprising

determining unit configured to determine whether or not an input signal of the input signals is used;

inhibitor configured to inhibit weighting the input signal if the input signal is not used.

4. Apparatus, comprising

channel filter configured to filter a received signal by a filter function having a passband in frequency domain to obtain an intermediate complex signal;

transformer configured to perform a Fourier transform on the complex intermediate signal to obtain plural output signals having respective different frequencies;

weighting unit configured to weight each of plural output signals with a respective weight; a frequency response of the filter function in the passband has a ripple relative to an ideal frequency response in the passband;

each of the weights is an inverse of a respective frequency response at the respective frequency with respect to the ideal frequency response in the passband.

5. The apparatus according to claim 4, wherein

each of the output signals is a subcarrier;

at least a subset of the subcarriers belongs to a numerology;

each of the subcarriers of the numerology is orthogonal to each of the respective other subcarriers of the numerology.

6. The apparatus according to any of claims 4 to 5, further comprising

determining unit configured to determine whether or not an output signal of the output signals is used;

inhibitor configured to inhibit weighting the output signal if the output signal is not used.

7. The apparatus according to any of the preceding claims, wherein

the frequency response in the passband is normalized with the average frequency response in the passband being set to 1 ; and

the inverse of the respective frequency response is calculated as a reciprocal of the respective normalized frequency response multiplied by a positive weighting factor.

8. The apparatus according to claim 7, wherein the weighting factor is predefined.

9. The apparatus according to claim 8, further comprising,

if depending on any of claim 1 to 3, a monitor configured to monitor an error vector magnitude of the output signal;

if depending on any of claims 4 to 6, a monitor configured to monitor a throughput of the output signals; and

adapting means configured to adapt the weighting factor depending on a result of the monitoring.

10. The apparatus according to any of the preceding claims, further comprising a limiter configured to at least one of • limit each of the weights such that each of the weights is not larger than a predefined maximum value; and

• limit each of the weights such that each of the weights is not smaller than a predefined minimum value.

1 1 . The apparatus according to any of the preceding claims, wherein the ideal frequency response in the passband is an average frequency response in the passband.

12. The apparatus according to any of the preceding claims, further comprising

response transforming means adapted to Fourier transform the filter function to obtain a Fourier transformed filter function;

determining means adapted to determine a respective deviation of the Fourier transformed filter function from the ideal filter function for each of the different frequencies;

inverting means adapted to invert, for each of the different frequencies, the deviation in order to obtain the respective weight;

setting means adapted to set the obtained weights in the weighting unit.

13. Method, comprising

weighting each of plural input signals with a respective weight;

performing an inverse Fourier transform on the plural weighted input signals to obtain a complex intermediate signal;

filtering the complex intermediate signal by a filter function having a passband in frequency domain to obtain an output signal, wherein

a frequency response of the filter function in the passband has a ripple relative to an ideal frequency response in the passband;

each of the weights is an inverse of a respective frequency response with respect to the ideal frequency response in the passband.

14. The method according to claim 13, wherein

each of the input signals is a subcarrier;

at least a subset of the subcarriers belongs to a numerology;

each of the subcarriers of the numerology is orthogonal to each of the respective other subcarriers of the numerology.

15. The method according to any of claims 13 to 14, further comprising

determining whether or not an input signal of the input signals is used;

inhibiting the weighting of the input signal if the input signal is not used.

16. Method, comprising

filtering a received signal by a filter function having a passband in frequency domain to obtain an intermediate complex signal;

performing a Fourier transform on the complex intermediate signal to obtain plural output signals having respective different frequencies;

weighting each of plural output signals with a respective weight;

a frequency response of the filter function in the passband has a ripple relative to an ideal frequency response in the passband;

each of the weights is an inverse of a respective frequency response at the respective frequency with respect to the ideal frequency response in the passband.

17. The method according to claim 16, wherein

each of the output signals is a subcarrier;

at least a subset of the subcarriers belongs to a numerology;

each of the subcarriers of the numerology is orthogonal to each of the respective other subcarriers of the numerology.

18. The method according to any of claims 16 to 17, further comprising

determining whether or not an output signal of the output signals is used;

inhibiting the weighting of the output signal if the output signal is not used.

19. The method according to any of claims 13 to 18, wherein

the frequency response in the passband is normalized with the average frequency response in the passband being set to 1 ; and

the inverse of the respective frequency response is calculated as a reciprocal of the respective normalized frequency response multiplied by a positive weighting factor.

20. The method according to claim 19, wherein the weighting factor is predefined.

21 . The method according to claim 20, further comprising, if depending on any of claims 13 to 15, a monitor configured to monitor an error vector magnitude of the output signal;

if depending on any of claims 16 to 18, a monitor configured to monitor a throughput of the output signals; and

adapting the weighting factor depending on a result of the monitoring.

22. The method according to any of claims 13 to 21 , further comprising at least one of

• limiting each of the weights such that each of the weights is not larger than a predefined maximum value; and

• limiting each of the weights such that each of the weights is not smaller than a predefined minimum value.

23. The method according to any of claims 13 to 22, wherein the ideal frequency response in the passband is an average frequency response in the passband.

24. The method according to any of claims 13 to 23, further comprising

Fourier transforming the filter function to obtain a Fourier transformed filter function;

determining a respective deviation of the Fourier transformed filter function from the ideal filter function for each of the different frequencies;

inverting, for each of the different frequencies, the deviation in order to obtain the respective weight;

setting the obtained weights in the weighting unit.

25. A computer program product comprising a set of instructions which, when executed on an apparatus, is configured to cause the apparatus to carry out the method according to any of claims 13 to 24.

26. The computer program product according to claim 25, embodied as a computer-readable medium or directly loadable into a computer.

Description:
Inband ripple compensation

Field of the invention

The present invention relates to an apparatus, a method, and a computer program product related to channel filtering in a transmitter and a receiver, respectively.

Abbreviations

3GPP 3 rd Generation Partnership Project

4G / 5G 4 th / 5 th Generation

ASIC Application-specific Integrated Circuit

CP Cyclic Prefix

DFT Discrete Fourier Transform

DL Downlink

DSP Digital Signal Processor

EVM Error Vector Magnitude

FDM Frequency Division Multiplex

FFT Fast Fourier Transform

FIR Finite Impulse Response

FPGA Field Programmable Gate Array

FRM Frequency Response Masking

HW Flardware

iFFT inverse Fast Fourier Transform

LTE Long Term Evolution

MIMO Multiple Input - Multiple Output

mMIMO massive MIMO

NR New Radio

OFDM Orthogonal Frequency Division Multiplex

QAM Quadrature Amplitude Modulation

RF Radio Frequency

RX Receive

SC-FDM Single Channel FDM

TRX Transmit

TS Technical Specification TX Transmit

UL Uplink

Background of the invention

High spectrum usage is an upcoming and important requirement for 5G and LTE (narrowed carrier) allowing efficient usage of the channel bandwidth. Since the guard band between adjacent channels is shrinking, channel filters will have to become steeper and hence require a much longer filter impulse response which is challenging and costly for implementation in HW (FPGA, ASIC, DSPs, vector processors).

The longer filter impulse response has two main problems:

1 . Increased computational effort, which results in larger complexity and higher power consumption.

2. Longer delay, i.e. higher latency of the signal.

There are probably several options to reduce computational effort, but none of these options reduces the latency as well. One example is the FRM approach, shown in Fig. 1 and described in [1 ] Simplified frequency responses inside an exemplary FRM filter according to Fig. 1 are shown in Fig. 2.

As shown in Fig. 1 , an FRM channel filter consists of a delay element z Nt and three programmable filters H, Ha and Hb. The filter H has only one non-zero tap every N taps. Filter H can be seen as an oversampled filter response with an originally low sample rate. The frequency response consists of N repeated parts (aliases), each of which may contain steep slopes.

The original idea of utilizing this FRM approach is illustrated in [1 ] and Fig. 2. Filter H is designed in a way that it just reproduces the steep slopes in the transition range of the channel filter. Between these two slopes, repetitions of the filter’s H frequency response occur.

Filter Ha selects the number of repetitions for the passband. The gaps in the passband are filled with selected repetitions of the complementary filter H response as follows: The output of filter H is subtracted from the output of the delay element with filter’s H group delay to obtain the complementary filtered signal. Filter Fib selects the number of repetitions for complementary filtered signal. The responses of the upper and lower paths shown in Fig. 1 add up to an overall frequency response with a steep transition range.

Depending on the oversampling ratio N for the filter FI, the overall frequency response may have a very steep transition range. The steepness corresponds with a filter of the same length as filter FI, but all its taps being nonzero. Flence, the effort for filter FI reduces by about a factor of N. The overall savings are somewhat lower, since a few taps have to be spent for filters Fla and Fib.

The FRM approach can be optimized further for lowering the number of taps. In this case, individual frequency responses for filters FI, Fla and Fib may not be assigned easily to the original idea, but still fulfil the requirements for their combined frequency response.

The FRM method is superior to use a single FIR filter in the critical case of steep slope and narrow transition band for high bandwidth efficiency. It allows reducing the number of taps by a factor, which depends on the bandwidth efficiency. For filters with moderate bandwidth efficiency, the savings are low and a traditional single FIR filter performs better.

A drawback of using FRM method is the increased filter group delay. Though the increase of delay is not large and always smaller than the group delay of filter Fla or Fib, it only pays off for high bandwidth efficiency. For that reason, the FRM method shall be used typically for steep filters only, i.e. for high bandwidth efficiency.

A carrier may support one or more numerologies each comprising plural subcarriers. The subcarriers of each single numerology are orthogonal to each other and have a same subcarrier spacing between adjacent subcarriers. The subcarrier spacings of different numerologies may be different or the same. The subcarriers of different numerologies may or may not be orthogonal to each other (see [4], [5]).

References: [1 ] Y. C. Lim and Y. Lian, "The optimal design of one- and two- dimensional FIR filters using the frequency response masking technique," IEEE Trans. Circuits Syst. II, vol. 40, p. 88-95, Feb. 1993.

[2] 3GPP TS36.104 V15.0.0:

http://www.3gpp.Org/ftp//Specs/archive/36_series/36.104/3610 4-f00.zip

[3] http://www.3gpp.org/ftp/TSG_RAN/WG4_Radio/TSGR4_AHs/TSGR4_AH - 1801 /DOCS/R4-1800956.zip

[4] 3GPP TS38.104 V15.0.0:

http://www.3gpp.Org/ftp//Specs/archive/38_series/38.104/3810 4-f00.zip

[5] 3GPP TS38.21 1 V15.0.0

http://www.3gpp.Org/ftp//Specs/archive/38_series/38.21 1/3821 1 -fOO.zip

Summary of the invention

It is an object of the present invention to improve the prior art.

According to a first aspect of the invention, there is provided an apparatus, comprising weighting unit configured to weight each of plural input signals with a respective weight; transformer configured to perform an inverse Fourier transform on the plural weighted input signals to obtain a complex intermediate signal; channel filter configured to filter the complex intermediate signal by a filter function having a passband in frequency domain to obtain an output signal, wherein a frequency response of the filter function in the passband has a ripple relative to an ideal frequency response in the passband; each of the weights is an inverse of a respective frequency response with respect to the ideal frequency response in the passband.

Each of the input signals may be a subcarrier; at least a subset of the subcarriers may belong to a numerology; each of the subcarriers of the numerology may be orthogonal to each of the respective other subcarriers of the numerology.

The apparatus may further comprise determining unit configured to determine whether or not an input signal of the input signals is used; inhibitor configured to inhibit weighting the input signal if the input signal is not used. According to a second aspect of the invention, there is provided an apparatus, comprising channel filter configured to filter a received signal by a filter function having a passband in frequency domain to obtain an intermediate complex signal; transformer configured to perform a Fourier transform on the complex intermediate signal to obtain plural output signals having respective different frequencies; weighting unit configured to weight each of plural output signals with a respective weight; a frequency response of the filter function in the passband has a ripple relative to an ideal frequency response in the passband; each of the weights is an inverse of a respective frequency response at the respective frequency with respect to the ideal frequency response in the passband.

Each of the output signals may be a subcarrier; at least a subset of the subcarriers may belong to a numerology; each of the subcarriers of the numerology may be orthogonal to each of the respective other subcarriers of the numerology.

The apparatus may further comprise determining unit configured to determine whether or not an output signal of the output signals is used; inhibitor configured to inhibit weighting the output signal if the output signal is not used.

According to the first and second aspects, the frequency response in the passband may be normalized with the average frequency response in the passband being set to 1 ; and the inverse of the respective frequency response may be calculated as a reciprocal of the respective normalized frequency response multiplied by a positive weighting factor. The weighting factor may be predefined.

According to the first and second aspects, the apparatus may further comprise, a monitor configured to monitor an error vector magnitude of the output signal and a monitor configured to monitor a throughput of the output signals, respectively; and adapting means configured to adapt the weighting factor depending on a result of the monitoring.

According to the first and second aspects, the apparatus may further comprise a limiter configured to at least one of limit each of the weights such that each of the weights is not larger than a predefined maximum value; and limit each of the weights such that each of the weights is not smaller than a predefined minimum value. According to the first and second aspects, the ideal frequency response in the passband may be an average frequency response in the passband.

According to the first and second aspects, the apparatus may further comprise response transforming means adapted to Fourier transform the filter function to obtain a Fourier transformed filter function; determining means adapted to determine a respective deviation of the Fourier transformed filter function from the ideal filter function for each of the different frequencies; inverting means adapted to invert, for each of the different frequencies, the deviation in order to obtain the respective weight; setting means adapted to set the obtained weights in the weighting unit.

According to a third aspect of the invention, there is provided a method, comprising weighting each of plural input signals with a respective weight; performing an inverse Fourier transform on the plural weighted input signals to obtain a complex intermediate signal; filtering the complex intermediate signal by a filter function having a passband in frequency domain to obtain an output signal, wherein a frequency response of the filter function in the passband has a ripple relative to an ideal frequency response in the passband; each of the weights is an inverse of a respective frequency response with respect to the ideal frequency response in the passband.

Each of the input signals may be a subcarrier; at least a subset of the subcarriers may belong to a numerology; each of the subcarriers of the numerology may be orthogonal to each of the respective other subcarriers of the numerology.

The method may further comprise determining whether or not an input signal of the input signals is used; inhibiting the weighting of the input signal if the input signal is not used.

According to a fourth aspect of the invention, there is provided a method, comprising filtering a received signal by a filter function having a passband in frequency domain to obtain an intermediate complex signal; performing a Fourier transform on the complex intermediate signal to obtain plural output signals having respective different frequencies; weighting each of plural output signals with a respective weight; a frequency response of the filter function in the passband has a ripple relative to an ideal frequency response in the passband; each of the weights is an inverse of a respective frequency response at the respective frequency with respect to the ideal frequency response in the passband.

Each of the output signals may be a subcarrier; at least a subset of the subcarriers may belong to a numerology; each of the subcarriers of the numerology may be orthogonal to each of the respective other subcarriers of the numerology.

The method may further comprise determining whether or not an output signal of the output signals is used; inhibiting the weighting of the output signal if the output signal is not used.

According to the third and fourth aspects, the frequency response in the passband may be normalized with the average frequency response in the passband being set to 1 ; and the inverse of the respective frequency response may be calculated as a reciprocal of the respective normalized frequency response multiplied by a positive weighting factor. The weighting factor may be predefined.

According to the third and fourth aspects, the method may further comprise, if a monitor configured to monitor an error vector magnitude of the output signal, and a monitor configured to monitor a throughput of the output signals, respectively; and adapting the weighting factor depending on a result of the monitoring.

According to the third and fourth aspects, the method may further comprise at least one of limiting each of the weights such that each of the weights is not larger than a predefined maximum value; and limiting each of the weights such that each of the weights is not smaller than a predefined minimum value.

According to the third and fourth aspects, the ideal frequency response in the passband may be an average frequency response in the passband.

According to the third and fourth aspects, the method may further comprise Fourier transforming the filter function to obtain a Fourier transformed filter function; determining a respective deviation of the Fourier transformed filter function from the ideal filter function for each of the different frequencies; inverting, for each of the different frequencies, the deviation in order to obtain the respective weight; setting the obtained weights in the weighting unit.

Each of the methods according to the third and fourth aspects may be a method of filtering.

According to a fifth aspect of the invention, there is provided a computer program product comprising a set of instructions which, when executed on an apparatus, is configured to cause the apparatus to carry out the method according to any of the third and fourth aspects. The computer program product may be embodied as a computer-readable medium or directly loadable into a computer.

According to some embodiments of the invention, at least one of the following advantages may be achieved:

• Shorter channel filter impulse response, hence less signal latency and less implementation effort;

• Higher bandwidth efficiency and spectrum usage can be supported with the same channel filter length;

• Higher stop band attenuation can be provided with the same channel filter length;

• Invention can be combined with other filter tap saving methods;

• Reduced requirements to the channel filter unit saving HW resources and cost;

• Enabling of reduced guard band.

In general, relaxed inband ripple requirements for the channel filter can be traded against - shorter filter length,

higher spectrum usage,

narrower carrier (reduced guard band), and/or

higher stop band attenuation. It is to be understood that any of the above modifications can be applied singly or in combination to the respective aspects to which they refer, unless they are explicitly stated as excluding alternatives.

Brief description of the drawings Further details, features, objects, and advantages are apparent from the following detailed description of the preferred embodiments of the present invention which is to be taken in conjunction with the appended drawings, wherein:

Fig. 1 shows a block diagram of a FRM structure according to [1 j;

Fig. 2 shows simplified frequency responses inside an exemplary FRM filter according to Fig. 1 .

Fig. 3 shows a simplified block diagram for inband ripple compensation in TX according to some embodiments of the invention.

Fig. 4 shows a simplified block diagram for inband ripple compensation in TX with plural numerologies according to some embodiments of the invention.

Fig. 5 shows a simplified block diagram for inband ripple compensation in TX with plural numerologies according to some embodiments of the invention.

Fig. 6 shows a simplified block diagram for inband ripple compensation in RX according to some embodiments of the invention.

Fig. 7 shows a simplified block diagram for inband ripple compensation in RX with plural numerologies according to some embodiments of the invention.

Fig. 8 shows a simplified block diagram for inband ripple compensation in RX with plural numerologies according to some embodiments of the invention.

Fig. 9 illustrates positions of some FFT windows (for OFDM symbol extraction) relevant for determining EVM according to 3GPP.

Fig. 10 shows EVM depending on EVM window length for uncompensated shortened filters compared with uncompensated long filter.

Fig. 1 1 shows EVM depending on EVM window length for compensated shortened filters compared with uncompensated long filter (127 taps).

Fig. 12 shows an apparatus according to an embodiment of the invention;

Fig. 13 shows a method according to an embodiment of the invention;

Fig. 14 shows an apparatus according to an embodiment of the invention;

Fig. 15 shows a method according to an embodiment of the invention; and

Fig. 16 shows an apparatus according to an embodiment of the invention.

Detailed description of certain embodiments

Herein below, certain embodiments of the present invention are described in detail with reference to the accompanying drawings, wherein the features of the embodiments can be freely combined with each other unless otherwise described. However, it is to be expressly understood that the description of certain embodiments is given by way of example only, and that it is by no way intended to be understood as limiting the invention to the disclosed details.

Moreover, it is to be understood that the apparatus is configured to perform the corresponding method, although in some cases only the apparatus or only the method are described.

Some embodiments of the invention provide a method to achieve high bandwidth efficiency with short filter impulse responses in particular for OFDM modulated signals and/or DFTs-OFDM (used in e.g. in SC-FDM).

Some embodiments of the invention utilize the fact that the signal for filtering is OFDM modulated. That is, the subcarriers of the signal are orthogonal to each other. Thus, part of the spectral shaping is done after FFT for reception (e.g. UL of a base station) and prior to iFFT for transmission (e.g. DL for a base station). In the following, the part of the spectral shaping being performed after FFT or prior to iFFT, respectively, is sometimes denoted as being performed“inside FFT” and“inside iFFT”, respectively, because of its close relationship to FFT and iFFT, respectively, in the signal flow, compared to its distance to channel filtering in the signal flow. The expression“inside (i)FFT” does not exclude that the part is implemented as a unit separate from the unit performing (i)FFT.

Performing part of the spectral shaping inside FFT and iFFT, respectively, relaxes the requirements for the channel filter considerably. The additional effort inside the FFT / iFFT is marginal and less than a single filter tap of the channel filter.

In addition, for mMIMO systems supporting the implementation in the radio unit of frequency domain digital beamforming, more savings are achieved and they scale with the ratio between the number of spatial layers and the number of TRX pipes.

Embodiments of the invention may be either used on their own or combined with the FRM approach described in the prior art section or with other methods for additional reduction of implementation effort and signal latency. Fig. 3 provides a simplified block diagram for inband ripple compensation in a transmitter according to some embodiments of the invention. A real transmitter may have more blocks than those shown in Fig. 3. The inband ripple compensation is performed by the application of subcarrier specific weights prior to iFFT operation. In case of mixed subcarrier numerologies as in 5G NR, URLLC application and possibly in future also for 4G, the ripple compensations are calculated and applied at the position of all subcarriers of all subcarrier numerologies within the carrier. After iFFT, the resulting complex signal (intermediate signal) comprising a sequence of symbols goes to CP insertion, wherein a guard period is inserted between adjacent symbols and a part at the end of the subsequent symbol is copied into the guard period preceding the respective symbol (called“CP insertion”). The intermediate signal with CP inserted is then forwarded to the channel filter, and from there to upconversion to RF. The process from iFFT to upconversion is conventionally known.

Fig. 4 shows a simplified block diagram for inband ripple compensation in a transmitter with multiple (here: 2) numerologies according to some embodiments of the invention. In Fig. 4, the subcarriers of numerology 1 (“used subcarriers 1”) have different subcarrier spacing than those of numerology 2 (“used subcarriers 2”), as indicated by the different spacings between the input lines. As shown in Fig. 4, subcarrier specific weights are assigned to the subcarriers of all the numerologies. Different channel filters are used for numerology 1 and numerology 2. The weights applied to the subcarriers of numerology 1 correspond to the ripple of channel filter 1 (the inverse thereof), and the weights applied to the subcarriers of numerology 2 correspond to the ripple of channel filter 2 (the inverse thereof).

Fig. 5 shows a simplified block diagram for inband ripple compensation in another transmitter with multiple (here: 2) numerologies according to some embodiments of the invention. In contrast to the embodiments of Fig. 4, in Fig. 5, the signals from both numerologies are added before the channel filter. Thus, a single channel filter is applied to both numerologies. Correspondingly, the subcarrier specific weights for both numerologies correspond to the ripple of the (single) channel filter,

In embodiments with more than two numerologies, the embodiments of Figs. 4 and 5 may be combined to a hybrid solutions, wherein the signals of some numerologies are separately filtered by respective channel filters, and the signals of other numerologies are added first and then filtered by (one or more) common channel filters. In each of these cases, the subcarrier specific weights correspond to the ripple (the inverse of the ripple) of the respective channel filter.

Some embodiments of the invention apply similarly for reception (see Fig. 6 showing a simplified block diagram of a receiver). A real receiver may have more blocks than those shown in Fig. 6. In reception, a received signal passes through channel filter, CP removal, and FFT, as conventionally known. The subcarriers obtained by FFT are weighted with subcarrier specific weights which are inverse to the ripples of the channel filter. That is, the inband ripple of the channel filter is compensated after FFT operation by subcarrier specific weights. In case of mixed subcarrier numerologies as in 5G NR, URLLC application and possibly in future also for 4G, the ripple compensations are calculated and applied at the position of all subcarriers of all subcarrier numerologies within the carrier. A basic difference to transmission (downlink for a base station) is, that there is no 3GPP EVM requirement for reception (UL for the base station). So, throughput evaluation may be carried out instead, which is expected to allow for larger ripple being compensated.

Figs. 7 and 8 show further embodiments related to RX with plural numerologies. These embodiments correspond to those of Figs. 4 and 5 in TX, respectively. Hybrid solutions are possible in RX, too.

According to some embodiments of the invention, the channel filter may fulfill very relaxed inband ripple requirements and therefore gets along with a reduced number of taps. The resulting larger inband ripple is inverted and thereafter provided as subcarrier specific weights before iFFT operation.

More in detail, a ripple is considered as a deviation from an ideal signal response. For example, the ideal signal response is constant in the passband, independent from the frequency. For example, an average of the signal response in the passband may be considered as the ideal signal response. The borders of the passband may be defined as those frequencies, where the deviation from the average is larger than a predefined percentage, e.g. 5% or 10%, depending on implementation. If the thus determined borders are not symmetric around the intended center frequency, one may define the passband symmetrically around center frequency with one border being the determined border closer to the center frequency. Another ideal signal response may be one which compensates (equalizes) a frequency response of one or more other components of the system, such as an analog RF filter. In such cases, the other component(s) belong to the embodiment, too. In general, in a method of determining the subcarrier specific weights, the ideal signal response may be arbitrarily set.

For each frequency, a weight may be calculated as a reciprocal of a quotient of the actual signal response at the respective frequency and the average signal response in the passband. I.e., if the actual signal response is higher than the average (quotient >1 ), the weight of the corresponding subcarrier will be less than 1 , and vice versa. A power of the quotient may be used instead of the quotient (e.g. quotient 2 ).

In some embodiments, the inverse of the quotient (or the power thereof) may be multiplied with a positive weighting factor different from 1 before being applied as a weight. In some embodiments, the weights may be limited by a predefined maximum value (larger than 1 ) and/or a predefined minimum value (smaller than 1 ). If the calculated weight is larger than the predefined maximum value, the weight is set to the maximum value. If the calculated weight is smaller than the predefined minimum value, the weight is set to the predefined minimum value.

In some embodiments, weights may be applied to the used subcarriers but not to the unused subcarriers. The frequency positions of used and unused subcarriers are known from the configuration of the carrier and the corresponding FFT/iFFT size. As an example, LTE10 uses 600 out of 1024 subcarriers. In DL LTE10, this is 300 contiguous used subcarriers below the unused DC subcarrier and 300 contiguous used subcarriers above.

To avoid complicated implementations, temporarily used subcarriers may be always provided with weights. There is no harm if weights are provided to unused subcarriers, but there is no benefit, because unused subcarriers are empty. Flence, these embodiments require less multiplications than data appearing at iFFT output. CP insertion increases the number of data again, so a single channel filter tap according to the prior art requires significantly more multiplications than the weight application to the used subcarriers. An example for LTE20 DL illustrates the effort and the achieved savings:

Imagine that one has to spend an additional single filter tap, the effort would be a (real by complex) multiplication at time domain sample rate (e.g. 30.72 MS/s for LTE20). Before CP insertion, the rate is somewhat lower (28.67 MS/s for LTE20), whereas before iFFT multiplications must be performed just for the 1200 used out of 2048 subcarriers at an even lower rate (16.8 MS/s for LTE20).

After channel filtering, the signal may be upconverted to RF. To assess the signal quality, EVM may be evaluated at the channel filter output. In some embodiments, the weighting factor may be adapted based on a result of EVM evaluation. In some embodiments, EVM may be measured online and the weighting factor (such as the exponent explained further below) may be adapted based on the measurement result.

In an example embodiment, in order not to increase too much the required iFFT resolution, weights were limited to a maximum of 2, which corresponds to -6dB inband ripple. Because of symbol-wise compensation and CP insertion, no complete compensation is expected anyways.

A performance study was carried out for an N10 5G 10 MHz carrier with reduced guard band, so that a steep filter with 127 taps is needed, if no ripple compensation is provided. This 127 tap filter and the corresponding EVM serves as a reference for prior art without inband ripple compensation.

There were three additional filters designed with reduced number of taps and increased inband ripple. The EVM results are shown in Table 1 . Because of the larger inband ripple, EVM increased dramatically without ripple compensation but to a much lesser extent with ripple compensation. But even with inband ripple compensation, EVM could not always be reduced below the reference value of the 127tap filter according to 3GPP TS 36.104 [2]

Table 1 : EVM results for different channel filter lengths

3GPP TS 36.104 [2] defines a window length for EVM measurement used to quantify the performance of a digital radio transmitter or receiver. If the current LTE EVM window length is preserved in 3GPP for NR (see 3GPP TS 38.104), only about 30% of the channel filter taps can be saved.

A reason for the remaining EVM in Table 1 even with ripple compensation may be understood, if the 3GPP EVM definition is investigated more carefully, as explained in [3]:

For EVM evaluation, 3GPP defines in [2] two FFT time window positions for both, the signal under test as well as the ideal reference signal. Fig. 9 illustrates the time window positions for FFT transformation (cyan) and the inter-symbol interference (red), coming from the delay spread of adjacent symbols. Typically, the filters have a symmetric impulse response; hence the optimum FFT window (green) with lowest impact of inter symbol interference is located at the center position between the symbol boundaries.

In detail, Fig. 9 illustrates FFT time window position in OFDM symbol n, which is suffering from inter-symbol interference from adjacent symbols n+1 and n-1 . For symmetric delay spread, the centered green FFT window constitutes the optimum. Both 3GPP defined FFT windows (cyan) experience higher inter-symbol interference. Cyclic prefix (CPn and CPn+1 ) boundaries are marked with dashed lines. 3GPP defines an offset of W E VM/2 from that optimum position in both directions. Officially, WEVM is named“EVM window length W”. The values are copied from 3GPP and provided in Table 2.

Table 2: EVM window length for normal CP, as copied from Table E.5.1-1 in 3GPP TS36.141

For LTE10, W E VM is 66 out of 72 cyclic prefix samples. This leaves just 3 samples on either side to protect from inter-symbol interference. Consequently, some inter-symbol interference will leak into the FFT window and generate EVM. While this was not that much of an issue at the time the EVM definition was established, it creates severe problems now:

- Around 1 percent of EVM contribution have not been a problem for modulation schemes up to 64QAM. Flowever, for modulation schemes as high as 256QAM or 1024 QAM, this constitutes a problem.

Bandwidth efficiency above 90% requires steeper filters with longer filter impulse response. As a result, more inter-symbol interference is expected to leak from adjacent OFDM symbols.

For these reasons, the inventors evaluated EVM for different EVM window lengths.

Without ripple compensation, EVM varies only a little with the EVM window length and choice of the test model. Fig. 10 shows EVM depending on EVM window length (see 3GPP 36.104 Annex E.5.1 ) for uncompensated shortened filters (55 and 37 taps) compared with uncompensated long filter (127 taps). EVM was evaluated for test models E-TM1 .1 and E-TM3.1 . The curves for both test models are almost identical and hence E-TM1 .1 results are not visible in the figure. The results for 91 taps are not present in this figure (about 3.1 % EVM). The situation changes with the iFFT ripple compensation (see Fig. 1 1 ). Fig. 1 1 shows EVM depending on EVM window length (see 3GPP 36.104 Annex E.5.1 ) for compensated shortened filters (91 , 55 and 37 taps) compared with uncompensated long filter (127 taps). EVM was evaluated for test models E-TM1 .1 and E-TM3.1 . Though significantly decreased, EVM heavily depends on the EVM window length. While EVM reaches 2.7% for 55 taps and 4.8% for 37 taps at default EVM window length (66 samples), EVM can shrink close to zero at low EVM window lengths. Thus, the EVM after ripple compensation comes mainly from inter-symbol interference.

From this point of view, an EVM window length of about 40 samples would be desirable. Flowever, EVM for the 91 tap filter can be reduced down to 0.88% at default EVM window length, which is below the EVM of the uncompensated 127 tap filter.

Though inband ripple compensation improves EVM in all cases, EVM contributions from inter-symbol interference are often above reference EVM for a long and uncompensated channel filter. If the current 3GPP EVM window length is preserved, only 30% of the channel filter taps can be saved.

There are two arguments for shortening the EVM window length:

lower EVM limits for high order modulation schemes (e.g. 256QAM and 1024QAM)

higher bandwidth efficiency, which requires steeper and thus longer filters. Currently, no reduction of EVM window length is envisaged. Flowever, if 3GPP EVM window length can be reduced to 60% of its original value, up to 70% of the channel filter taps can be saved. The topic of reducing EVM window length will be proposed to standardization, using above two arguments.

In some embodiments of the invention, the determination and setting of the weights may be performed in an automated way. That is, the following steps may be performed:

1 . Fourier transform of FIR

2. Evaluate ripple as the deviation (factor) from the ideal (wanted) frequency response over the passband, i.e. frequency range of the used subcarriers. In other words: By which factor function the passband frequency response of the FIR deviates from the ideal frequency response at each frequency of the passband? 3. Sample the ripple at subcarrier frequencies for all subcarrier numerologies within the carrier.

4. Invert the ripple to obtain the weights.

5. Set the weights in the weighting unit of the respective transmitter or receiver.

The order of the steps can vary in a way that sampling is performed after each of the other steps. Moreover sampling (step 3) may be combined with one out of the three other steps 1 , 2, and 4.

Fig. 12 shows an apparatus according to an embodiment of the invention. The apparatus may be transmitter such as a base station (e.g. a gNB or an eNB) or a terminal (such as a UE) or an element thereof. Fig. 13 shows a method according to an embodiment of the invention. The apparatus according to Fig. 12 may perform the method of Fig. 13 but is not limited to this method. The method of Fig. 13 may be performed by the apparatus of Fig. 12 but is not limited to being performed by this apparatus.

The apparatus comprises weighting means 10, transforming means 20, and filtering means 30. The weighting means 10, transforming means 20, and filtering means 30 may be a weighting unit, transformer, and channel filter, respectively. The weighting means 10, transforming means 20, and filtering means 30 may be a weighting processor, transforming processor, and filtering processor, respectively.

The weighting means 10 weights each of plural input signals having respective different frequencies with a respective weight (S10). Each of the weights is an inverse of a respective frequency response at the respective frequency with respect to an ideal frequency response in a passband of the filtering means 30.

The transforming means 20 performs an inverse Fourier transform on the weighted input signals to obtain a complex intermediate signal (S20).

Fig. 14 shows an apparatus according to an embodiment of the invention. The apparatus may be receiver such as a base station (e.g. a gNB or an eNB) or a terminal (such as a UE) or an element thereof. Fig. 15 shows a method according to an embodiment of the invention. The apparatus according to Fig. 14 may perform the method of Fig. 15 but is not limited to this method. The method of Fig. 15 may be performed by the apparatus of Fig. 14 but is not limited to being performed by this apparatus.

The apparatus comprises weighting means 130, transforming means 120, and filtering means 1 10. The weighting means 130, transforming means 120, and filtering means 1 10 may be a weighting unit, transformer, and channel filter, respectively. The weighting means 130, transforming means 120, and filtering means 1 10 may be a weighting processor, transforming processor, and filtering processor, respectively.

The filtering means 1 10 filters a received signal by a filter function having a passband in frequency domain to obtain an intermediate complex signal (S1 10). A frequency response of the filter function in the passband has a ripple relative to an ideal frequency response in the passband. For example, the ideal frequency response in the passband may be an average frequency response in the passband.

The transforming means 120 performs a Fourier transform on the complex intermediate signal to obtain plural output signals having respective different frequencies (S120).

The weighting means 130 weights each of the plural output signals with a respective weight (S130). Each of the weights is an inverse of a respective frequency response at the respective frequency with respect to the ideal frequency response in the passband.

Fig. 16 shows an apparatus according to an embodiment of the invention. The apparatus comprises at least one processor 410, at least one memory 420 including computer program code, and the at least one processor 410, with the at least one memory 420 and the computer program code, being arranged to cause the apparatus to at least perform the method according to one of Figs. 13 and 15.

Embodiments of the invention are described wherein a guard period with a CP is foreseen between symbols of the signal output from iFFT. Flowever, in some embodiments of the invention, nothing may be transmitted in the guard period. In such embodiments (e.g. for ultrawideband), the receiver may generate CP itself by copying the last part of the subsequent symbol into the guard period. In still other embodiments, the guard period may be omitted such that symbols follow directly one after the other. These embodiments might have a higher nominal data rate but they might suffer under higher intersignal interference such that the effective data rate might not be increased.

If embodiments of the invention are applied to mixed numerologies (i.e. multiple numerologies in one carrier), the subcarrier spacing of one numerology may be the same as the subcarrier spacing of another numerology or different therefrom. The subcarriers of different numerologies may be or may not be orthogonal to those of another numerology.

Embodiments of the invention may be applied to any radio technique using OFDM, such as LTE and LTE-A.

One piece of information may be transmitted in one or plural messages from one entity to another entity. Each of these messages may comprise further (different) pieces of information.

Names of network elements, protocols, and methods are based on current standards. In other versions or other technologies, the names of these network elements and/or protocols and/or methods may be different, as long as they provide a corresponding functionality.

If not otherwise stated or otherwise made clear from the context, the statement that two entities are different means that they perform different functions. It does not necessarily mean that they are based on different hardware. That is, each of the entities described in the present description may be based on a different hardware, or some or all of the entities may be based on the same hardware. It does not necessarily mean that they are based on different software. That is, each of the entities described in the present description may be based on different software, or some or all of the entities may be based on the same software. Each of the entities described in the present description may be embodied in the cloud.

According to the above description, it should thus be apparent that example embodiments of the present invention provide, for example, a base station such as a eNB or a gNB, or a component thereof, an apparatus embodying the same, a method for controlling and/or operating the same, and computer program(s) controlling and/or operating the same as well as mediums carrying such computer program(s) and forming computer program product(s). According to the above description, it should thus be apparent that example embodiments of the present invention provide, for example, a terminal such as a UE, a MTC device, an loT device etc, or a component thereof, an apparatus embodying the same, a method for controlling and/or operating the same, and computer program(s) controlling and/or operating the same as well as mediums carrying such computer program(s) and forming computer program product(s).

Implementations of any of the above described blocks, apparatuses, systems, techniques or methods include, as non-limiting examples, implementations as hardware, software, firmware, special purpose circuits or logic, general purpose hardware or controller or other computing devices, or some combination thereof.

It is to be understood that what is described above is what is presently considered the preferred embodiments of the present invention. However, it should be noted that the description of the preferred embodiments is given by way of example only and that various modifications may be made without departing from the scope of the invention as defined by the appended claims.