Background of the invention Local networks often make use of a communication line, such as a communication bus, over which a set of nodes communicates. A driver module in a master node applies power to the line, the driver module being switched to produce step changes in the power in the line to transmit signals to receivers in remote slave nodes over the line. The switched power signal activates the multiplexed remote nodes connected to the line and the line also selectively transmits signals from the remote nodes back to a central processing unit.

Such a bus is used in automotive vehicles, for example, the bus comprising either a single line or a twisted pair of conductors in which the current flows, the close coupling between the pair of conductors reducing their sensitivity to electromagnetic interference ('EMI'), that is to say reception of noise induced in the wires of the bus, and improving their electromagnetic compatibility ('EMC'), that is to say the radiation of parasitic fields by the currents flowing in the wires of the bus; both are critical parameters, especially in automotive applications.

Historically, in automotive applications, functions such as door locks, seat positions, electric mirrors, and window operations have been controlled directly by electrical direct current delivered by wires and switches. Such functions may today be controlled by ECUs (Electronic Control Units) together with sensors and actuators in a multiplexed Controller Area Network (CAN). The Controller Area Network (CAN) standard (ISO 11898) allows data to be transmitted by switching a voltage, at a frequency of 250kbauds to 1baud for example, to the multiplexed receiver modules over the twisted pair cable. The receiver modules may be

actuators that perform a function, for example by generating mechanical power required, or sensors that respond to activation by making measurements and transmitting the results back to the ECU over the bus.

The CAN bus was designed to be used as a vehicle serial data bus, and satisfies the demands of real-time processing, reliable operation in a vehicle's EMI environment, is cost-effective, and provides a reasonable data bandwidth.

However, connecting with the main body network directly via a CAN bus system can be expensive because of increased costs per node and because high overall network traffic can make management extremely difficult. To help reduce costs, the logical extension is to structure the network hierarchically.

A variant on the CAN standard is the LIN (Local Interconnect Network) sub- bus standard (see ISO 7498), which is an extension to the CAN bus, at lower speed and on a single wire bus, to provide connection to local network clusters. A LIN sub-bus system uses a single-wire implementation (enhanced IS09141), which can significantly reduce manufacturing and component costs. Component costs are further reduced by self-synchronization, without crystal or ceramics resonator, in the slave node. The system is based on common Universal asynchronous receiver and transmitter serial communications interface (UART/SCI) hardware that is shared by most micro-controllers, for a more flexible, lower-cost silicon implementation.

It is often necessary to control accurately the shape of the leading and/or trailing edges of switched signals transmitted over the communication line. This is particularly the case where it is desired to minimise electromagnetic interference, by limiting the basic frequencies of electromagnetic emissions to certain acceptable frequency ranges and restricting the amplitudes of emissions of harmonics of the basic frequencies outside the acceptable range.

It is possible to generate signals to be transmitted with controlled shape by generating a step function current charging a capacitor to approximate the desired shape of the voltage signal, as shown in Figure 1. Such a generator is readily accommodated in an integrated circuit using digital techniques. However, the discontinuous changes in the charging current produce large rates of change of the voltage across the capacitor and applied to the communication line, which are

a source of EMI harmonics. Also, the capacitor stage has high impedance, with correspondingly low current produced by this stage, so that several follower stages are needed and its components are subjected to the full voltage swings across the capacitor, so that they need to withstand such relatively high voltages.

It is possible to generate signals to be transmitted with controlled shape by analogue circuits, such as those described in US patent specifications 5 530 388, US patent specification 6072340 and US patent specification 6259303, for example. However, the circuits described in these specifications do not give an optimal compromise between the requirements of low EMI and low circuit complexity, especially in the context of an integrated circuit implementation.

Summary of the invention The present invention provides a transmitter for a controlled-shape switched signal on a communication line as described in the accompanying claims.

Brief description of the drawings Figure 1 is a block schematic diagram of a known signal generator in a transmitter for a communication bus line, Figure 2 is a block schematic diagram of a signal generator in a transmitter for a communication bus line, in accordance with one embodiment of the present invention, given by way of example, Figure 3 is a more detailed schematic diagram of the signal generator of Figure 2, Figure 4 is a diagram illustrating the variation with time of an output voltage of the signal generator of Figures 2 and 3, Figure 5 is a diagram illustrating the variation of a feedback current appearing in operation of the signal generator of Figures 2 and 3, as a function of a feedback voltage, Figure 6 is a diagram illustrating the variations with time of a charging current and the output voltage of the signal generator of Figures 2 and 3,

Figure 7 is a schematic diagram of another embodiment of a signal generator in a transmitter for a communication bus line in accordance with the present invention, and Figure 8 is a diagram similar to Figure 5 for the signal generator of Figure 7.

< : < Detailed description of the preferred embodiments Figure 1 shows a known communication system, comprising a LIN communication bus line 1, a power supply line 2, and a ground line 3 enabling communication between a transmitter node 4 and a receiver node 5. The receiver node 5 is shown schematically as comprising a diode 6 connected to the positive supply line 2, a resistor 7 connected in series with the negative pole of the diode 6, and a capacitor 8 connected between the other pole of the resistor 7 and the ground line 3, the LIN bus being a single conductor connected to the junction between the resistor 7 and the capacitor 8. It will be appreciated that the receiver node 5 is in practice associated with elements (not shown) such as signal processors and actuators for exploiting the received signals. The communication system comprising the lines 1,2 and 3 and the transmitter node 4 and receiver node 5 may be installed in a vehicle or may be included in industrial applications, for example. It will also be appreciated that the system may be adapted to other communication standards, such as the CAN standard. According to the LIN standard, as shown, the physical LIN bus medium is a single wire connected via the termination resistor (pull-up) 7 to a positive voltage level, which, in automotive applications, will typically be the positive battery node. The idle state (communication pulse de-asserted) is high or recessive and the active state (communication pulse asserted) is low or dominant.

The transmitter node comprises an output stage 9 comprising a diode 10 whose positive pole is connected to the positive voltage supply line 2 and whose negative pole is connected to one terminal of a resistor 11, the emitter and

collector terminals of a PNP transistor 12 being connected in series between the other terminal of the resistor 11 and the ground line 3.

The signal generator also comprises an input stage 13. A rectangular pulse data signal is applied to an input terminal 14, which is connected to control the opening and closing of a switch 15 connected to the ground line 3. A first current source 16 is connected in series with a second current source 17 through the switch 15 to ground. The connection 18 between the current sources 16 and 17 is connected to one pole of a capacitor 19 whose other pole is connected to the ground line 3. A current shape generator 20 receives a triggering input from the data signal input terminal 14, and controls the currents passed by the current sources 16 and 17, so that the currents change step-wise as indicated schematically. The current passed by the source 17 when the switch 15 is closed is 2. lush, where ISH is the current passed by the current source 16.

In operation, the capacitor is charged to the voltage of the supply line 2 while the input signal at the terminal 14 is de-asserted. When the input signal at the terminal 14 is asserted, the switch 15 is closed and the net current ISH supplied by the current sources 16 and 17 increases step-wise to discharge the capacitor 19, the voltage across the capacitor 19 decreasing as a function of the integral of the current ISH. At the falling edge of the input pulse, the switch 15 is opened, and the net current flowing to the capacitor 19, from the current source 16 alone, is inverted and increases step-wise until the capacitor is fully charged.

In practice, the order of magnitude of the current ISH is low, typically 10 micro amps maximum, for example, and due to this high impedance of the input stage, several follower stages 21 are required. The follower stages 21 apply a current proportional to the voltage across the capacitor 19 to the base of the transistor 12, so as to apply the output signal to the LIN bus 1.

The use of the digital current shape generator 20 enables different shapes of the rising and falling edges of the signal applied to the LIN bus 1 to be synthesised. The signal generator therefore enables the corners of the rectangular input pulse to be rounded. However, the step-wise nature of the charging current of the capacitor 19 introduces high rates of change of the charging current with time and introduces harmonics of the base frequency of the pulse edge on the LIN

bus 1. Also, the current shape generator 20 is a relatively complex circuit, typically including several voltage comparators. Moreover, the current sources 16 and 17 are exposed to the full range of voltages across the capacitor 19, increasing the semiconductor area of the elements required to withstand the high voltages.

Figure 2 shows a signal generator in a transmitter in accordance with one embodiment of the present invention. The receiver node is again shown at 5 and the transmitter node at 4, with an output stage 9. A single follower stage 21 supplies the base of the transistor 12 of the output stage 9 with a current that is proportional to the voltage across the capacitor 19.

The voltage across the capacitor 19 is also supplied to a voltage generator 22 in a feedback loop. The voltage generator 22 produces a first voltage V that is equal to the voltage Vo across the capacitor 19 up to a clamp value, of 5 volts in this example, and a second voltage VH that is equal to the supply voltage Vsup minus the voltage Vo across the capacitor 19, the voltage VH being also limited to the same clamp value. A selection circuit 23 produces a feedback voltage VE by selecting the lower of the two values VL or VH. The feedback voltage VE is applied across a resistor R in a feedback current generator so as to produce a feedback current IE which is equal to VE/R, the feedback current IE being limited to a clamp value IMAX. The feedback current IE is applied to a charging current generator 25 that selectively inverts the feedback current IE to produce a charging current Ic that is applied to the capacitor 19. The input signal is applied to trigger the voltage generator 22 and pilot the charging current generator 25.

The voltage generator 22, selection circuit 23, feedback current generator 24 and charging current generator 25 constitute a feedback loop. In operation, at the rising edge of the rectangular data pulse applied to the input terminal 14, the voltage across the capacitor 19, initially at the supply voltage Vsup is applied to the voltage generator 22. The voltage VH is initially smaller than the voltage VL and is selected by the selection circuit 23. The feedback voltage VE applied to the resistor R in the feedback current generator 24 produces a feedback current IE that is applied by the charging current generator 25 to the capacitor 19 in a direction so as to reduce the charge of the capacitor and hence its voltage Vo. Accordingly, the voltages VH and VE increase exponentially until the clamp voltage of the generator

22 is reached. The feedback VE and the feedback current IE are then maintained at constant values, so that the voltage Vo continues decreasing but linearly with time.

When the voltage VL becomes smaller than the voltage VH, the selection circuit 23 selects the voltage VL. The voltage VE then decreases exponentially with time, so that the feedback current IE also decreases exponentially with time, and the voltage Vo across the capacitor 19 changes progressively more and more slowly.

This half cycle is then repeated at the falling edge of the input pulse, with the charging current Ic inverted by the inverter 25, so that the voltage Vo across the capacitor 19 progressively approaches the voltage Vsup.

A preferred embodiment of the signal generator shown in Figure 2 is illustrated in Figure 3. The voltage generator 22 comprises a first voltage generator element 26, whose input is connected to receive the Vo voltage across the capacitor 19 and a second voltage generator element 27 having an inverted input connected to receive the voltage Vo and a direct input connected to receive the supply voltage Vsup. The output of the first generator element 26 produces the feedback voltage VL and the output of the second generator element 27 produces the feedback voltage VH.

The selection circuit 23 comprises a pair of diodes 28 and 29 connected respectively to receive the voltages VL and VH on their negative poles, the positive poles being connected to the base of a transistor 30. The collector of the transistor 30 is connected to an internal power supply line 31 at a substantially lower voltage than the LIN bus power supply line 2. A current generator 39 is connected between the supply line 31 and the base of the transistor 30. The emitter of the transistor 30 is connected to one terminal of the resistor R of the feedback current generator 24.

The feedback current generator 24 also comprises a first field effect transistor ('FET') 32 whose drain is connected to the other terminal of the Resistor, and whose source is connected to the ground line 3. The gate of the FET 32 is connected to the gate of a further FET 32, whose source is also connected to the ground line 3 and whose drain is shorted to its gate and is also connected through a current generator 34 to the voltage supply line 31.

The charging current generator and inverter 25 comprises first and second current mirrors 35 and 36 that are driven by the current IE in the transistor 30 and the resistor R as indicated by the arrows 37 and 38, the current flowing in the mirror 35 being equal to IE and the current generated in the mirror 36 being 2. JE.

The current mirror 35 is connected to the capacitor 19 directly and the mirror 36 is connected to the capacitor 19 through the switch 15.

In operation, the diodes 28 and 29 select the lower of the voltages VL and VH at the outputs of the voltage generator elements 26 and 27. As long as the selected voltage is positive, it is applied across the resistor R and the FET 32, the FET 33 and current generator 34 serving to clamp the feedback current IE at a maximum value IMAX. The operation of the switch 15 serves to select the inversion or not of the charging current Ic applied to the capacitor 19. A starting current for the transistor 30 is provided by the current generator 39 connected between the voltage supply line 31 and the base of the transistor 30.

Referring now to Figure 4 and 5, it will be seen that, during a first phase 40, starting at the rising edge of the data signal at the terminal 14, the voltage Vo decreases as an exponential function of time, its rate of decrease progressively increasing. During a second phase 41, the voltage Vo decreases linearly with time, and during a third phase 42, the voltage Vo continues to decrease as an exponential function of time, but with its rate of change progressively decreasing.

The voltage Vo subsequently remains constant during a fourth phase 43. It will be appreciated that, throughout the three phases 40 to 42, the feedback current IE and the charging current lc have continuous magnitudes that are progressive functions of the Vo, so that the rate of change of the Vo never exceeds maximum values as would be the case with step-wise, discontinuous changes in the charging current. Subsequent phases 44,45 and 46 mirror the phases 42,41 and 40 with the current inverted so that the capacitor is progressively charged up to the Vsup in response to the falling edge of the data signal at the terminal 14.

The charging current is proportional to the feedback voltage VE from the selection circuit 23 divided by the value of the resistor R until the current reaches the maximum value IMAX defined by the generator 34 and FET 33. Figure 6 shows the resulting values of the charging current Ic and the output Vo as functions of

time. This half cycle of variation of the output voltage corresponds with a close approximation to a pure sinusoidal half cycle, so that electromagnetic emissions on the LIN bus 1 are limited to a single base frequency with little contribution from harmonics of that base frequency. The base frequency is a relatively low frequency, which can readily be chosen to be compatible with acceptable standards.

Figure 7 shows a variant of the embodiment of the invention shown in Figure 3 in which the resistor R is replaced by resistors Ri and R2 in parallel and the FET 32 is replaced by FETS 46 and 47 in series respectively with the resistors R1 and R2, The values of the resistors Ri and R2 are chosen so that the application of a voltage VE to the resistors Ri and R2 with the FETS 46 and 47 conducting will divide the feedback current IE so that a part IR of the feedback current te flows through the resistor Ri and a part equal to three times IR flows through the resistor R2. The variation of the overall feedback current IE with the voltage VE is shown in Figure 8, which is similar to the diagram of Figure 5. The sum of the part currents in the Ri and R2 is equal to four times IR until the gate-source voltage of the FET 47 reaches that of the FET 33 and the feedback current IE then reduces to the part current IR until the part current IR reaches the value IMAX. The resulting feedback current IE remains a continuous function of the feedback VE but its rate of change is initially faster than in the embodiment of Figures 2 and 3 and subsequently slower. It is found that this variant gives an even closer approximation to a sinusoidal half cycle of the rising and falling edges of the Vo with consequently even less emissions of harmonics of the base frequency.