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Title:
VARIABLE DC-DC CONVERTER AND NO-LOAD CLAMP
Document Type and Number:
WIPO Patent Application WO/2020/021279
Kind Code:
A1
Abstract:
A variable DC-DC converter is provided. When unloaded or lightly loaded, the main output of a DC-DC converter is subject to voltage spikes. The variable DC-DC converter is therefore fitted with a no-load voltage clamp in which a rectified, filtered and loaded output, termed a control signal, is used to drive a switch. The switch switches in a load resistance across the output terminals of the converter when the terminals are unloaded or lightly loaded. Due to a combined rectification, smoothing and filtering action of the control signal circuit, the control signal presents a steady voltage of exactly the output voltage selected by the user of the converter based on their specific output requirements. The control signal is therefore not subject to the voltage spikes that the main output is subject to. The circuit compares the control signal to the output voltage, and switches in the load resistance when the voltage at the output rises above the control signal.

Inventors:
FRANK WARNES FRANK (GB)
Application Number:
PCT/GB2019/052099
Publication Date:
January 30, 2020
Filing Date:
July 26, 2019
Export Citation:
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Assignee:
MURATA MANUFACTURING CO (JP)
MURATA POWER SOLUTIONS MILTON KEYNES LTD (GB)
International Classes:
H02M3/335
Foreign References:
JPS57185523A1982-11-15
JPH0580185U1993-10-29
JPS6359766A1988-03-15
Attorney, Agent or Firm:
REDDIE AND GROSE LLP (GB)
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Claims:
CLAIMS

1 . A variable DC-DC converter, comprising:

an input circuit connected to a primary winding, the input circuit having a switch for switching a voltage across the primary winding and a controller for supplying a driving frequency to the switch to output a desired voltage at the output;

an output circuit connected to a secondary winding, the secondary winding having a pair of winding terminals;

the output circuit including:

a first rectifier;

a pair of output terminals coupled respectively to each of the winding terminals; a smoothing circuit connected between the output terminals;

a voltage clamp connected between the secondary winding terminals, the voltage clamp including:

a switch, connected between the secondary winding terminals, for connecting/disconnecting a shunt load between the output terminals;

a low-pass filter connected between the secondary winding terminals and having a filter output, wherein the filter output outputs a control signal to the switch;

a second rectifier connected in series with the low-pass filter;

a load resistance connected to the filter output for loading the low-pass filter; and

wherein the low-pass filter provides the control signal output to the switch, such that the switch connects the shunt load across the secondary winding terminals, when the actual voltage across the output terminals becomes higher than the desired voltage, and disconnects it at other times.

2. The variable DC-DC converter of claim 1 , wherein the low-pass filter comprises a resistance and a capacitor connected in series across the secondary winding terminals, the low- pass filter output being located between the resistance and capacitor.

3. The variable DC-DC converter of claim 1 or 2, wherein the switch is a bipolar junction transistor, the base of the transistor being connected to the low-pass filter output, and the collector and emitter being connected between the output terminals.

4. The variable DC-DC converter of claim 1 or 2, wherein the switch is an operational amplifier, and wherein the inverting input of the operational amplifier is connected to the low- pass filter output, and the non-inverting input and output are connected between the output terminals.

5. The variable DC-DC converter of claim 4, wherein the voltage clamp comprises a third rectifier, wherein the non-inverting output is connected to the output terminal via the third rectifier.

6. The variable DC-DC converter of any preceding claim, wherein the smoothing circuit comprises a capacitor connected in parallel between the output terminals.

7. The variable DC-DC converter of claim 6, wherein the smoothing circuit further comprises an inductor, in series with the capacitor.

8. The variable DC-DC converter of any of preceding claim, wherein the converter is a flyback converter.

9. The variable DC-DC converter of any of claims 1 to 7, wherein the converter is a forward converter.

10. The variable DC-DC converter of any preceding claim, wherein the shunt load is a shunt resistor.

1 1 . The variable DC-DC converter of any preceding claim, wherein in operation the actual voltage across the terminals becomes higher than the desired voltage due to leakage inductance between the primary winding and the secondary winding.

12. The variable DC-DC converter of any preceding claim, wherein in operation the actual voltage across the terminals becomes higher than the desired voltage due to a minimum on- time limitation of the controller. 13. The variable DC-DC converter of any preceding claim, wherein the first, second or third rectifiers are diodes.

14. The variable DC-DC converter of any preceding claim, further comprising a second primary winding connected to the controller via a smoothing circuit and a voltage divider.

15. The variable DC-DC converter of claim 14, wherein the voltage divider is connected to ground and the ground-leg of the voltage divider comprises a variable resistance.

16. The variable DC-DC converter of claim 15, wherein the duty cycle of the driving frequency is set by a combination of the second primary winding voltage and the variable resistance.

Description:
VARIABLE DC-DC CONVERTER AND NO-LOAD CLAMP

Background of the Invention

The invention relates to a variable DC-DC converter, and in particular one having a no-load clamp.

Isolated DC-DC converters use a transformer to electrically isolate a DC power supply side of the converter from a DC output side of the converter. A switch in the DC supply circuit switches on and off a primary winding of the transformer which induces a voltage in the secondary winding which is coupled to the DC output circuit. A pulse width modulator (PWM) controller operates the switch to provide the switched voltage in the primary transformer winding. The switched voltage is transferred to the secondary winding by magnetic coupling, and the DC output voltage is derived by a combination of rectification and filtering of the secondary winding voltage. The voltage output at the transformer primary is a function of the switched-on time (T on ) of the switch in each switching cycle T to tai. The DC voltage at the output is a function of the ratio of T on /Ttotai (the duty cycle) and the input voltage.

To provide closed loop control of the switch a further winding may be provided on the transformer in phase with the output side winding. This is designed to allow the PWM controller to detect over-voltages and reduce the T on of the switch to lower the voltage.

In a DC-DC converter, the DC output side circuitry conventionally comprises a diode or diodes and a smoothing circuit. When the voltage is induced in the secondary winding by the primary winding, power is applied across the output terminals and also stored in the capacitor or inductor of the smoothing circuit. When the capacitor or inductor is charged and the current collapses in the secondary winding, the voltage discharges over the output providing a constant output voltage across the output connection terminals.

In normal operation, the DC output of the DC output side of the converter is loaded, and the secondary winding supplies a voltage through the diode when induced by the primary windings. The voltage supplied by the secondary winding charges a smoothing circuit and supplies power to the load. The capacitor charges up to a desired value and then discharges over the load. This means that the voltage at the secondary winding increases to match the supply side voltage (multiplied by the transformer turns ratio).

A problem occurs when there is no load, or a very small load, connected across the DC output. In this state the voltage at the secondary winding can increase to more than 200% of the supply side voltage (multiplied by the turns ratio). This is due to undamped leakage inductance in the transformer. In normal operation, the voltage supplied by the leakage inductance is dissipated across the load. When the output is unloaded this voltage charges the capacitor, causing the voltage at the output to increase up to the voltage supplied by the leakage inductance, i.e. more than 200% of the expected voltage.

A further problem occurs due to the leakage inductances which are seen across the unloaded secondary windings. As the voltage at the output increases, due to there being no load to dissipate across, the PWM controller tries to reduce the T on further and further to make up for the over-voltage at the output. At a certain point, the PWM controller cannot reduce the switched-on time T on any further. This means that the PWM may not be able to achieve the low end of the converter’s control because the circuit’s required T on , to provide the lowest rated output voltage, is longer than the PWM’s minimum T on .

Mechanical and software solutions have been developed to address the problem of over-voltage at the output in lightly loaded or unloaded DC-DC converters. These have included attempts to improve the coupling between the windings to reduce the leakage inductance, as well as introducing pulse-skipping in the PWM which effectively multiplies the T totai to allow for longer T on times at lower voltages. These invariably affect the reliability and ripple voltage of the device.

Circuitry based solutions are also known from the prior art. For example, an output circuit where a load resistor is permanently connected across the output, is shown in Figure 2 as will be described later. This is a brute force type solution, which will prevent the problems associated with a lightly or unloaded output, but at the cost of having a constant load providing considerable inefficiency and reducing the maximum current capability of the circuit.

A second example may have an opto-isolator circuit which attempts to increase the reliability of the feedback circuit. This solves the problem of the leakage inductance effects not being reflected in the auxiliary feedback winding, and reduces the duty cycle, but does not solve the problem of minimum Ton. This is a complex and costly solution for a comparatively inexpensive piece of equipment.

A third example may have an output circuit with a Zener diode connected across the output, which will clamp at a voltage slightly higher than the output voltage (such as when the over-voltage spikes are present). This method can be applied to a fixed output dc-dc converter, but is not suitable for an adjustable DC-DC converter.

A fourth example may have an output circuit incorporating a constant current minimum load which is switched in when the output load goes below the minimum load level. However this also presents a permanent loss because of the current sense resistor and causes inaccuracy in the output voltage.

JP63-059766 shows a voltage clamp for a converter, which can switch in an internal load resistor when the main output of the converter rises above a desired value. A voltage divider provides a fixed reference voltage to the base of a transistor. Due to the way that the reference voltage is generated, the circuit does not operate sufficiently when connected to a variable DC-DC converter. The values of the resistors which make up the potential divider have to be selected for the desired output voltage. There exists a need for a truly variable no-load voltage clamp for a variable DC-DC converter.

We have appreciated that it would be desirable to provide a solution to the problems with the prior art discussed above.

SUMMARY OF THE INVENTION

The invention is defined in the independent claims to which reference should now be made. Advantageous features are set forth in the dependent claims.

Examples of the invention introduce an internal load across the output only when the converter is unloaded or lightly loaded and presenting over-voltage spikes at the transformer secondary windings, and prevent the internal load from being introduced across the output when an external load is connected across the output so as not to introduce a constant load which causes a permanent inefficiency in the circuit.

BRIEF DESCRIPTION OF THE FIGURES

Embodiments of the invention will now be described, by way of illustration only, and with reference to the drawings, in which:

Figure 1A is a set of voltage/time graphs showing the secondary winding voltage waveform of an unloaded output of a DC-DC converter;

Figure 1 B is a set of voltage/time graphs showing the secondary winding voltage waveform of the output of a DC-DC converter incorporating the present invention;

Figure 2 shows a prior art circuit diagram of a no load clamp for a DC-DC converter;

Figure 3 is a circuit diagram describing a first embodiment of the present invention;

Figure 4 is a circuit diagram describing a second embodiment of the present invention;

Figure 5 is a circuit diagram describing a third embodiment of the present invention;

Figure 6 is a circuit diagram describing a fourth embodiment of the present invention;

Figure 7 is a circuit diagram showing a DC-DC converter incorporating the present invention;

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

In brief, a DC output side circuit for a DC-DC converter will now be described in which a rectified, filtered and loaded output, known herein as a control signal, is used to drive a switch, which in one embodiment is a transistor. The switch can switch in a load resistance across the output terminals when the terminals are unloaded or lightly loaded. The control signal is not subject to the over-voltage issues that the main output is subject to. Due to a combined rectification, smoothing and filtering action of the control signal circuit, the control signal presents a steady voltage of exactly the desired voltage. The desired voltage is the output voltage selected by the user of the converter based on their specific output requirements. A rectified and unloaded main DC output is connected across the collector and emitter of the transistor, along with an internal load resistor. When the voltage at the emitter becomes higher than the loaded voltage (i.e. when there is no load across the output terminals) by an amount greater than the transistor base/emitter voltage, the transistor allows current flow through the emitter and collector and switches in the internal load resistor to damp the over-voltage spikes. In this sense the circuit effectively compares the control signal to the output voltage, and switches in the internal load resistance when the voltage at the output rises above the control signal.

Alternatively, the transistor can be replaced with an operational amplifier operating as a comparator, or replaced with an NPN transistor in which the load resistor is switched in when the base is higher than the emitter voltage.

An example will now be described in more detail.

Figure 1 a shows an example of an un-loaded output voltage waveform measured across the secondary side winding of a DC-DC converter not employing the present invention. The first diagram 10 in Figure 1 a shows the waveform at the secondary windings. The waveform first goes negative. This is a reflection of the primary switch connecting the primary winding across Vin. During this period primary current rises due to the inductance of the winding. When the primary winding is disconnected, it transfers the stored energy to the secondary winding and the voltage rises quickly in the positive direction. During this period secondary current is delivered to the output capacitor. As there is no load to discharge the capacitor and damp the secondary voltage waveform there is a spike that appears on the leading edge of the waveform and the waveform has a curved top causing the output capacitor to charge up to a higher voltage This is shown in the magnified waveform portion in the second diagram 12 of Figure 1a. This causes the PWM controller to attempt to reduce the duty cycle further than is possible. The third diagram 14 in Figure 1a shows a longer time scale of the waveform and it can be seen that minimum duty cycle cannot be reached and the PWM controller attempts to reduce the output voltage by missing pulses.

The first diagram 16 in Figure 1 b shows an example of a loaded output voltage waveform measured across the secondary side winding of a transformer of a DC-DC converter. It can be seen that when the output of the DC-DC converter is loaded, there is no voltage spike before the steady voltage is established. It is therefore possible for the PWM controller to produce continuous pulses at or above the minimum duty cycle, as shown in the second diagram 18 in Figure 1 b.

A circuit comprising the no-load clamp described below will present a waveform similar to the one shown in Figure 1 b regardless of whether an external load is connected or not.

Figure 2 shows a circuit diagram describing a first prior art circuit which has a permanent load resistor 214 connected across the output. The load resistor allows the capacitor 218 to discharge across the output when no external load is connected, however the resistor presents a constant resistance to the circuit which inhibits the current capacity of the circuit, and the current draw of the constant resistance is increased if the output is adjusted to a higher voltage.

Figure 3 shows a circuit diagram forming a first embodiment of the present invention. The circuit 30 forming a no-load clamp for an isolated adjustable DC-DC converter as applied to the secondary winding 32S of an adjustable DC-DC converter. Rectifier 316 is connected by its anode to a first terminal of the secondary winding 32S. The cathode of rectifier 316 is connected via capacitor 318 to the second terminal of the secondary winding 32S. Rectifier 34 is connected by its anode to the first terminal of the secondary winding 32S before rectifier 316. Resistance 36 is connected by a first end to the cathode of rectifier 34. Capacitor 38 and resistance 310 are connected in parallel by a first end to the second end of resistance 36. The second ends of capacitor 38 and resistance 310 are connected to the second output of the secondary winding 32S. The node created by the connection of capacitor 38 and resistance 310 to the second end of resistance 36 (node 36-38-310) is connected to the base of transistor 312. The collector of transistor 312 is connected via internal load resistor 314 to the second terminal of the secondary winding 32S. The emitter of transistor 312 is connected to the cathode of rectifier 316. The primary and secondary windings describe a transformer.

The primary and secondary windings provide an isolated link between a DC supply side connected to the primary winding 32P and a DC output side connected to the secondary winding 32S. Rectifiers 34 and 316 force the direction of current travel in the circuit and are thus connected to the positive side of the transformer secondary winding 32S.

Rectifier 34 along with capacitor 38 operate as a rectified output of the secondary winding 32S across the load resistor 310. This provides a permanently loaded output across the base of transistor 312 and the 0V terminal of the secondary winding 32S. The voltage supplied by the secondary winding 32S charges capacitor 38 which then discharges across resistance 310 providing a rectified output. As noted in the summary of invention, without the resistance 310 the output would be subject to high ripple voltages. In addition, resistance 36 operates with capacitor 38 to create a low-pass RC filter to provide a more stable output across the base of transistor 312 and the 0V terminal of the secondary winding 32S. With resistance 310 operating as a load across the rectifying circuit formed by rectifier 34 and capacitor 38, and also with the filtering action of resistance 36 and capacitor 38, a rectified output which overcomes the problems of an unloaded DC output of a DC-DC converter is provided. Particularly because of the filtering action of resistance 36 and capacitor 38, and the current gain of the transistor 312, resistance 310 can have a high value, and therefore low leakage current. This load is negligible in comparison to a load required to stabilise the main output of the DC output side of the DC-DC converter. As noted in the background of invention, a permanent high load resistor capable of providing the required load to stabilise the DC output causes inefficiencies and lowers the maximum current capacity of the output.

In the first embodiment described by circuit 30, transistor 312 is a PNP transistor. A PNP transistor will switch on when the voltage applied at the base is low compared to the voltage applied at the emitter. The voltage at the base must be lower than the voltage at the emitter by a certain amount to allow current flow through the transistor. Typically the voltage difference is approximately 0.6V.

The base of transistor 312 receives the control signal from the rectifier/filter circuit of components 34 to 310. Due to the action of components 34 to 310 this voltage will be an accurate output voltage corresponding to the desired output voltage. However this voltage is not suitable for use as the main output voltage of the DC-DC converter, as the resistors 36 and 310, particularly series resistor 36, would provide a constant load on the output, thereby reducing efficiency of the converter. As noted in the background of invention and the description accompanying Fig1 simply providing a resistor across the output of the converter introduces inefficiencies as the operative load connected across the output is increased by the permanent resistive load of the circuit.

The rectifier 316 and capacitor 318 represent a known rectification circuit for a DC- DC converter. If only these components are used, an unloaded or lightly loaded DC-DC converter will have the output shown in Figure 1a across the secondary winding. Lightly loaded may be understood as 1 % or less of the full rated load. As noted in the summary of invention, it is an object of the invention to introduce an internal load across the output only when the converter is unloaded and presenting an output as shown in Figure 1a. It is also an object of the invention to prevent the internal load from being introduced across the output when an external load is connected across the output. The output at the node joining the cathode of rectifier 316 to the first end of capacitor 318 (node 316-318) is therefore a rectified, but potentially unloaded output from the DC-DC converter. This output is subject to effects caused by the voltage spikes caused by leakage inductance between the primary and secondary windings when there is no load connected across the output.

The output from the node 36-38-310 is therefore compared to the output from node 316-318 by the transistor 312. When the output from node 316-318 climbs to a point 0.6V above the control signal output from node 36-38-310 due to there being no external load connected across the DC-DC converter output, the transistor 312 switches on and clamps the internal load resistance 314 across the output, thereby presenting a load to the DC-DC converter only when the output voltage exceeds the transistor voltage by an amount greater than the base-emitter voltage rating of the transistor.

In an ideal circuit, secondary winding 32S would present a square wave. The voltage supplied to primary winding 32P is a square wave of an amplitude equal to the supply voltage, and a duty cycle equal to the supply voltage/desired voltage. Due to leakage inductances at the secondary winding, the voltage at the secondary winding will peak at an amplitude much higher than the supply voltage, multiplied by the turns ratio of the transformer, if there is no load for the voltage to dissipate across.

When the supply to the primary winding is on, energy is stored in the air gap between the primary and secondary windings. When the supply to the primary winding is switched off, the energy in the air gap dissipates through the secondary winding, and induces a voltage in the secondary winding. When there is a load applied at the output of the circuit, the voltage spike which would be caused by the leakage inductance simply dissipates through the load. The voltage begins to charge capacitor 318, but any extra energy due to the leakage inductance goes to the load. The voltage waveform at the inlet to the capacitor does not spike.

When there is no load across the output, any extra energy due to the leakage inductance goes to capacitor 318 and remains there, as there is nothing for the capacitor to discharge across. The same voltage waveform is presented at capacitor 38, after going through resistance 36. There, the voltage waveform acts as if it were presented to a loaded circuit. Any extra energy presented at the capacitor 38 dissipates through the load resistance 310. In addition, series resistance 36 operates with capacitor 38 to reduce any spikes by acting as a low-pass filter. This is achievable by having a very small value of series resistance 36, such as 10 ohms. This presents a very small loss in the circuit, much smaller than would be required to permanently connect a load resistance for the main output. The parallel resistance 310 which operates as a load to capacitor 38 may have a value of around 20k ohms. Capacitor 38 may have a capacitance of 1 n farad.

Whilst the secondary winding is conducting, voltage is presented at the top of capacitor 38, through resistance 36 and diode 34 to the base of transistor 312, and to the top of capacitor 318 through diode 316. When the secondary winding stops conducting, the charge in capacitor 38 discharges over resistance 310 and continues to present a constant voltage at the base of transistor 312. As there is no load connected across the main output, the charge in capacitor 318 cannot dissipate because rectifier 316 becomes reverse biased preventing the charge from dissipating over the RC filter and load resistance circuit.

The emitter of transistor 312 is therefore presented with a voltage higher than the desired voltage, whilst the desired voltage is presented to the base of transistor 312. Once the voltage at the emitter reaches a point higher than the voltage at the base plus the transistor switching voltage, transistor 312 switches on so that resistor 314 is connected across the load. This allows the overvoltage present at the emitter to dissipate through resistor 314, thereby maintaining the output voltage at the desired voltage.

Because the voltage at the emitter is compared to a control signal having the desired voltage, the circuit can be applied to a variable DC-DC converter.

The operation of the circuit above is described with respect to a flyback converter. In a forward converter the secondary winding 32S conducts at the same time as primary winding 32P.

Figure 4 shows a second embodiment of the invention whereby the transistor 312 of Figure 3 is replaced by an operational amplifier 412. The initial output circuit of rectifier 44, resistance 46, capacitor 48 and load resistance 410 are connected and work in the same way as in the circuit of Figure 2 to provide a loaded, filtered and rectified control signal to one side of the op-amp 412 from the secondary side winding 42S. Rectifier 416 and capacitor 418 provide a final output circuit, which is an unloaded rectified output to the main outputs and also to the other side of the op-amp 412.

Op-amp 412 operates as a comparator. When the voltage at the non-inverting terminal of op-amp 412 becomes higher than the voltage at the inverting terminal, the op- amp 412 saturates and switches in the internal load resistor 414 to load the voltage. As in the previous embodiments, the circuit is comparing a filtered and loaded signal to an unloaded main output, and loading the output when it presents a voltage a certain amount higher than the filtered and loaded signal.

Rectifier 420 provides a voltage drop in the main output signal to the op-amp. This replicates the function of the base-emitter voltage drop of the transistor embodiment, so that the internal load resistance is only switched in once the main output voltage rises above the control signal voltage by a certain amount, for example 0.6V.

Figure 5 shows a third embodiment of the invention whereby the input to the power terminal of op-amp 412 is replaced by a single wire, and the input to the inverting terminal of op-amp 412 is connected to ground by resistance 522. This provides the same input voltage to op-amp 512 as the input to op-amp 412 in Figure 4, and the circuit operates in the same manner.

Figure 6 shows a fourth embodiment of the invention. The circuit operates by comparing a control signal with a main output signal, however the transistor used is an NPN transistor 612.

Figure 7 shows a DC-DC flyback converter incorporating an embodiment of the invention. The following description is for explanation of the DC-DC converter concept only, and the output side circuit can be applied to any suitable design of DC-DC flyback converter primary side. DC voltage source 71 provides power to the primary winding 72P1. For a flyback converter, the primary and secondary windings can be arranged around a core in a manner similar to that of a transformer, however voltage induced in the secondary winding 72S is not induced by‘transformer action’. Energy is stored in an air gap between the two windings, and thus when the voltage in the primary winding 72P1 collapses, this induces a voltage in the secondary winding 72S.

PWM controller 73 provides a driving voltage to the gate of transistor 75 to switch on and off the supply from DC voltage source 71 to primary winding 72P1. Rectifier 77 and capacitor 79 create a rectified voltage output from primary winding 72P2. This winding 72P2 is not driven by the voltage source, but instead reflects the instantaneous voltage from secondary winding 72S. This is a feedback winding which is connected to a voltage feedback terminal on the PWM controller. The voltage presented at the feedback terminal is compared to an internal reference voltage. Determining whether the feedback voltage is higher or lower than the reference voltage allows the PWM 73 controller to alter the duty cycle based on the voltage at the output. Through the action of negative feedback, the PWM controller 73 will increase or decrease the duty cycle in order to always maintain the same feedback voltage as the reference voltage. Resistors 71 1 and 713 in conjunction with variable resistor 715 create a potential divider, and the output from the potential divider is presented at the voltage feedback of the PWM controller.

By varying the resistance of variable resistor 715 the desired output voltage of the DC output side can be set. The PWM controller 73 will accept an input voltage from the potential divider and translate this into a duty cycle for the Vout terminal driving the gate of the switching transistor 75. Resistor 713 is present to set the maximum output voltage.

When the terminal Vfb supplied by the potential divider from the primary side auxiliary winding 72P2 receives an overvoltage, this sets the duty cycle ratio lower. However, at light output loads, the auxiliary winding 72P2 suffers from poor cross-regulation with the secondary windings, meaning that the output voltage can still rise above the desired voltage, and there are still problems associated with the minimum duty-cycle of the PWM controller. If the controller has a fixed duty cycle, the auxiliary winding and feedback circuit would not be required. Components 74, 76, 78, 710, 712, 714, 716 and 718 can operate in the same manner as components 24 to 218 of Figure 2. The embodiment of the invention incorporated into the DC-DC converter of Figure 7 is equivalent to the circuit of Figure 2, however the circuits of Figures 4, 5 and 6 and any other circuit described herein can be used in place of the output circuit.

The invention can also be implemented in a DC-DC forward converter. The invention applied to the forward converter works in the same manner as applied to the flyback converter 70 of Figure 7. The operation of the converter itself however is different, in that current is induced in the secondary side winding in phase with the primary side winding by the transformer effect. There is no energy stored in an air gap between the primary and secondary windings. The smoothing circuit of the forward converter comprises different components to that of the flyback converter, for instance, the smoothing circuit of the forward converter comprises an inductance.

Leakage inductance is a parasitic element of a circuit which can be present in any arrangement of inductors or windings. The circuits described in the above embodiments are therefore suitable for connection to any DC-DC converter or AC-DC converter which is subject to problems caused by a minimum on time limitation and/or leakage inductance as described above.

References in the preceding examples to a resistance may be understood to refer to a standard resistive component such as a resistor, or a thin film, metal film, wire wound, carbon film etc. resistor. Alternatively, references to a resistance may refer to a resistive length of wire or a section of wire of an appropriate length to provide a resistance, or an impedance presented by a component other than a resistor.

References in the preceding examples to a rectifier may be understood to refer to a diode. Alternatively, references to a rectifier may refer to any other component capable of forward biasing a circuit, or to an arrangement of components designed to have the same effect.

References in the preceding examples to a transformer indicate that there is a primary winding and a secondary winding coupled so that one winding induces a voltage in the other winding. The example of a transformer is not limited to any arrangement of windings and a core, and can indeed contain no core, a core with an air gap, a solid core, and any number of primary and secondary windings. The example of a transformer is not intended to limit the operation of a primary and secondary winding to inducing voltage via transformer action. References in the preceding examples to a PNP or NPN transistor may be understood to also refer to, respectively, P or N channel field effect transistors (FETs), or any other switch capable of having the desired effect, such as an operation amplifier.

References in the preceding examples to a capacitor may be understood to refer to any type of capacitor of a suitable size and voltage rating, such as a ceramic, film, power film or electrolytic capacitor.

Embodiments of the present invention may take the form of an embedded converter device, wherein the windings are disposed around a magnetic core embedded in a substrate. The converter device may advantageously be used as part of power switching electronic devices.

Described above are a number of embodiments with various optional features. It should be appreciated that, with the exception of any mutually exclusive features, any combination of one or more of the optional features are possible.

Various modifications to the embodiments described above are also possible and will occur to those skilled in the art without departing from the scope of the invention which is defined by the following claims.