Field of the Invention

This invention relates to an isolating transmission line transformer, and particularly to an isolating transmission line transformer for use within a data communications circuit or system.

Background of the Invention

Data communications and measurement equipment often needs to couple broadband signals to and from transmission lines with some D.C. and low frequency isolation, e.g. to reject common mode signals such as mains hum in ^earth loops'. A D.C. isolating transformer is commonly employed for this purpose. It is generally accepted, however, that the parasitic reactance of such known transformers will limit the upper usable frequency (fU) that may be communicated over the transmission line by introducing loss and mismatch. Further, (fL) the lower frequency limit will be limited by a shunt reactance to make it difficult to increase the ratio fU/fL beyond a certain limit, typically 100,000. There is therefore placed a limitation on the achievable overall bandwidth.

Another form of transformer is a Transmission line Transformer (TLT) in which the physical properties of the wires used for the transformer windings are considered and disposed in such a way as to also form part of a transmission line.

Currently only conventional isolating transformers are used in local and wide -area networks (LANs and WANs) and, in their current form, by virtue of the above characteristics, limit bandwidth- and are therefore not conducive to optimising the potential benefits of high speed networks, fibre optic backbones and networks, for example. Further information on TLTs is described in Sevick, J., Transmission Line Transformers, Noble Publishing Corp., 4 ^th edition, 2001 but this reference does not refer to an Isolating TLT.

US8456267 discloses an isolating TLT exhibiting a high impedance port, typically to couple analogue radio equipment to high impedance antennas, without significant loss.

US 7924130 discloses an isolation magnetic device having a single port and with multiple windings, the latter of which limits the upper frequency to an estimated 2 GHz operation. Applicant is of the view that the device disclosed therein will not meet isolation and return loss specifications for stable transmission in addition to producing a variation in performance, e.g. between individual Ethernet lanes and from device to device. Summary of the Invention

In a broad sense, there is provided an Isolating Transmission Line Transformer (ITLT) for use in data communications, the ITLT being arranged with first and second ports connected to respective first and second windings, the ports being d.c. isolated from one another.

A first aspect of the invention provides an isolating Transmission Line Transformer (ITLT) for use in a data communications system, the ITLT comprising:

- a core;

- a first port for connection to a data source;

- a second port for connection to a transmission or data receiver line, said second port being on substantially the opposite side of the core to the first port; a transmission line formed by first and second conductors, the first conductor being connected in series to the first port and the second conductor being connected in series to the second port to provide d.c. isolation between the ports, wherein the conductors are arranged side-by-side though or around the core, and each executes a single turn or winding through or around the core.

The transmission line preferably has a characteristic impedance substantially equal to half of that presented at the first and second ports.

The first and second conductors may be twisted about each other as they pass through or around the core. The first and second conductors may be twisted about each other such that the gap between the conductors remains substantially constant.

The core may be a binocular -type core comprising two substantially parallel bores extending between first and second ends of the core, the first port being positioned adjacent the first end and comprising two terminals with the first conductor extending from one terminal, through one of the bores, and returning through the other bore to be connected to the other terminal, and the second port being positioned adjacent the second end and comprising two terminals with the second conductor extending from one terminal, through one of the bores, and returning through the other bore to be connected to the other terminal. In another embodiment, the core may comprise first and second binocular -type cores, arranged side by side in general alignment, each of the first and second cores having first and second substantially parallel bores extending between first and second ends, the first port being positioned generally between the first and second cores and comprising two terminals with the first conductor extending from one terminal, through the first bore of the first core, and returning through its second bore and into the second bore of the second core, and returning through the first bore of the second core to the other terminal; and the second port being positioned generally between the first and second cores and comprising two terminals with the second conductor extending from one terminal, through the second bore of the second core, and returning through its first bore and into the first bore of the first core, and returning through the second bore of the first core to the other terminal.

In another embodiment, the core may be a pot core. The core may be formed of a ferrite material having a permeability of approximately ΙΟ,ΟΟΟμ. Other permeability figures can be used.

A second aspect of the invention provides a method of using an ITLT, according to any preceding definition, in a data communications system wherein one port is connected to a computer, computer modem, or data communications equipment and the other port is connected to a further transmission line or the like, and in which the data communications system is configured to transmit and/or receive data to and/or from the further transmissions line.

A third aspect of the invention provides method of constructing an Isolating Transmission Line Transformer (ITLT) for use in a communications system, the method comprising: providing a core; winding a transmission line comprising first and second conductors around the core to provide first and second coils, the first conductor being connected in series to a first port and the second conductor being connected in series to a second port, wherein the first and second ports are provided substantially opposite one another relative to the core; and each of the conductors executes only a single winding around or through the core.

A further aspect of the invention provides an Isolating Transmission Line Transformer (ITLT) for use in a data communications system, the ITLT comprising:

- a core;

- a first port for connection to a data source;

- a second port for connection to a transmission or data receiver line; and

a transmission line formed by first and second conductors, each conductor being wound about the core to provide first and second coils, the first conductor being connected in series to the first port and the second conductor being connected in series to the second port to provide d.c. isolation between the ports, the TLT being arranged such that there is (i) a substantially constant inter-winding gap (g) between the first and second conductors of the coils, and (ii) a substantially constant intra-winding gap (G) between adjacent windings of the same coil.

Each of the first and second ports may comprise first and second terminals, one or both of which is/are spaced apart by a distance substantially equal to g.

The ITLT may be arranged such that the transmission line has a characteristic impedance Zo which is, substantially, a predetermined fraction (1/N) of the characteristic impedance (s) presented at the first and second ports, wherein N is an integer. The ITLT may be arranged to provide a 1:1 impedance transformation ratio, and wherein Zo is substantially one-half of the impedance presented at the first and second ports. The ITLT may comprise multiple such transmissions lines connected in parallel between the first and second ports.

The intra-winding spacing G may be arranged so as to substantially minimise or reduce intra-winding capacitance.

The surface-area of each coil may be arranged so as to provide a predetermined minimum magnetising inductance.

The size of the core may be arranged to provide an increased bandwidth.

The ITLT may be arranged such that the first and second ports exhibit a substantially constant resistive characteristic impedance over a bandwidth of substantially greater than 100,000 x Fl, wherein Fl is the usable lower frequency.

The ITLT may be arranged such that the first and second ports exhibit a substantially constant resistive characteristic impedance over the frequency range 100 kHz > 4 GHz. The range may in other embodiments be extended up to 5 GHz. The range may in some embodiments extend up to 10 GHz, and beyond.

The ITLT may be arranged such that there is a substantially constant transmission delay between the ports over said frequency range.

The TLT may be configured to be connected using one port to a computer, computer modem, or data communications equipment such as a source of voice data, and at the other port to a further transmission line.

The ITLT may be formed of coaxial cable, stripline, microstrip or on a surface-mounted IC, PCB or Flexi PCB.

A further aspect comprises a data communications system comprising the ITLT of any preceding definition. A further aspect of the invention provides a data communications system, comprising:

- a data source providing data at a given bit rate or range of bit rates;

- a first transmission line or receiver line for carrying the data; and

an Isolating Transmission Line Transformer (ITLT) disposed between the data source and first transmission or receiver line for providing d.c. isolation between the two, the ITLT comprising:

- a core;

- a first port connected to the data source;

- a second port connected to the transmission or data receiver line; and

- a transmission line formed by first and second conductors, each conductor being wound about the core to provide first and second coils, the first conductor being connected in series to the first port and the second conductor being connected in series to the second port to provide d.c. isolation between the ports, the ITLT being arranged such that there is (i) a substantially constant inter-winding gap (g) between the first and second conductors of the coils, and (ii) a substantially constant intra-winding gap (G) between adjacent windings of the same coil. One of the ports may be positioned within the associated coil. Preferably still, said port is connected to a point at the centre of the coil, or substantially so. A further aspect of the invention comprises a method of constructing an Isolating Transmission Line Transformer (ITLT) for use in a communications system, the method comprising:

- providing a core;

- winding a transmission line comprising first and second conductors around the core to provide first and second coils, the first conductor being connected in series to a first port and the second conductor being connected in series to a second port, the windings being such that there is a substantially constant inter-winding gap (g) between the first and second conductors of the coils, and (ii) a substantially constant intra-winding gap (G) between adjacent windings of the same coil .

Brief Description of the Drawings

The invention will now be described, by way of non-limiting example, with reference to the accompanying drawings, in which:

Figure 1 is a system block diagram showing a data source coupled to a transmission line via a transmission line transformer; Figure 2 is a schematic diagram of a typical lumped transformer model, showing parasitic elements, which is useful for understanding the background of the invention;

Figure 3 is a schematic diagram of a typical isolating transformer that is characteristically dispersive and of limited bandwidth, useful for understanding the background of the invention;

Figure 4 is a schematic diagram of an isolating transmission line transformer which is useful for understanding the invention; Figure 5 is a close-up view of the coils of ^~ he Figure 4 embodiment, indicating an inter-winding gap and stray capacitance;

Figure 6 is a close-up view of the coils of the Figure 4 embodiment, indicating the intra-winding gap and stray capacitance;

Figures 7a and 7b show cross-sectional and axial views of a coaxial cable transmission- line which may be employed in the embodiment ;

Figures 8a and 8b show cross-sectional and axial views of a twin transmission line which may be employed in the embodiment;

Figure 9 is a perspective view of a physical implementation of the Figure 4 embodiment;

Figures 10a and 10b are, respectively, topological representations of a known transmission line transformer and a transformer in accordance with the invention;

Figures 11a and lib are alternative topological representations respectively corresponding with Figures 10a and 10b;

Figure 12 is a performance graph showing reflection delays relating to a known transmission line transformer;

Figures 13a and 13b are performance graphs relating to minute or smaller reflection delays in a transformer in accordance with the invention;

Figure 14a and 14b are top plan and side views of a physical implementation based on a Pot Core Ferrite using the Figure 4 embodiment, using only one turn for each conductor;

Figure 15 is a sectional view of an alternative physical implementation of a bead / binocular core a transformer according to the invention;

Figure 16 is a perspective view of the Figure 15 implementation; and

Figure 17 is a sectional view of an alternative physical implementation of a two bead / binocular transformer according to the invention. Detailed Description of Preferred Embodiments

Embodiments herein describes an isolating transmission line transformer (hereafter "ITLT") and data communications systems employing one or more such ITLTs which, by virtue of their design and construction, provide d.c. isolation with substantially seamless coupling between a source of data at one port and another data transmission means at the other port, particularly a transmission line (or data receiver line) for onwards transmission (or reception) of the data. In some embodiments, multiple ITLTs may be used to couple multiple transmission or reception lines together with regeneration to provide transmission and reception over greater distances. Advantageously, the ITLT of the present design and construction is found to permit data transmission and receive speeds with a much higher data rate than is conventionally known or available, whilst keeping the lower usable frequency relatively constant, or controllable. This provides a greater overall bandwidth than is currently available (the current bandwidth typically being in the order of 100,000 times the lower usable frequency).

Figure 1 shows a typical system in which the ITLT can be employed, comprising a digital data source 3 or a digital data receiver 3, the ITLT 1, and a transmission line 5 which provides transmission of the data to or from the distant end. The digital data source or receiver 3 is connected to the ITLT 1 by respective two-terminal ports, and the ITLT to the transmission line 5 by respective two-terminal ports, as shown.

The data source or receiver 3 can be a computer (e.g. a PC or laptop) , a data network, whether a LAN or WAN, audio equipment, digital television/video, telecommunications equipment or test and measurement equipment, to give some examples. Any source of digital data operating at broadband speeds can be used, particularly speeds above 256 kbit/s and up to 100 Gbit/s, and potentially beyond. The current state of the art limits current broadband bandwidth to the order of 1000MHz (10G Base-T for example is limited to 500MHz) whereas this embodiment enables the bandwidth to be increased to4000 MHz and upwards.

The electrical transmission line used in the construction of ITLT 1 can be any form of transmission line, such as parallel line, coaxial cable, stripline and microstrip, PCB or Flexi PCB and the like. The transmission line 5 can be embodied on a surface mounted integrated circuit (IC) or chip.

The ITLT 1 comprises the first and second ports, and at least two conductors forming a transmission line, wherein each conductor is wound about a core, e.g. a toroidal ferrite core, to provide first and second coils formed of adjacent windings, the first conductor being connected in series to the first port and the second conductor being connected in series to the second port. By virtue of this structure, there is d.c. and some low- frequency isolation between the ports, as is required, for example to reject common-mode signals such as mains hum in earth loops. As will be explained below, the transmission line of the ITLT 1 will have a known characteristic impedance Zo, this being provided by the manufacturer of the transmission line and/or which can be measured. By virtue of the design and arrangement of the ITLT 1, the characteristic impedance (s) Zl and Z2 which is/are presented at the first and second ports may be the same or different than Zo. Ultimately, however, it is important in the present context for the port characteristic impedances Zl and Z2 to substantially match the respective resistive impedances of the data source or receiver 3 and the transmission line 5. This will ensure seamless, or near seamless coupling by minimising reflections and therefore loss.

As will be appreciated, in conventional transformers, the characteristic port impedance (s) is or are frequency dependent and hence there is a limitation on usable bandwidth, particularly the upper usable frequency fU.

In the present embodiment, the design and arrangement of the ITLT 1 is such as to provide a relatively flat characteristic impedance and frequency response over a much wider bandwidth than conventional isolating transformers.

For context, Figure 2 depicts in schematic form a typical lumped model of an isolating transformer, or TLT, which is useful for understanding the limiting behaviour of conventional Isolating Transformers or TLT ¹ s . LI and L2 represent the physical coils formed of multiple windings, which provide mutual inductance M, whereas the additional elements L3, L4, L5, L6, CO, CI, C2 and C3 represent parasitic elements that limit performance, particularly high frequency performance.

In this embodiment, we provide, and will describe, an ITLT with a 1:1 impedance transformation ratio, i.e. whereby the characteristic impedances Zi = Z2 are appropriate where the data source or receiver 3 and transmission line 5 have the same characteristic impedance for seamless connection. However, it will be appreciated that other transformation ratios can be used, e.g. 1:2, 1:4, 1:9, 4:1, 9:1, Further the ITLT is not limited to just two ports, and multi-port topologies can be employed.

Figure 3 shows an embodiment of a commonly used TLT alternative for an Isolating Transformer that typically does not produce characteristic impedances at its ports, nor a constant transmission delay between them and as a result is necessarily dispersive and of limited bandwidth. Figure 4 is an embodiment of an ITLT which is useful for understanding the invention, formed of a first conductor 17 connected in series to first and second terminals of a first port (Port 1) and wound around a core to provide a first coil 19 formed of a plurality of windings. A second conductor 21 is connected in series to first and second terminals of a second port (Port 2) and wound around the core to provide a second coil 23 formed of the same number of windings. The ITLT provides a 1:1 transformation ratio. The dotted lines between the coils 19,23 indicate that the coils physically form a transmission line and indeed in this embodiment are formed by a length of RG179 Coaxial Cable of characteristic impedance 50 ohms, although other forms of transmission line with other characteristic impedances can be used. It will be noted that this embodiment of an Isolating TLT employs a different topology in that the second port (Port 2) has a centre output point (tap) within the second coil 23, which is found to be advantageous. In some embodiments, the second port may be slightly off-centre.

In Figure 4, at the physical, constructional level, windings 19 and 23 are arranged around the core in such a way as to form a transmission line between them.

Figure 7a shows the cross-section of a coaxial cable 31 employed in this embodiment, which is used for the first and second coils 19, 23, although alternative transmission lines can be used. As will be appreciated, a coaxial cable comprises an inner conductor 33, surrounded by a tubular insulating layer, surrounded by a tubular conducting shield 35. Figure 7b shows the cable 31 along part of its axial length. The gap "g" between the outer surface of the core 33 and the inner surface of the outer shield 35 is substantially constant throughout the length, this being the inter-winding gap. The inner conductor 33 in this case provides the first coil 19 and the shield 35 the second coil 23.

Figure 8a and 8b show the cross-sectional areas of a twin transmission line which is an additional example of what can be used in the construction of the coils for TLT 1 and the relationship of the respective gap.

Referring to Figure 9, an example of how the coaxial cable which can be used in the Figure 4 embodiment is physically arranged around a core 41, as well the ports. In this case, a cylindrical core 41 is shown in part, although a toroidal core can be employed. The inter-winding gap g between the conductors is maintained constant throughout the entire length of the coil around the core, as is intra-winding gap G. Referring back to Figures 5 and 6, as a result of this physical arrangement, the stray inter and intra -winding capacitances Cg and CG are constant and distributed. The inter-winding stray capacitance Cg is subsumed into the transmission line formed by the two coils (Figure 4) 19, 23 and is inversely proportional to the inter-winding gap g. The intra-winding stray capacitance CG in this structure is inversely proportional to the intra- winding gap G. Increasing this gap G has the effect of increasing the upper frequency limit and therefore the bandwidth. In some embodiments, the conductors of the coils (Figure 4) 19, 23 are of constant cross-section and therefore of constant surface area. In some embodiments, the dimensions of the core are also relevant, in that the bandwidth can be controlled by changing the dimensions; reducing one or both of the core diameter and/or length. This has the effect of decreasing the lower frequency (OCL) . The material of the core is also relevant, in that we employ a ferrite core with selected permeability, for example 10000 μ, as will be discussed later on.

In some embodiments, the length and the construction of the winding can also be used to control bandwidth, in that the shorter the length of the winding, the higher the usable upper frequency (fU) . Overall, therefore, there is an incentive to miniaturise . Returning to the specific embodiment shown schematically in Figure 4, using this 1:1 topology, employed physically using a 1.2 metre length of RG179 50 ohm coaxial cable, with the abovementioned constant inter and intra gap spacing wound around the core, a 5.1 mH magnetising inductance was recorded. It was also observed through measurement that there was no upper frequency limit observed or at least a very high upper frequency limit using the particular test signal.

It was also observed that this embodiment, demonstrated a substantially constant characteristic impedance Zo of 100 ohms and a transit delay of 6 nS, independent of frequency above the low frequency cut-off fl, which was 1.5kHz.

This result is not consistent with traditional Isolating Transformers and TLT models. Indeed, applying the numerical parameters to traditional distributed parameter models gave a predicted upper frequency limit in the order of l/(2x6nS) of 83 MHz. However, with this embodiment, no such upper limit was observed. Figure 4 provides in schematic form a model more consistent with these findings, indicating a way of designing and constructing an ITLT for seamless connection between a source and transmission line to provide greater bandwidth. Further, by cascading multiple transmission lines using such ITLTs and a shunt magnetising inductance provides an increase in the magnitude of (fU) in comparison to well-known and current predictive models.

Reflections captured from the input port (Port 1) were found to indicate a constant resistive characteristic impedance and a constant transport delay (time delay) in much the same way as a transmission cable does. In the embodiment shown in Figure 4, the characteristic impedance at both ports was found to be twice that of the characteristic impedance Zo of the transmission line used to form the Isolating TLT, using the 1:1 topology. So, in this case, 100 ohms characteristic impedance was presented at both outputs, making this Isolating TLT suitable for connection to a 100 ohm data source and receiver 3 and 100 ohm transmission line 5, with the resultant matching being maintained over the wide bandwidth.

It was deduced that the TLT (d.c. isolation aside) could be accurately modelled by a shunt inductance, i.e. the magnetising inductance of the core, in series with the transmission line segments (L-section, T-section and/or Pi-section models would work in this regard) . As such, it is possible to construct a TLT for d.c. isolation that offers very wide bandwidth, with a substantial increase in fU which in itself appears to be limited only by the transmission line loss itself.

This embodiment, as mentioned, provides a substantially constant and resistive characteristic impedance at Ports 1 and 2. The leakage inductance of a conventional isolating transformer and TLT is modelled as a lumped element inductance that is not inductively coupled to anything else and which appears in series with the 100% coupled mutual inductances of the conventional isolating transformer and TLT. In the present embodiment, however, indications are that whilst there are still leakage inductances, these do not appear (when modelled) as a single lumped element at the ports, but are distributed. They appear, or are modelled, as a series of small incremental inductances, not coupled to anything else, and distributed between incremental spaced elements of mutual inductance and incremental spaced elements of inter-winding capacitance. This model results in a ladder network of series inductances (Ls) in the two legs of the windings linked by shunt capacitive elements interspersed with mutually spaced inductive elements. This ladder network can be recognised as being identical, or substantially identical, to the incremental lumped element model of an actual transmission line, with unsurprisingly the same properties in common therewith, namely a characteristic impedance that is constant and a transmission term that is substantially a constant propagation delay. In summary, this embodiment has taken the lumped parasitic leakage inductance (L) and the inter winding capacitance (C) of traditionally constructed isolating transformers / TLTs with primary and secondary coils wound on a core) and distributed these as the distributed L and C of a transmission line with characteristic impedance SQRT (L/C) by winding the primary and secondary coils together as a transmission line.

In terms of a specific design using Figure 4 topology, therefore, being 1:1, the choice of transmission line with which to construct the Isolating TLT should have a characteristic impedance half that of the impedances required at the ports, i.e. those of the data source and receiver 3 and the transmission line 5. The resulting matching remains flat over a wide frequency band, as does the observed transmission delay. The only observed significant component of the reflections induced at the ports are due to the intrinsic shunt magnetising impedance of the Isolating TLT. However, these reflections due to parasitic leakage inductance and the inter-winding capacitance of a traditional (non-TLT) isolating transformer have been substantially, or completely, subsumed into the constant resistive characteristic impedance and transmission delay of this ITLT. The notable result of this is the substantial increase in upper frequency / bandwidth, limited only by the loss of the transmission cable 5 it is connected to, the bandwidth of the circuits and other logic components it is being integrated with, and the shunt magnetising impedance of the Isolating TLT. The factor of the relationships between characteristic impedance at the ports, and that of the constituent transmission line of the 1:1 ITLT also means that using two transmission lines of characteristic impedance Zo, connected in parallel, can provide an overall composite Isolating TLT with a characteristic impedance substantially equal to Zo at the ports. This is of benefit in that transmission lines with commonly available characteristic impedances (e.g. 50 ohm) can be used between systems requiring the same impedance, e.g. 50 ohm, notwithstanding the aforementioned relationship. So, by connecting two 1:1 Isolating TLTs (as depicted in Figure 4) in parallel, to provide a composite Isolating TLT, the use of 50 ohm transmission line for the Isolating TLTs will provide 50 ohms at the first and second ports. More than two parallel Isolating TLTs can be used for similar purposes, to provide the required impedances at the ports. More than two ports can also be provided, where required. To recap, (fL) is maintained by the shunt magnetising impedance, which is inversely proportional to the intrinsic magnetising inductance. This magnetising inductance increases with the increasing inductance factor of the core, and as the square of the number of turns. The upper frequency limit due to the shunt magnetising impedance is due in turn to (parasitic) intra- winding capacitances of the coils, distinct from the inter- winding capacitance between coils. The upper frequency limit is inversely proportional to the intra-winding capacitance. The intra-winding capacitance can be beneficially reduced, further increasing the upper frequency limit (fU) by reducing the length and diameter of the constituent transmission line from which the embodiment is constructed. This, taken together, means that miniaturisation of the embodiment is effectively increasing the upper frequency limit without further increasing the lower frequency limit to the extent that the magnetising inductance can be maintained during miniaturisation, e.g. by keeping the number of turns constant while maintaining the reluctance of the core constant, i.e. for a give core material, maintaining the ratio of magnetic path cross-section and length. This process is constrained only by the need to avoid excessive loss, e.g. Cu loss of thin conductors, and the power handling capability of the ITLT as the ITLT will need to be of a certain minimum size in order to handle a given amount of power without distortion and/or destruction.

Figures 10 and 11 provide a more generalised comparison between the topologies of the known and present embodiment transformers, as previously introduced in relation to Figures 3 and 4 respectively, although using only single windings for each wire for reasons to be explained. Of note is that in the known, Figure 10(a) and 11(a) embodiment, the characteristic impedance is not constant, and bandwidth is limited. The Figure 10(b) and 11(b) topology indicates a significant attribute of the present embodiment, which is that there are two ports which are, mechanically and topologically,_opposite. This produces a constant resistive impedance and increased bandwidth. Referring to Figure 12, a graphical indication of the voltage versus time response for the known Figure 3 / 11(a) transformer is shown, in which Zc is the characteristic impedance of the transmission line, e.g. 100 ohms, and Zx is the characteristic impedance of the transformer. OC and SC represent Open Circuit and Short Circuit conditions respectively. As Figure 12 shows, the Figure 3 (and 11(a)) embodiment has a different termination point that results in a significant reflection that causes a change in the impedance thus limiting the bandwidth of the transformer .

Referring to Figures 13(a) and (b) , the response for the Figure 4 / 11(b) transformer is shown.. Referring to Figure 13(a), the termination point is different, and although X shows some ambiguity between transformer and transmission line, for presentation purposes only the net result of the Figure 4 / 11(b) topology is shown in Figure 13(b) which is a substantially seamless transmission line transformer.

For optimal performance, in further embodiments, as well as having the ports at opposite ends, mechanically speaking, a single turn or winding is employed, which it has been discovered, takes the upper frequency beyond 2 GHz and beyond 10 GHz.

Figures 14(a) and 14(b) shows such an embodiment 61 of the invention, employing a. pair of conductors 64, 65 wound around the central part 63 of a ferrite pot core 62, each conductor extending between mechanically opposite ports 1 and 2, and executed using a single turn or winding, following the Figure 4 / 11(b) topology. There is no intra winding capacitance, and it does not limit low / high bandwidth combinations. The conductors are insulated from one another, and preferably have a substantially constant gap. In a preferred embodiment, the pot core 62 has a diameter of approximately 12.5 mm and the diameter of the central part 63 has a bore of approximately 0.2 mm. The permeability of the ferrite material is approximately ΙΟ,ΟΟΟμ. This embodiment exhibits under testing an open circuit inductance (OCL) of 160 μΗ and a bandwidth of 10 GHz. Variations of one or more of these parameters may provide higher bandwidths.

Referring now to Figures 15 to 17, alternative practical embodiments of the above are shown and described in terms of how they may be manufactured and produced.

Referring to Figure 15, a top view of such a transformer 70 is shown. It comprises a binocular (or bead) core 71 with two parallel bores 74, 75 through which twisted conductors 73, 76 pass to provide a transmission line. The core can actually be toroidal, binocular or a pot, but a binocular core provides a natural fit for the present embodiment (s) .

A first port (Port 1) is provided to one side of the core 71, and comprises a first conductor 73 which runs from one port terminal, through the first bore 74, whereafter it exits and returns back through the second bore 75 and terminates at the other port terminal. A second port (Port 2) is provided on the mechanically opposite side to the core 71, and comprises a second conductor 76 which runs from one port terminal, through the second bore, whereafter it exits and returns back through the first bore 74 and terminates at the other port terminal. The conductors 73, 76 therefore execute a single turn or winding, as with the previous embodiment, which is found to exhibit particularly advantageous results. Conductors 73 and 76 are twisted together within the core 71 as shown, but are insulated from one another by surrounding insulating material and have a substantially constant gap.

Effectively, each conductor 73, 76 is a U-shaped arrangement pulled from opposite ends through the core 71.

Figure 16 shows the Figure 15 arrangement in perspective view.

In one example, the Zc at Port 1 and Port 2 is 100 ohms, in which case the transmission line is arranged to be Zc/2 = 50 ohms . Other example sizes with additional Common Mode Coupling (CMC) are given as follows.

To achieve 100 kHz at 37.5 mA / 15000 μί for an OCL 350 μΗ, the dimensions would be Outer Diameter (OD) of 4 mm, Inner Diameter (ID) of 0.5 mm and length of 38 mm. For four lanes, this equates to a package size of 20 mm x 45 mm x 6 mm.

To achieve 100 kHz at 8 mA / 15000 μί for an OCL 120 μΗ, the dimensions would be OD of 4 mm, ID of 0.5 mm and length of 12 mm. For four lanes, this equates to a package size of 20 mm x 20 mm x 6 mm.

Figure 17 is an alternative construction 80, in which, effectively, the binocular core is divided into two parts 81a, 81b, but has the same general dimensions overall. In this case, the ports 1 and 2 are still mechanically opposed, but are between the two core parts 81a, 81b. More specifically, a first port (Port 1) is provided two one side of the core parts 81a, 81b, generally at the gap between the two, and comprises a first conductor 83 which runs from one port terminal, through the first bore 85a, whereafter it exits at one end and returns back through the second bore 84a, through to the other second bore 84b, exiting at the other end and returning back through the other first bore 85b and terminating at the other port terminal. The second port (Port 2) is provided on the opposite side of the core parts 81a, 81b, again generally at the gap between the two. A second conductor 86 runs from one port terminal, through the second bore 84a, whereafter it exits at one end and returns back through the first bore 85a, through to the other first bore 85b, exiting at the other end and returning back through the other second bore 84b and terminating at the other port terminal. Conductors 83, 86 and 76 are twisted together within the core parts 81a, 81b, as shown, but are insulated from one another by surrounding insulating material and may have a substantially constant gap.

Analysis by simulation of the Figure 17 embodiment shows that it doubles the parasitic resonance than with the Figure 15 and 16 example. A 20 mm single bead construction has a 6 to 7 GHz resonance, whereas two 10 mm beads, as in Figure 17, result in a resonance of 12 - 14 GHz. Either structures meet all the backward compatibility requirements of historic systems as well as evolving 40 GBase-T and 100 GBase-T standards, as would using the above toroidal or pot core construction. A pot core geometry is free of this resonance, and a bead geometry that accepts wire loops which is as wide as is long substantially supresses this parasitic mode, being similar or equivalent to a square pot core . In a preferred embodiment of the Figures 15 to 17 embodiments, the pot core 71, 81 has a length of approximately 15 mm and the diameter of the central bores 74, 75, 84, 85 is approximately 0.2 - 0.5 mm. The permeability of the ferrite material is approximately ΙΟ,ΟΟΟμ. These embodiments exhibit under testing an open circuit inductance (OCL) of 160 μΗ and a bandwidth of 10 GHz and beyond. Variations of one or more of these parameters may provide higher bandwidths .

The construction exhibits the aforementioned advantageous effects, making it particularly suited to wide bandwidth data transmission. For example, high bandwidth operation well beyond 2 GHz has been demonstrated, with insertion losses within the - 3dB standard. The use of only a single turn or winding for each conductor extends the upper frequency limit. Any worsening of the open circuit inductance (OCL) can be counteracted by, for example, dimensional changes to the core (e.g. the bore) and/or the permeability of the core material.

It will be appreciated that the above described embodiments are purely illustrative and are not limiting on the scope of the invention. Other variations and modifications will be apparent to persons skilled in the art upon reading the present application.

Moreover, the disclosure of the present application should be understood to include any novel features or any novel combination of features either explicitly or implicitly disclosed herein or any generalization thereof and during the prosecution of the present application or of any application derived therefrom, new claims may be formulated to cover any such features and/or combination of such features.