ISOLATED DC-DC CONVERTER

TECHNICAL FIELD

[0001] The aspects of the disclosed embodiments relate generally to power conversion apparatus, and more particularly to impedance networks used in multi-port DC-DC power converters.

BACKGROUND

[0002] Three port DC-DC power converters capable of providing multiple variable DC voltage sources are useful in many modern power applications. These can include for example, electric vehicles, electric aircraft, and micro-grid deployments used in wind and solar energy systems.

[0003] A three port DC-DC power converter can be formed using a two-stage approach. In these two stage solutions, three DC voltage sources are integrated together using two conventional DC-DC converters. Unfortunately, these solutions result in an additional switching network not present in other three-port solutions. Two stage solutions also fail to provide a direct path between all the DC sources.

[0004] Three-port DC-DC converters have been proposed based on an integrated three winding transformer and three inductive impedances. Such an approach requires complex control to decouple the ports. A change in power on one can cause disturbances on both the other ports. Controlling these disturbances can be quite complex because of coupling among the ports. Design of the integrated transformer is also complex due to interaction between the multiple windings with their magnetic flux and leakage currents coupled together.

[0005] Using two transformers rather than a more complex three winding transformer can simplify the transformer design, however prior solutions based on this approach rely on asymmetrical switching networks with one network having an increased number of switches. These solutions also require a complex control strategy to mitigate coupling between the ports. Furthermore, due to the use of purely inductive impedances, the drawbacks associated with symmetrical impedances further increases control complexity.

[0006] Thus, there is a need for three port DC-DC power converters that employ simple switching networks while allowing for simplified transformer designs and simplified control strategies. Accordingly, it would be desirable to provide an apparatus that addresses at least some of the problems described above.

SUMMARY

[0007] The aspects of the disclosed embodiments are directed to a quasi-decoupled three port DC-DC converter apparatus appropriate for applications requiring multiple DC power sources. The aspects of the disclosed embodiments provide decoupled three port DC power from a converter employing two-level decoupling strategies. This and other objectives are addressed by the subject matter of the independent claim. Further advantageous modifications can be found in the dependent claims.

[0008] According to a first aspect, the above and further objectives and advantages are obtained by an apparatus. In one embodiment, the apparatus includes a first transformer and a second transformer, where a first winding of the first transformer is coupled in parallel with a first winding of the second transformer. The apparatus includes a first switching network having a first DC power rail and a first AC power source, where the first AC power source is coupled to the first winding of the first transformer through a first impedance, and the first DC power rail is configured to couple with a first DC power source. The apparatus further includes a second switching network having a second DC power rail and a second AC power source, where the second AC power source is coupled to a second winding of the first transformer through a second impedance, and the second DC power rail is configured to couple with a second DC power source. The apparatus includes a third switching network having a third DC power rail and a third AC power source, where the third AC power source is coupled to a second winding of the second transformer through a third impedance, and the third DC power rail is configured to couple with a third DC power source. The first impedance comprises a purely inductive impedance, the second impedance comprises a zero impedance, and the third impedance comprises a series resonant impedance. The use of both circuit level and control level decoupling leverages the available control parameters to provide simplified optimal control of all three ports.

[0009] In a first possible implementation form of the apparatus according to the first aspect the first impedance comprises the zero impedance, the second impedance comprises the purely inductive impedance, and the third impedance comprises the series resonant impedance. Changing the circuit level architecture provides flexibility with regard to power transfer among the three ports.

[0010] In a possible implementation form of the apparatus, the first impedance comprises the series resonant impedance, the second impedance comprises the purely inductive impedance, and the third impedance comprises the zero impedance. A third circuit level architecture provides additional flexibility for power distribution among the three ports.

[0011] In a possible implementation form of the apparatus, the series resonant impedance comprises an inductor in series with a capacitor. A simple two component implementation of the series resonant impedance can reduce converter complexity and reduce cost.

[0012] In a possible implementation form of the apparatus, the first switching network, the second switching network, and the third switching network each comprise a full bridge switching network. Full bridge switching networks offer advantages in many applications and can be advantageously employed with any of the claimed high frequency impedance networks.

[0013] In a possible implementation form of the apparatus, the first switching network, the second switching network, and the third switching network each comprise a half bridge switching network. In certain embodiments a half bridge switching network can be desirable and can be advantageously employed with any of the claimed high frequency impedance networks.

[0014] In a possible implementation form of the apparatus, where the first switching network, the second switching network, and the third switching network are operated based on a dual active bridge switching strategy. The apparatus is configured to have a first internal phase shift of the first switching network set to zero when the first impedance comprises a zero impedance, a second internal phase shift of the second switching network set to zero when the second impedance comprises a zero impedance, and a third internal phase shift of the third switching network set to zero when the third impedance comprises a zero impedance. Employing control level decoupling along with the circuit level decoupling improves controllability of the converter. [0015] In a possible implementation form of the apparatus, the first switching network, the second switching network, and the third switching network are operated at a first switching frequency when a first power transfer rate is desired, the first switching network, the second switching network and the third switching network are operated at a second switching frequency when a second power transfer rate is desired, where the first switching frequency is different than the second switching frequency. The use of asymmetric impedances allows the switching frequency to be advantageously employed as an additional control parameter leading to improved controllability of the converter.

[0016] These and other aspects, implementation forms, and advantages of the exemplary embodiments will become apparent from the embodiments described herein considered in conjunction with the accompanying drawings. It is to be understood, however, that the description and drawings are designed solely for purposes of illustration and not as a definition of the limits of the disclosed invention, for which reference should be made to the appended claims. Additional aspects and advantages of the invention will be set forth in the description that follows, and in part will be obvious from the description, or may be learned by practice of the invention. Moreover, the aspects and advantages of the invention may be realized and obtained by means of the instrumentalities and combinations particularly pointed out in the appended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

[0017] In the following detailed portion of the present disclosure, the invention will be explained in more detail with reference to the example embodiments shown in the drawings, in which like references indicate like elements and: [0018] Figure 1 illustrates a block diagram of an exemplary three port DC-DC converter incorporating aspects of the disclosed embodiments;

[0019] Figure 2 illustrates a block diagram of an exemplary quasi-decoupled three port DC-DC converter incorporating aspects of the disclosed embodiments; [0020] Figure 3 illustrates exemplary circuit diagrams showing equivalent star and delta transformations of a high frequency impedance network incorporating aspects of the disclosed embodiments;

[0021] Figure 4 illustrates a block diagram of an exemplary three-port DC-DC converter having an asymmetric impedance network incorporating aspects of the disclosed embodiments; [0022] Figure 5 illustrates exemplary circuit diagrams showing equivalent star and delta transformations of a high frequency impedance network incorporating aspects of the disclosed embodiments;

[0023] Figure 6 illustrates circuit diagrams showing equivalent star and delta transformations of a high frequency impedance network incorporating aspects of the disclosed embodiments;

[0024] Figure 7 illustrates a schematic diagram of an exemplary full bridge switching network incorporating aspects of the disclosed embodiments;

[0025] Figure 8 illustrates a schematic diagram of an exemplary half bridge switching network incorporating aspects of the disclosed embodiments; [0026] Figure 9 illustrates a schematic diagram of a three-port DC-DC converter topology having full bridge switching networks incorporating aspects of the disclosed embodiments;

[0027] Figure 10 illustrates graphs showing operating waveforms of a three-port DC-DC converter incorporating aspects of the disclosed embodiments;

[0028] Figure 11 illustrates graphs showing operating waveforms of a three-port DC-DC converter with circuit control level decoupling incorporating aspects of the disclosed embodiments.

[0029] Figure 12 illustrates an exemplary embodiment of a three-port DC-DC converter apparatus incorporating aspects of the disclosed embodiments.

DETAILED DESCRIPTION OF THE DISCLOSED EMBODIMENTS

[0030] Referring to Figure 1, a simplified block diagram of a three-port DC-DC converter apparatus 100 is illustrated. The apparatus 100 of the disclosed embodiments is directed to a three- port DC-DC converter topology which may be referred to more particularly herein as a quasidecoupled three-port DC-DC converter apparatus. The apparatus 100 employs a high frequency impedance network 120 having two transformers Ti, T2 and three asymmetric impedances, Z ₁ , Z ₂ , and Z ₃ configured to decouple the three ports 102, 104, 106, thereby providing simplified control of power at each of the three ports 102, 104, 106. The apparatus 100 is suitable for use in many modern power applications requiring three variable DC power sources such as electric vehicles, electric aircraft, and micro-grid deployments.

[0031] As shown in Figure 1, the high frequency impedance network 120 is coupled to the three ports 102, 104, 106. Each port 102, 104, 106 includes a variable DC voltage or power source Vi, V2, V3. Each port 102, 104, 106 includes a switching network 108, 110, 112 coupled to a DC power rail 114, 116, 118 and an AC power source VACI, V _AC2 , VAC3. When operated appropriately the switching networks 108, 110, 112 are configured to convert between the DC power sources Vi, V2, V3, which are coupled to the DC power rails 114, 116, 118 respectively, and corresponding AC power sources VACI, V _AC2 , VAC3. AS will be discussed further below, any appropriate switching network capable of converting between a DC power and an AC power may be advantageously employed as the switching networks 108, 110, 112. The switching networks 108, 110, 112 need not all be the same. In certain embodiments different switching networks, such as half bridge and full bridge switching networks, may be advantageously employed in the different ports 102, 104 and 106 of the apparatus 100.

[0032] As used herein the terms “power source” and “power” are used interchangeably to refer to power or device capable of sourcing and/or sinking electric power. For example, a battery may be used as a DC power source or DC power that is capable of sourcing power, such as for driving a motor or other electric device. A battery is also capable of acting as a power sink when receiving and storing power during charging. Those skilled in the art will readily recognize that each of the three ports 102, 104, 106 may be configured to receive power from the DC power sources Vi, V2, V3 as well as send power to the DC power sources Vi, V2, V3.

[0033] The high frequency impedance network 120 couples the three ports 102, 104, 106 and provides electrical isolation among the ports 102, 104, 106. Electric isolation is provided by a pair of transformers Ti, T2, where a first winding 122 of the first transformer Ti is coupled in parallel with a first winding 126 of the second transformer T2. Certain conventional three-port converters employ a single transformer having three windings. As is known in the art, design of a three winding transformer can become very complex and lead to inefficiencies in the design and operation of conventional converters. This disadvantage is overcome through the use of the pair of two winding transformers T ₁ , T ₂ , thereby significantly simplifying design of the transformers. [0034] The first winding 122 of the first transformer T1 and the first winding 126 of the second transformer T ₂ are coupled with the AC power source V _AC1 of the first port 102, through a 5 first impedance Z ₁ . A second winding 124 of the first transformer T ₁ is coupled through a second impedance Z ₂ with the AC power source V _AC2 of the second port 104, and a second winding 128 of the second transformer T ₂ is coupled through a third impedance Z ₃ with the AC power source VAC3 of the third port 106. The transformers T1, T2 may be configured to incorporate any appropriate turns ratio configured to support the desired DC power sources V ₁ , V ₂ , V ₃ of each port 10 102, 104, 106. [0035] An important aspect of the disclosed embodiments is selection of impedances Z ₁ , Z ₂ , Z ₃ used to connect the transformers T ₁ , T ₂ with the switching networks 108, 110, 112. It has been shown that using the same type of impedance, such as a pure inductance, for all three impedances Z ₁ , Z ₂ , Z ₃ , results in an undesirable coupling among the ports where a change in power 15 on one port results in a power disturbance on the other two ports. Further, use of the same type of impedance for all three impedances Z ₁ , Z ₂ , Z ₃ , referred to herein as symmetrical impedances, makes it difficult to independently control each of the ports, especially when different ports are configured to deliver different power levels. [0036] In contrast with conventional solutions, such as solutions employing symmetrical 20 impedances, the exemplary apparatus 100 is configured with asymmetric impedances Z ₁ , Z ₂ , Z ₃ . Each impedance Z ₁ , Z ₂ , and Z ₃ may be implemented using a different type of impedance network and may include any desired combination of inductors and capacitors. For example, in one embodiment the first impedance Z ₁ may be a pure inductance, the second impedance Z ₂ may be a zero impedance and the third impedance Z ₃ may be a series resonant impedance. The resulting asymmetrical impedance networks, Z ₁ , Z ₂ , and Z ₃ exhibit different characteristics and may be selected to reduce coupling among the ports 102, 104, 106. An asymmetrical choice of impedances Z ₁ , Z ₂ , and Z ₃ can offer more freedom, aid decoupling, and help reduce control complexity.

[0037] As used herein, the term zero impedance, or setting an impedance to zero, Z=0, refers to a direct connection or coupling of elements where the coupling has a negligible amount of inductance, and capacitance.

[0038] As used herein the term asymmetrical impedances refers to impedances of different types having differing frequency characteristics. For example, a pure inductance, a series resonant impedance, and a zero impedance are examples of different types of impedances. A one henry inductor and a two henry inductor differ only in value and have similar frequency characteristics. Thus, a one henry inductor and a two henry inductor are not considered to be different types of impedances and do not constitute asymmetric impedances.

[0039] Figure 2 illustrates a simplified block diagram of an exemplary quasi-decoupled three port DC-DC converter apparatus 200 incorporating aspects of the disclosed embodiments. The apparatus 200 illustrated in Figure 2 is similar to the apparatus 100 described above and with reference to Figure 1 where like references indicate like elements. As will be discussed in more detail below, setting one of the three impedances Z ₁ , Z ₂ , Z ₃ in the high frequency impedance network 220, such as the second impedance Z ₂ , to zero, effectively decouples the corresponding port 104 from the other two ports, 102 and 106. In the exemplary apparatus 200 the second impedance Z ₂ is set to a zero impedance (Z ₂ =0), while the first impedance Z ₁ is set to a pure inductance formed by the inductor L202, and the third impedance Z ₃ is set to a series resonant impedance formed by an inductor L204 in series with a capacitor C206. As will be discussed further below, the use of different impedance types, such as zero impedance, purely inductive impedance , and series resonant impedance, for the impedances Z ₁ , Z ₂ , Z ₃ of apparatus 200 illustrate a high frequency impedance network 220 having asymmetric impedances Z ₁ , Z ₂ , Z ₃ .

[0040] The asymmetric impedances, namely the zero impedance; pure inductance; and series resonant impedance, may when desired be shifted to or implemented in different ports of the network 220. For example, in one embodiment the first impedance Z ₁ may be set to the zero impedance, the second impedance Z ₂ may be set to the series resonant impedance, and the third impedance Z ₃ may be set to the pure inductance. Alternatively, the first impedance Z ₁ may be set to the series resonant impedance, the second impedance Z ₂ may be set to the pure inductance and the third impedance Z ₃ may be set to the zero impedance.

[0041] In the illustrated embodiment of Figure 2 the pure inductance is represented as a single inductor L202. In alternate embodiments, any inductive network that results in an equivalent pure inductance may be advantageously employed without straying from the spirit and scope of the present disclosure. Similarly, the series resonant impedance is illustrated as a single inductor L204 coupled in series with a single capacitor C206. In alternate embodiments, any series resonant impedance network may be advantageously employed without straying from the spirit and scope of the present disclosure.

[0042] Figure 3 illustrates circuit diagrams 302, 304, showing equivalent star 306 and delta 308 transformations of the high frequency impedance network 120 described above and with reference to the apparatus 100. Elements shown in Figure 3 are similar to those shown in Figure 1 where like references represent like elements. In the equivalent circuit diagrams 302 and 304, the impedances Z ₂ ' and Z ₃ ' represent equivalent impedance values of the impedances Z ₂ and Z ₃ respectively, after being reflected across their respective transformers T ₁ and T ₂ . The first circuit diagram 302 depicts a star 306 equivalent transformation of the high frequency impedance network 120 coupling the three ports 102, 104, 106. The second circuit diagram 304 depicts a delta 308 equivalent transformation of the high frequency impedance network 120 coupling the three ports 102, 104, and 106.

[0043] Examining the star 302 and delta 304 equivalent circuits depicted in Figure 3 provides an appreciation of the coupling among the three ports 102, 104, 108 resulting from having three non-zero impedances in the high frequency impedance network 120. This coupling among the ports 102, 104, 106 is highlighted by the equivalent delta transformation 308 where each of the three equivalent impedances Z ₁₂ ', Z ₁₃ ', Z ₂₃ ' includes a combination of the actual impedances Z ₁ , Z ₂ , Z ₃ . The notation Z ₁₂ ' represents an equivalent impedance derived from the first Z ₁ , second Z ₂ , and third Z ₃ impedance, and the prime indicates that the impedance is reflected across a transformer where appropriate.

[0044] The equivalent delta transformation 304 shows that a change in power on one of the three ports, such as the first port 102, will be reflected on the other two ports, such as ports 104 and 106, because all ports share multiple and possibly asymmetrical impedances, Z ₁ , Z ₂ ', and Z ₃ '. Decoupling of the ports 102, 104, 106, under these circumstances requires complex control strategies.

[0045] Adverse effects created by coupling among the ports 102, 104, 106 can be reduced by setting one of the three impedances Z ₁ , Z ₂ , Z ₃ to a zero value. Figure 4 illustrates a simplified block diagram of a three-port DC-DC converter 400 where one of the impedances in the high frequency impedance network 420 is set to zero. In this example, impedance Z ₃ =0. The apparatus 400 is similar to the apparatus 100 described above and with reference to Figure 1 where like references indicate like elements.

[0046] Figure 5 illustrates circuit diagrams 502, 504 showing equivalent star and delta transformations, respectively, of the high frequency impedance network 420 described above. Configuring the high frequency impedance network 420 with the third impedance Z ₃ set to zero, Z ₃ =0, effectively decouples the first port 102 and the second port 104. As described above, the impedance Z ₂ ' represents the value of impedance Z ₂ reflected across the transformer Ti.

[0047] The delta transformation 504 shows that by setting the third impedance Z ₃ to zero, linkage between the first port 102 and the second port 104 has been broken and the first port 102 and second port 104 are completely decoupled. The delta transformation 504 shows that when transferring power between the first port 102 and second port 104 of the apparatus 400, the third port 106 acts as a bridge or buffer. Power transfer between the first port 102 and the third port 106, or between the second port 104 and the third port 106 by the apparatus 400, can be performed directly using the first impedance Z ₁ and the second impedance Z ₂ respectively.

[0048] Any one of the impedances Z ₁ , Z ₂ , Z ₃ can be set to zero as desired to aid in decoupling of the ports. As described above, setting the third impedance to zero, Z ₃ =0, decouples the first port 102 from the second port 104. Similarly, setting the first impedance to zero, Z ₁ =0, decouples the second port 104 and the third port 106.

[0049] An important consideration affecting performance of the above described three- port DC-DC converters is the choice of impedances used for each of the three impedances Z ₁ , Z ₂ , Z ₃ . A conventional choice employs symmetric impedances where all three impedances are purely inductive impedances. When operating the switching networks as dual active bridge (DAB) converters, the use of symmetric impedances requires complex control schemes adding cost and making optimal control difficult to attain. In contrast, the embodiments disclosed herein employ asymmetrical impedances which, as will be described further below, exhibit different characteristics allowing significantly simplified control strategies.

[0050] The benefits provided by asymmetrical impedances can be understood from the equivalent delta transformation 504 described above and with reference to Figure 5. As discussed above, coupling between the first port 102 and the second port 104 can be removed by setting the third impedance Z ₃ to zero. The third port 106, which is acting as abridge or buffer, is still coupled to each of the other ports 102, 104. Because of this coupling, a change in power on the third port 106 results in a power disturbance at both the first port 102 and the second port 104. If both the first impedance Z ₁ and the second impedance Z ₂ are the same, such as purely inductive, the disturbance transfer to the first port 102 and the second port 104 will have the same rate, making it difficult to control each of the ports independently.

[0051] The above described control difficulty can be mitigated by using asymmetric impedances. For example, as shown in the embodiment illustrated in Figure 4, when the third impedance Zsis set to zero, the first impedance Z ₁ and the second impedance Z ₂ can be set to different types of impedances, such as a pure inductance and a series resonant impedance. In an embodiment having asymmetric impedances for the first impedance Z ₁ and the second impedance Z ₂ , an appropriate choice of impedances Z ₁ and Z ₂ will provide different power-transfer characteristics to each of the ports. These different power transfer characteristics allow different rates of power transfer among the ports. Having differing transfer rates simplifies circuit control level decoupling of the ports.

[0052] To illustrate this, consider a case where the apparatus 400 is transferring a low power from the third port 106 to the first port 102 and a high power from the third port 106 to the second port 104 simultaneously. With symmetric impedances it is difficult for the third port 106 to accommodate sending different power levels to each of the other ports 102, 104. In contrast, the differing power transfer characteristics provided through the use of asymmetric impedances can simplify delivery of different power levels to each of the first 102 and second 104 ports.

[0053] In general, many combinations of impedances can be formed using inductors and capacitors. In the simplest case, all three inductances Z ₁ , Z ₂ , Z ₃ can be pure inductances. Unfortunately, as can be seen from conventional DAB converters based on pure inductances, complex phase-shift modulation methods are required to achieve efficient power conversion.

[0054] A second order impedance network can be formed as a series resonant network formed using an inductance and a capacitance in series. Coupling two switching networks with a series resonant impedance yields a converter topology known as dual-active-bridge series-resonant DC-DC converter (DABSRC). The DABSRC requires both frequency variation and phase-shift modulation to converge for an efficient power conversion. Replacing all three impedances Z ₁ , Z ₂ , Z ₃ with a series resonant impedance leads to a symmetrical DABSRC. However, because all switching networks 108, 110, 112 must operate at the same switching frequency in a three-port converter, having symmetric series resonant impedances creates a difficult power decoupling problem due to having the same power transfer characteristics among all the ports. [0055] For example, the use of symmetrical series resonant impedances in the example described above with respect to apparatus 400, delivering full power to one port and low power to the other port requires operating each switching network 108, 110, 112 at different frequencies. Unfortunately, it is not possible to operate different ports at different switching frequencies in a three-port DC-DC converter.

[0056] As described briefly above, control difficulties caused by having similar power transfer characteristics among all the ports can be mitigated through the use of asymmetric impedances. As an aid to understanding consider the implementation form of the apparatus 200 described above and with reference to Figure 2. The apparatus 200 decouples the first port 102 from the third port 106 by setting the second impedance Z ₂ to zero, Z ₂ =0. Implementing the first impedance Z ₁ using a pure inductance L202, forms a DAB converter between the first port 102 and the second port 104, while implementing the third impedance Z ₃ as a series resonant impedance formed by L204, C206, forms a DAB SRC between the second port 104 and the third port 106.

[0057] Forming two different types of converters, DAB and DAB SRC, each having different power transfer characteristics, provides the freedom to control the two types of converters independently. This independent control capability is due in part to the enhanced decoupling provided by the above described choice of impedances Z ₁ , Z ₂ , Z ₃ . The apparatus 200 can be viewed as a quasi-decoupled three-port DC-DC converter, where a zero impedance is used to decouple two of the ports, and asymmetric impedances are used to allow independent control.

[0058] Figure 6 illustrates circuit diagrams 602, 604 showing equivalent star and delta transformations, respectively, of the high frequency impedance network 220 described above and with reference to the apparatus 200. Elements shown in Figure 6 are similar to those shown in Figure 2 where like references represent like elements. In the equivalent circuit diagrams 602 and 604, the impedances Z ₃ ', L’204, C’206 represent equivalent impedance values of the impedances Z ₃ , L204, C206 shown in Figure 2 respectively, after being reflected across the transformer T2.

[0059] The first circuit diagram 602 depicts a star equivalent transformation of the high frequency impedance network 220 coupling the three ports 102, 104, 106. The second circuit diagram 604 depicts a delta equivalent transformation of the high frequency impedance network 220 coupling the three ports 102, 104, and 106.

[0060] The delta transformation 604 shows that setting the second impedance to zero, Z ₂ =0, decouples the first port 102 from the third port 106. As described above setting the second impedance to zero, Z ₂ =0. simplifies control of the DC power sources Vi, V2, V3.

[0061] The use of asymmetric impedances for the first impedance Z ₁ and the second impedance Z ₂ in the apparatus 200 can be viewed as forming two different converter types, a DAB converter between the first port 102 and the second port 104, and a DAB SRC between the second port 104 and the third port 106. The power transfer characteristics of each converter type, DAB and DAB SRC, provides a means to control each converter independently, thereby allowing control of different power transfer rates to each port.

[0062] As described above each port 102, 104, 106 of a three-port DC-DC converter includes a switching network 108, 110, 112 configured to convert between a DC power source Vi, V2, V3 and a corresponding AC power source VACI, V _AC2 , VAC3. Any appropriate switching network 108, 110, 112 configured to convert between a DC power source and an AC power source may be advantageously employed in the converter embodiments disclosed herein. [0063] The main purpose of switching networks 108, 110, 112 in a three-port DC-DC converter, such as the exemplary apparatus 100 described above, is to excite the impedances Z ₁ , Z ₂ , Z ₃ by providing an AC voltage V _AC1 , V _AC2 , V _AC3 to each impedance Z ₁ , Z ₂ , Z ₃ , respectively The shape of the generated AC voltage is dependent on the type of switching network. However, any appropriate AC voltage may be beneficially employed. The choice of switching network is independent of the choice of impedances and is a matter of design choice. The switching network may be selected based on the desired converter specifications. In certain embodiments it may be advantageous to use different types of switching networks for different converter ports.

[0064] Figure 7 illustrates a schematic diagram 700 of an exemplary full bridge switching network 702 coupled to a DC power source V _P . The full bridge switching network 702 is appropriate for use as any of the switching networks 108, 110, 112 described above and with reference to the disclosed embodiments. The exemplary full bridge switching network 702 includes a DC power rail 704, which includes a positive DC rail 706 and a negative DC rail 708. The DC power rail 704 is coupled in parallel with a DC power source V _P . In certain embodiments it is advantageous to include a filter capacitor Cp coupled in parallel with the DC power source V _P to remove unwanted high frequency fluctuations from the DC power source V _P .

[0065] The exemplary full bridge switching network 702 includes a first pair of switches Sa, Sc coupled in series between the positive DC power 706, and the negative DC power 708 and forming a first central node 710. A second pair of switches Sb, Sd is coupled in series between the positive DC power 706, and the negative DC power 708 and forming a second central node 712. The four switches Sa, Sc, Sb, Sd are operated to provide an AC power source VAC across the two central nodes 710, 712. [0066] In the illustrated embodiment each switch Sa, Sc, Sb, Sd is coupled in parallel with a respective free-wheeling diode D _a , D _c , D _b , D _d to protect the corresponding switch Sa, Sc, Sb, Sd from voltage stresses. The diodes D _a , D _c , D _b , D _d may be integrated with the switches themselves or when desired may be implemented using separate devices. The exemplary full bridge switching network 702 is illustrated with field effect transistors having diode protection, however those skilled in the art will readily recognize that any appropriate switching device capable of switching the desired power at the desired frequencies may be advantageously employed without straying from the spirit and scope of the present disclosure.

[0067] Figure 8 illustrates a schematic diagram 800 of an exemplary half bridge switching network 802 coupled to a DC power source V _P . The half bridge switching network 802 is appropriate for use as any of the switching networks 108, 110, 112 described above and with reference to the disclosed embodiments. The exemplary half bridge switching network 802 includes a pair of switches S _e , S _f coupled in series between a positive DC voltage 806 and a negative DC voltage 808 forming a central node 812 between the two switches S _e , S _f . A DC power source V _P is coupled across the DC power rail 804. In one embodiment, an output capacitor C _P is coupled in parallel with the DC power source V _P to provide conditioning of the DC power source V _P . Any appropriate type of switching device capable of switching the DC power source V _P to create the AC power source VAC may be used for the switches S _e , S _f .

[0068] In the example of Figure 8, each switch S _e , S _f includes an integrated protection diode D _e , D _f . In operation the switches S _e , S _f may be controlled to generate an AC voltage across the central node 812 and the negative DC voltage 808. [0069] Figure 9 illustrates an exemplary embodiment of a three-port DC-DC converter 900 implemented using full bridge switching networks 908, 910, 912 in each port 902, 904, 906. The full-bridge switching network is the most widely used network, however other types of switching networks may be advantageously employed as a matter of design choice. The full bridge switching networks 908, 910, 912 may employ any appropriate full bridge switching network such as the full bridge switching network 702 described above and with reference to Figure 7.

[0070] The exemplary converter 900 employs a high frequency impedance network 920 similar to the high frequency impedance network 220 described above and with reference to apparatus 200 where the first impedance Z ₁ is a pure inductance formed by an inductor L202, the second impedance Z ₂ is set to zero, Z ₂ =0, and the third impedance Z ₃ is a series resonant impedance formed by an inductor L204 in series with a capacitor C206. The transformer Ti, has a turns ratio Ni, and the transformer T2 has a turns ratio N2 where Ni and N2 may, when desired, be set to different values.

[0071] Figure 10 illustrates graphs 1000 showing operating waveforms for the three-port DC-DC converter 900 shown in Figure 9 and described above. The graphs 1000 depict time along the horizontal axis 1004 increasing to the right. Magnitude of voltages and currents is depicted along a vertical axis 1002 increasing upwards. The three AC voltages VACI, V _AC2 , VAC3 generated by each of the switching networks 908, 910, 912 respectively are shown. The AC currents i _p1 , i _P2 , i _P 3 associated with each AC voltage VAC1, V _AC2 , VAC3 respectively, are graphed along with the AC voltages to illustrate the voltage-current relationship of each port. Switch states, on or off, used to generate the AC voltages V _AC1 , V _AC2 , V _AC3 by the switching networks 908, 910, 912 are shown along the bottom 1006 with arrows indicating the corresponding switching time to through t ₈ . [0072] Phase relationships among the AC voltages V _AC1 , V _AC2 , V _AC3 , referred to herein as the external phase shift, control power transfer between the ports. The external phase shift between the first AC voltage VACI and the second AC voltage V _AC2 is represented as <φ ₁₂ , and the external phase shift between the second AC voltage V _AC2 and the third AC voltage VAC3 is represented as (p23. The internal phase shift of each the AC voltage is labelled as ai, α ₂ , 013 for each AC voltage VACI, V _AC2 , VAC3 respectively. The internal phase shift represents a difference between the nominal start of a period of an AC voltage and the actual turn on time of the corresponding switching network.

[0073] The control parameters available for controlling the three-port DC-DC converter are shown in the graph 1000 and include the external phase shifts φ12 , φ23 , the internal phase shifts α1, α2, α3, and the switching frequency f _s . Where the switching frequency f _s is equal to 1/(2T) and is the switching frequency of the three switching networks 108, 110, 112. The switching frequency fs used in a three-port DC-DC converter, such as the converter 100, is the same for all three switching networks 108, 110, 112. The average output power P _o1 , P _o2 , P _o3 of each port 902, 904, 906 is given by equations (1), (2), and (3), when P _o1 is sending power and P _o3 and P _o3 are the receiving power: where ai, α ₂ , 013 are the internal phase shifts within each AC voltage VACI, V _AC2 , V _AC3 ; (φ ₁₂ , (p23 are the external phase shifts between the first VACI and second V _AC2 AC voltages and the second V _AC2 and third VAC3 AC voltages respectively; Z ₁ and Z ₃ are impedances given by equations (4) and (5) with Z ₁ being a pure inductance and Z ₃ being a series resonance:

The quantity M12 is the voltage gain across the first impedance Z ₁ given by equation (6):

Mi2=(N ₁ V ₂ )/V ₁ (6) where Ni is the transformer turns ratio of transformer T ₁ . M23 is the voltage gain across the third impedance Z ₃ given by equation (7): 2 ₃ =(N ₂ V ₃ )/(N ₁ V ₂ ) (7) where N2 is the transformer turns ratio of transformer T ₂ . fs is the switching frequency of all the switching networks 908, 910, 912 of Figure 9.

[0074] The term “internal phase shift α ₁ , α ₂ , α ₃ ” as used herein is a delay between a zero crossing 1008, 1010, 1012 of a fundamental of an AC voltage VACI, VAC ₂ , V _AC3 and the turn on time t ₂ , t4, t6 of the associated switching network 908, 910, 912 respectively. For example, it can be seen from the graph 1000 that the fundamental of the third AC voltage V _AC3 begins at time 1012 and the third switching network 912 is turned on at a later time te. The difference between the start of the period beginning at time 1012 and the turn on time t ₆ is referred to as the internal delay of the third AC voltage V _AC3 .

[0075] As discussed above, the use of dual transformers with asymmetric impedances having a combination of pure inductive and series resonant impedances was shown to decouple the ports of a three-port DC-DC converter at the circuit architecture level. Now, analysing the power equations (1) through (7), shows that the control parameter α ₂ , which corresponds to the internal phase shift of the second port 904, and the switching frequency f _s of the switching networks 908, 910, 912 are common to all power equations (1), (2), and (3), and are still coupling all three ports 902, 904, 908 at the circuit control level. Setting the internal phase shift of the second port to zero α ₂ =0 removes a significant portion of this coupling. Regarding the switching frequency f _s , it can be observed from equations (4) and (5) above that the first impedance Z ₁ , which is a pure inductance, and the third impedance Z ₃ , which is a series resonant impedance, vary at a different rates with respect to the switching frequency f _s .

[0076] Figure 11 illustrates graphs 1100 showing operating waveforms for the three-port DC-DC converter topology 900 shown in Figure 9 and described above. The graphs 1100 illustrate a case where the second switching network 910 is operated with the second internal phase shift α ₂ set to zero. The graphs 1100 depict time along the horizontal axis 1104 increasing to the right. Magnitude of voltages and currents is depicted along a vertical axis 1102 increasing upwards. The three AC voltages V _AC1 , V _AC2 , V _AC3 generated by each of the switching networks 908, 910, 912 respectively are shown with corresponding labels V _AC1 , V _AC2 , V _AC3 . The AC currents i _p1 , i _p2 , i _p3 associated with each AC voltage V _AC1 , V _AC2 , V _AC3 respectively, are graphed along with the AC voltages to illustrate the voltage-current relationship of each port 902, 904, 906. Switch states, on or off, used to generate the AC voltages V _AC1 , V _AC2 , V _AC3 by the switching networks 908, 910, 912 is shown along the bottom 1106 with arrows indicating the corresponding switching time to through t ₈ .

[0077] Setting the internal phase shift α ₂ of the second AC voltage V _AC2 to zero reduces the total control parameters to five. These control parameters including the internal phase shift of the first AC voltage ai, the internal phase shift of the third AC voltage 013, the external phase shift between the first and second AC voltage φ ₁₂ , the external phase shift between the second and third AC voltage φ ₂₃ , and the switching frequency of the ports f _s . These parameters can be used to form an optimal and efficient control modulation strategy for the power transfer between the ports 902, 904, 906 of the converter 900.

[0078] The average power of the three ports 902, 904, 906 with the internal phase shift of the second port α ₂ set to zero is shown by equations (8), (9), and (10), when P _o1 is sending power and P _o2 and P _o3 are the receiving powers: where the parameters are as described above.

[0079] These average power equations (8), (9), and (10) along with the associated waveforms illustrated in the graph 1100 show how asymmetric impedances can be used to form a quasi-decoupled three-port DC-DC converter providing significantly improved controllability.

Equations (8), (9), and (10) represent an apparatus, such as the apparatus 100 when full bridge switching networks are used, where the first impedance Z ₁ is a pure inductance, the second impedance Z ₂ is set to zero, and the third impedance Z ₃ is a series resonant impedance formed by an inductor in series with a capacitor. Shifting these impedance values among the three impedances Z ₁ , Z ₂ , Z ₃ and setting the internal phase shift associated with the zero impedance to zero leads to a family of three converter topologies.

[0080] The power equations for a three-port DC-DC converter, such as the apparatus 100 described above when full bridge switching networks are used, where the first impedance Z ₁ is set to zero, the first internal phase shift α ₁ is set to zero, the second impedance Z ₂ is a pure inductance, and the third impedance Z ₃ is a series resonant impedance formed by an inductor in series with a capacitor, are shown in equations (11), (12), and (13), when P _o1 is sending power and P _o2 and P _o3 are the receiving powers: where the second impedance Z ₂ is given by equation (14)

Z ₂ = 2 f _s L ₂ (14) the third impedance Z ₃ is given by equation (15) the voltage gain M12 across the second impedance Z ₂ is given by equation (16) 1 ₂ =(N ₁ V ₂ )/V ₁ (16) where Ni is the turns ratio of the first transformer Ti. The voltage gain across the third impedance M ₁₃ is given by equation (17) 1 ₃ =(N ₂ V ₃ )/V ₁ (17) where N2 is the turns ratio of the second transformer T ₂ . ( φ ₁₂ and φ ₂₃ are the external phase shifts between the first VACI and second V _AC2 AC voltages and the second V _AC2 and third V _AC3 AC voltages respectively.

[0081] Referring now to Figure 12 there can be seen an exemplary embodiment of a three- port DC-DC converter apparatus 1200 employing an alternate high frequency impedance network 1220. The exemplary apparatus 1200 is similar to the exemplary apparatus 100 described above and with reference to Figure 1 where like references indicate like elements.

[0082] In the illustrated embodiment the high frequency impedance network 1220 provides design and manufacturing benefits through the use of a dual transformer T ₁₂ , T ₂₂ configuration as described above where the first winding 1222 of the first transformer T ₁₂ is coupled in parallel with the first winding 1226 of the second transformer T ₂₂ . In the high frequency impedance network 1220 a first port 102 is coupled to a first winding 1222 of the first transformer T12 through a first impedance Z ₁ , a second port 104 is coupled to a second winding 1224 of the first transformer T12 through a second impedance Z ₂ , and a third port 106 is coupled to a second winding 1228 of the second transformer T ₂₂ through a third impedance Z ₃ .

[0083] Circuit level decoupling is achieved in the high frequency impedance network 1220 by setting the first impedance Z ₁ to a series resonant impedance formed by an inductor L1204 coupled in series with a capacitor C1206, setting the second impedance Z ₂ to a zero impedance, and setting the third impedance Z ₃ to a purely inductive impedance which is illustrated as an inductor L1202. Control level decoupling is achieved in the apparatus 1200 by controlling operation of the second port 104, which in the exemplary embodiment 1200 corresponds to the zero impedance Z ₂ , such that the second internal phase shift α ₂ of the second port 104 is maintained at zero. The second internal phase shift α ₂ of the second port 104 is the same phase shift α ₂ described above and with reference to Figure 10.

[0084] As an aid to understanding a simple series resonant impedance formed by an inductor in series with a capacitor was presented in the above discussion, however those skilled in the art will readily recognize that any resonant impedance network exhibiting different power transfer characteristics than a pure inductance may be advantageously employed in place of the series resonant impedance without straying from the spirit and scope of the present disclosure.

[0085] The family of quasi-decoupled three-port DC-DC converter topologies described above enable a low-cost solution for integration and power sharing among three variable DC voltage sources. The two-level decoupling approach disclosed herein includes both circuit architecture level and circuit control level decoupling to achieve quasi-decoupling among all three ports. The two-level decoupling approach eases control of the resulting three-port DC-DC converter, such as the apparatus 100 described above and with reference to Figure 1, simplifies control of the apparatus 100 by giving more freedom to manipulate control parameters to achieve desirable and efficient operation, such as by enabling soft-switching and minimizing circulating current. The use of two transformers eases manufacturing and production processes resulting in lower costs. Use of a series resonant impedance network can reduce the size of a required electromagnetic-interference (EMI) filter resulting in higher power density and lower cost. [0086] Thus, while there have been shown, described and pointed out, fundamental novel features of the invention as applied to the exemplary embodiments thereof, it will be understood that various omissions, substitutions and changes in the form and details of devices and methods illustrated, and in their operation, may be made by those skilled in the art without departing from the spirit and scope of the presently disclosed invention. Further, it is expressly intended that all combinations of those elements, which perform substantially the same function in substantially the same way to achieve the same results, are within the scope of the invention. Moreover, it should be recognized that structures and/or elements shown and/or described in connection with any disclosed form or embodiment of the invention may be incorporated in any other disclosed or described or suggested form or embodiment as a general matter of design choice. It is the intention, therefore, to be limited only as indicated by the scope of the claims appended hereto.