CONTROL CIRCUIT AND METHOD

This invention relates to a method of controlling an element in a gas sensor and to a pulse modulation control circuit for implementing the method. Specifically, the disclosed technique reduces the amount of power consumed by the circuit.

As will be described below, the control circuit is primarily envisaged for use with a gas sensitive element. However, it will be appreciated that the control circuit could be used in other applications, such as the control of heating elements or anemometers. Similarly, the method and circuit are particularly well adapted for use with pellistor type gas sensors and, as such, the description below will focus on such applications. However, it will be realised that the techniques could be used with other types of gas sensor including semistors, semiconductor sensors or electrochemical sensors. For example, the techniques could be used to maintain a high temperature solid oxide electrolyte sensor, such as a zirconium oxide sensor (commonly used in exhaust pipes), at its optimum operating temperature.

Calorimetric sensors, such as pellistors, have long been used for the measurement of combustible gases. The principles of operation are described, for example, in WO04/048955 and in "Calorimetric Chemical Sensors", PT. Walsh and T.A. Jones, Chapter 11 in "Sensors - A Comprehensive Survey" volume 2, editors W. Gopel, J. Hesse and J.N. Zemel.

Typically, a pellistor includes a gas sensitive element, which is provided with a catalytic coating on which combustible gases react, and (optionally) a compensator element which is catalyst- free. The gas reaction on the gas sensitive element evolves heat which changes the resistance of the element, and this is monitored in a detection circuit. Gas does not react on the compensator element, but its resistance changes in response to changes in the ambient conditions, such as temperature and/or humidity, thereby providing a reference against which the output from the gas sensitive element can be compared.

The elements can be controlled under constant current or constant voltage regimes, in which case the varying resistance of the elements is used to evaluate the concentration of combustible gas in the atmosphere. In conventional sensor devices, the two pellistor elements form part of a Wheatstone bridge circuit which generates a signal related to the difference between the resistance of the gas sensitive element and that of the compensator element.

Alternatively, the elements can be run in constant temperature mode, in which case the power supplied to each element is controlled such that the temperature (and therefore the resistance) of each remains constant. The variation in the power supplied required to maintain the element at constant temperature is used to evaluate the concentration of gas. This mode has the advantage that the pellistor elements remain at their optimum operating temperature throughout use.

As described in our co-pending International Application No. PCT/GB2006/001401, a primary concern for users of pellistor sensors is the power consumption of the device. Particularly in portable devices, it is desirable to reduce the power consumed by the pellistor element in order that smaller batteries are required and hence smaller, lighter and cheaper sensing devices may be designed. The primary aim of the approach described in the above International Application is to allow the two pellistor elements to be switched on and off individually, thereby offering the possibility of power saving. This is achieved by providing each element with its own individual pulse modulation controlled feedback circuit, which may be analogue or digital in nature and may be used to realise constant temperature, constant current or constant voltage control. In one proposed mode of operation, the gas sensitive element is continuously powered whilst the compensator element is operated for only a fraction of the time, such as ten seconds in every two minutes. This offers a power saving related to the beads themselves of approximately 40%.

In practical applications within a gas sensing instrument, the full power saving offered by such modes of operation may be compromised if the voltage rail used to supply the pulse modulation circuit has to be regulated, since such a step inevitably involves some power dissipation. Such wastage may be exacerbated by the use of individual circuits for the two elements as opposed to the situation which exists in, for example, a conventional Wheatstone bridge where the two elements are in series (but where they cannot be individually controlled in the required fashion).

In most systems, one of the main advantages of using a pulse modulation approach would normally be to overcome the need for such power regulation. However, in a gas detection instrument, special circumstances may apply which can result in erroneous signals if the power supply is not regulated:

• The pellistor elements typically have relatively slow thermal response times, of the order of seconds or hundreds of milliseconds, depending on the bead size. In effect, this defines the period over which a reliable measurement of the

resistance can be obtained and hence the necessary error signal for the feedback loop. However, if the power supply to the pulse modulator changes significantly over a comparable or shorter time scale, the pellistor element may be receiving an erroneous drive current. This would go undetected until the pellistor element "catches up" with the change in supplied power, during which time the resistance measurement (and so the measured gas concentration) would be incorrect.

• In a battery powered instrument, it is natural for the supply voltage to change as the level of charge in the battery alters. This normally occurs over a relatively long period of the order of hours, and so is well compensated by a standard pulse modulation method. However, if the instrument goes into an alarm mode, then the power drain may increase very suddenly due to the use of visual and audible indicators. Very often, these are turned on and off at a frequency of a few hertz and therefore result in battery output voltage fluctuations on a timescale which is comparable with or shorter than the thermal response time of the pellistor elements, and can therefore produce the unwanted effects described above.

As a result, additional power regulation is presently required to cope with such circumstances, and this results in a much reduced power saving compared with that which is theoretically available.

In accordance with the present invention, a pulse modulation control circuit for connection to an element in a gas sensor, comprises: a pulse modulator for generating a pulsed signal from a voltage supply, the pulsed signal supplying power to the element; and a controller adapted to: monitor a parameter which varies with the temperature of the element, and compare the monitored parameter to a first set-point (SP ₁ ) value to generate a first error value (E ₁ ); monitor the pulsed signal generated by the pulse modulator, calculate a value

(V _CALC ) related to the root mean square (RMS) voltage (V _RMS ) of the pulsed signal and compare the calculated RMS-related value (V _CALC ) to a second set-point value (SP ₂ ) to generate a second error value (E ₂ ), the second set-point value (SP ₂ ) being derived from the first error value (E ₁ ); and

control the pulse modulator in accordance with the second error value (E ₂ ), thereby controlling the power consumed from the voltage supply by the element.

The invention further provides a method of controlling an element in a gas sensor, comprising: generating a pulsed signal from a voltage supply, the pulsed signal supplying power to the element; monitoring a parameter which varies with the temperature of the element; comparing the monitored parameter to a first set-point value (SP ₁ ) to generate a first error value (E ₁ ); monitoring the pulsed signal generated by the pulse modulator to calculate a value (V _CALC ) related to the root mean square (RMS) voltage (V _RMS ) of the pulsed signal; comparing the calculated RMS-related value (V _CALC ) to a second set-point value

(SP ₂ ) to generate a second error value (E ₂ ), the second set-point value (SP ₂ ) being derived from the first error value (E ₁ ); and controlling the pulsed signal in accordance with the second error value (E ₂ ) thereby controlling the power consumed from the voltage supply by the element.

By monitoring the pulse signal and processing it in this way, the pulse modulated signal can be adjusted to take account of changes to the voltage supply. This does away with the need for regulating the voltage rail, and thereby enables the sensor to achieve its full power saving potential. Advantageously, the additional processing can be carried out using software without adding to the complexity of the pellistor pulse modulation drive circuit. Additionally, the invention potentially reduces costs in other parts of the instrument circuit by reducing the regulation demands on them. Effectively, the invention performs the functions of two feedback loops in order to accurately control power to the element. The first monitors a parameter such as the resistance of the element and compares this with a set point to generate an error value representative of the state of the pellistor element (compared to the ideal operational state of the element). The second feedback loop monitors the pulsed signal generated by the pulse modulator, which is supplied to the pellistor element. Importantly, the monitored signal is independent of the thermal response time of the pellistor element. As such, the second feedback loop operates at a rate much faster than the resistance controlling loop. This makes it possible to compensate for changes in the voltage rail occurring at a frequency greater than the rate at which the bead can respond to thermal changes, such as may occur in an alarm scenario as described above. This

second feedback loop compares a value related to the root mean square voltage of the pulsed signal (which is directly related to its power) to a second set point value, which is derived from the output of the first feedback loop. The eventual output from the second feedback loop therefore includes a correction for both the resistance variation and the voltage supply variation, and the pulse modulator can then adjust its output accordingly to maintain the resistance (or other monitored parameter) of the element substantially constant even when the voltage supply varies.

Preferably, the pulse modulator comprises a pulse modulation driver and a switching device, the switching device being connected in series between the voltage supply and the element, and the pulse modulation driver being arranged to operate the switching device to generate the pulsed signal. With this arrangement, in its "off state, the switch entirely isolates the element from the power supply, which provides an optimal power saving when the element is operated via a pulsed signal, and additionally the same switch can be used to turn off an element entirely if it is being used in an intermittent mode.

Advantageously, the second set-point value (SP ₂ ) is derived from the first error value (E ₁ ) according to the relationship SP ₂ = C ± E ₁ , where C is a predetermined constant. The predetermined constant C is typically determined based on the optimum operating condition of the pellistor element. The present inventor has found that a value of C = 1.75 ² is appropriate for typical pellistors.

Preferably, the parameter which varies with the temperature of the element is the resistance of the element. As discussed above, the resistance of a gas sensitive element is directly related to the concentration of combustible gas in the atmosphere under test. Conveniently, the controller calculates the resistance of the element from measurements of the voltage across and current through the element.

Preferably, the RMS-related value (V _CALC ) is the square of the RMS voltage (V _RMS ) °f the pulse signal. Alternatively, any other function of the root mean square voltage of the pulse signal could be used instead. However, it is advantageous to use the square of the RMS value (or a function of the square) so that the need to calculate a square root is avoided. Calculating a square root takes significant time and processing capacity and may slow the processing down to an unacceptable level since a fast response is required should the supply voltage change.

Conveniently, the controller calculates the RMS-related value (V _CALC ) from measurements of the peak voltage and duty cycle of the pulsed signal. Preferably, the controller calculates the RMS-related value (V _CALC ) according to the relationship v2

VCALC - {VR _M S) ² = DC.(V _≠ )

where V _RMS = root mean square voltage, OC = duty cycle and V _p11 = peak voltage of the pulsed signal.

In most cases, it is preferable that the controller is adapted to control the pulse modulator such that the second error value (E ₂ ) tends to zero. However, in alternative embodiments, it may be advantageous to have the second error value (E ₂ ) tend to some other limit.

Preferably the controller is adapted to control the pulse modulator by adjusting the duty cycle of the pulsed signal. Advantageously, this is achieved by adjusting the width of the pulses (i.e. pulse width modulation). The controller could additionally or alternatively be adapted to control the pulse modulator by adjusting the frequency of the pulse signal.

Advantageously, the controller is adapted to control the pulse modulator in accordance with the second error value so as to maintain the monitored parameter of the element substantially constant.

Preferably, the controller further provides the pulse modulation driver. To obtain a full power saving, it is preferable that the power supply is unregulated, since the invention does away with the need for any regulation.

In certain preferred embodiments of the invention, the element is a gas sensitive element. In other preferred embodiments of the invention, the element is a compensator element. Preferably, the controller comprises a first feedback loop for monitoring the parameter which varies with the temperature of the element, and comparing the monitored parameter to the first set point value (SP ₁ ) to generate the first error value (E ₁ ). Further preferably, the controller comprises a second feedback loop for monitoring the pulsed signal generated by the pulse modulator, calculating a value (V _CALC ) related to the root mean square (RMS) voltage of the pulsed signal and comparing the calculated RMS related value (V _CALC ) to a second set-point value (SP ₂ ) to generate the second error value (E ₂ ), the second set-point value (SP ₂ ) being derived from the first error value (E ₁ ).

In accordance with another aspect of the invention, a gas sensor assembly is provided comprising a gas sensitive element arranged in series with the voltage supply, and a first pulse modulation control circuit as described above, connected to the gas sensitive element. Preferably, the gas sensor assembly further comprises a compensator element arranged in series with the voltage supply independently of the gas sensitive element, and a second pulse modulation control circuit as described above connected to the compensator element. As such, both of the elements can be independently controlled in pulse modulation circuits without need for voltage supply regulation and a significant power saving is achievable. Examples of circuits and methods in accordance with the present invention will now be described with reference to the accompanying drawings, in which:

Figure 1 is a block diagram which schematically illustrates the functions of a conventional control circuit;

Figure 2 is a graph showing the effect of a power supply disturbance on the power supplied to the sensor element in the arrangement shown in Figure 1 ;

Figure 3 is a block diagram which schematically illustrates the functions of a control circuit according to an embodiment of the invention;

Figure 4 is a graph showing the effect of a power supply disturbance on the power supplied to the sensor elements in the arrangement shown in Figure 3; Figure 5 is a block diagram schematically illustrating in more detail the control circuit of Figure 3; and

Figure 6 is a schematic circuit diagram illustrating an exemplary sensor device implementing the control circuit of the embodiment.

The invention will now be described with reference to its application in pellistor type sensors. However, as mentioned above, the control circuit and method described herein could readily be applied to any other type of sensor such as semistors or semiconductor sensors. Similarly, the technique could readily be extended to applications such as heater element control.

As already described, the major advantage of using a pulse modulation drive circuit for the control of pellistors is the power saving obtained compared with a normal

Wheatstone bridge circuit. This advantage may be diminished or lost if a linear voltage regulator has to be used to control the power supply to the pulse modulation circuit in order to maintain accuracy. A conventional control circuit which may suffer from such problems, is shown in terms of its functional steps in Figure 1. The sensor element 1 , which may be a gas sensitive (detector) element or a compensator element, is

powered by a pulsed drive signal from pulse modulator 2. In practice, the pulse modulator 2 typically comprises a pulse modulation driver and a switching device, wherein the switching device is disposed in series with the sensor element 1 and a power supply (not shown in Figure 1). Further details of such an arrangement will be described below in relation to Figure 6.

The pulse modulator generates the pulsed signal by modulating power from a power supply (not shown), which may be subject to variations. Such fluctuations are represented in Figure 1 by voltage disturbance V _DD , shown as an error input. In other words, if there is no fluctuation, V ₀₀ = 0, and when the power supply deviates from its intended value, V ₀₀ has a positive or negative non-zero value.

As described above, each pellistor element is typically operated in one of constant temperature, constant current or constant voltage mode. In each case, a parameter of the sensor element 1 is monitored and the pulsed signal adjusted (usually in terms of its duty cycle) to maintain the monitored parameter at a constant value. To this end, a parameter controller module 3 is provided which receives input from feedback line 8 and compares the monitored parameter against a first set point value to generate an error value. In the example shown, the pellistor is operated in constant temperature mode and, as such, the monitored parameter is the resistance of the element, R. The resistance of the element 1 is monitored by taking measurements of the voltage across and the current through the element 1 and applying Ohm's Law. The first set point value, SP ₁ , is a predetermined constant calculated during calibration (i.e. in zero combustible gas), and corresponds to the resistance of the element 1 at its optimum operating temperature. An appropriate value for typical sensor elements has been found to be approximately 25 ohms. The parameter controller module 3 outputs a first error value E ₁ , which represents the difference between the monitored parameter (here the resistance) and the set point value, SP ₁ . This error signal E ₁ is used to adjust the output of the pulse modulator 2 to modify the power supplied to the sensor element 1 in order to keep the resistance at a constant level. Typically this is achieved by modifying the duty cycle of the pulsed signal, although its frequency could additionally or alternatively be varied. In most cases, adjustment of the duty cycle will be effected by modifying the pulse width; i.e. increasing or decreasing the proportion of each cycle for which the signal is "high". However, it is possible to achieve control by altering things other than the pulse width or frequency. For example, the amplitude of the pulsed signal could, in principle, be

adjusted to control the power supplied to the element. It will be appreciated that this is not a preferred technique in the present case, since it is desired that the power supply be unregulated and as such controlling the amplitude may be difficult and inaccurate. However, this has been found to be effective where the power supply fluctuations are restricted to a narrow band [please confirm this paragraph is correct].

However, where there is a non-zero voltage supply disturbance V _DD , unknown to the pulse modulator 2, the pulsed signal it generates has either a higher or a lower amplitude than expected, and as a result the power supply to the sensor element 1 is incorrect. For example, if the power supply voltage suddenly fell by 300 mV (V _DD = -300 mV), the pulsed signal would have an unexpectedly low amplitude and the resulting power (which is related to the amplitude and duty cycle of the pulsed signal) would also be reduced. This scenario is shown in Figure 2, where the top trace (P**) shows the power supplied to the element 1 changing by about 20 mW when the power supply (Vp ₅ ) changes by a step of 300 mV. As can be seen, after some time, the system corrects itself. However, the power to the detector, and therefore the calculated gas concentration, is incorrect for a significant duration. This is caused by the long thermal time constant of the sensor element 1.

By introducing a second feedback control loop, the effect of power supply disturbances can be minimised. Figure 3 shows a functional block diagram of an embodiment of the invention which implements this concept. Where appropriate, the same reference numerals as used in Figure 1 are used to indicate the same components. Thus, the sensor element 1 , pulse modulator 2 and parameter controller module 3 all correspond to the components shown in Figure 1.

A voltage controller module 4 is provided which takes an input from feedback line 9 which monitors the pulsed signal generated by the pulse modulator 2. As will be described in more detail below, certain features of the pulsed signal are used to calculate a value V _CALC which is compared against a second set point value, SP ₂ , to output a second error value E ₂ . The second set point value SP ₂ is derived from the first error value E ₁ output by the parameter controller 3. Thus the second error value E ₂ includes a correction relating to the resistance (or other parameter) of the sensor element 1 , as well as a correction for any voltage disturbance detected in the pulsed signal. The second error value E ₂ is used to control the pulse modulator 2 such that changes in the element resistance are compensated for, and variations in the voltage supply are additionally compensated for.

Figure 4 shows the effect of the 300 mV power supply disturbance on the sensor element power in the arrangement of Figure 3, and it will be seen that there is a much reduced deviation experienced, which is quickly corrected for. The technique is effective since the second feedback loop (consisting of voltage controller module 4 and feedback line 9) is independent of the long thermal response time of the sensor element 1. Thus the voltage correction is updated at the same frequency as the pulsed signal generated by the pulse modulator 2. As such, any variations in the power supply are very quickly corrected for.

It should be noted that, whilst this technique can deal with power supply fluctuations, there are still some over-riding constraints in the power supply. In particular, the power supply must be capable of providing a minimum level of power such that the pellistor element can run in its intended mode when the pulsed signal has a reasonable duty cycle. That is, the necessary power should be supplied to the element without having to select a very high (e.g. >80%) or very low (e.g. <20%) duty cycle. This is because the duty cycle has to be monitored by the controller and this is difficult at such extremes since either the pulses or the troughs become too short to measure accurately. Thus, it is preferred that the duty cycle of the pulsed signal be between 25% and 75% and the power supply should be able to supply sufficient power to allow for this. Figure 5 shows in more detail the relationship between the parameter controller module 3 and the voltage controller module 4. In practice, these functions will be carried out by a controller (typically a microcontroller, but this could comprise discrete components), however they are shown as separate functional components here for clarity. Preferably, the functions are implemented in software such that no changes to the hardware are required.

As already described, the parameter of the sensor 1 (usually the resistance, R) is evaluated by monitoring the current through and/or the voltage across the sensor element 1. The parameter controller module 3 compares this value input from feedback line 8 with a first set point value SP ₁ , typically abut 25 Ohms where the monitored parameter is resistance. A first error value E ₁ is generated and, if necessary, this is converted into a voltage using Ohm's law. This error signal is provided to the voltage controller 4 and used to derive a second set point value SP ₂ according to the relationship SP ₂ = C ± E ₁ , although other functions of E ₁ could be used. Hence when there is no deviation from the set point resistance, the first error value E ₁ will be zero and the second set point value SP ₂ equals the calibration constant

C. C is a predetermined constant related to the calibration voltage which is applied in an atmosphere of zero combustible gas to produce the desired resistance in the sensor element (i.e. when R = SP ₁ ). The calibration voltage is therefore determined by the first set point, SP ₁ , in that a voltage is supplied which maintains the element at the same temperature (and resistance) at which it would sit if it were powered by a DC signal rather than a pulsed signal.

The voltage controller module 4 compares the second set point value SP ₂ with a value V _CALC which is derived from monitoring the pulsed signal generated by the pulse modulator 2. In order to obtain a value which is directly related to the power being supplied to the sensor element 1, it is necessary to evaluate the root mean square (RMS) voltage V _RMS of the pulse signal, or a function of the RMS voltage. The RMS voltage (V _RMS ) can be derived from measurements of the pulse signal according to the relationship

where DC is the duty cycle of the pulse signal (i.e. the proportion of each cycle wherein the pulse is "high") and V _PK is the peak voltage of the pulsed signal (i.e. the amplitude). However, in practice it is preferable to avoid the calculation of a square root since this can significantly slow the processing of the signal and lead to an unacceptably long delay before any voltage variation is corrected for. As such, the value input to the voltage controller module 4, V _CALC , is preferably a function of the square of the RMS voltage (V _RMS ). In this embodiment,

Hence, so that a comparison can be carried out, the second set point value SP ₂ must also be in units of (volts) ² . The output from parameter controller 3 (first error signal E ₁ ,, is therefore squared and likewise the value C equals the square of the calibration voltage.

The calibration voltage is determined based on the optimum operating condition of the sensor element 1. Typical values of the calibration constant are C = (1.75 volts) ² for a gas sensitive element and C = (1.30 volts) ² for a compensator element. The result of the comparison between V _CALC and SP ₂ is a second error value E ₂ which is used to control the pulse modulator so as to adjust the pulsed signal to compensate for changes in the monitored parameter of the sensor element 1 and in the power supply.

To calibrate the system, a calibration terminal 7 is provided, which shorts the parameter controller 3 out of the circuit. The calibration voltage is thus applied in the form of a pulsed signal across the sensor 1 with any variations being corrected for by the voltage controller module 4. The resulting resistance (or other parameter) of the sensor element 1 is measured and this value is used as the first set point value SP ₁ for future operation of the device. As described above, this value is typically about 25 Ohms for a gas sensitive element operating in constant temperature mode.

Figure 6 shows a schematic circuit diagram of a sensor device having a detector (gas sensitive) element 11 and a compensator element 12, each of which is disposed in an individual circuit between a power supply 17 and ground G. Each circuit is provided with a respective switch, 13 and 14 in series with the element and the power supply such that, when the switch is in its off state, the respective element is entirely isolated from the power supply. As described in our co-pending International Application No. PCT/GB2006/001401 , in this arrangement, the detector and compensator elements can be operated individually and power can be saved by the use of certain intermittent operating regimes. In this example, each parallel circuit further comprises a load resistor 15,16 which together with the respective element act as a voltage divider such that parameters can be measured across the load resistance instead of across the element. However, there are other well known methods of measuring current which do not require load resistors and these components are therefore optional.

A processor 18 takes inputs 20,21 from each circuit and outputs pulsed signals 19 to the two switching devices 13 and 14. Control of each switch results in a pulsed signal through each of the parallel circuits and so provides power to the detector and compensator elements 11,12. Thus the pulse modulator of Figures 3 and 5 is in practice implemented by processor 18 in conjunction with the respective switch 13,14.

The resulting pulsed signal is monitored on lines 20 and 21 from which current measurements can also be taken. The peak voltage (V _PK ) across each pellistor element is measured by the processor via an ADC. Thus, the processor 18 can calculate V _CALC and the resistance R and perform the functions as set out in Figures 3 and 5 above. The resulting gas sensing device makes it possible to realise the full energy saving potential of the individual pulse modulation circuits.