BACKGROUND OF THE INVENTION The present invention relates to ferroresonant transformers, and deals more particularly with closed-loop ferroresonant transformers having controlled output voltage harmonics.

(1) Field of the Invention Ferroresonant transformers are often used for voltage regulation. The voltage regulation is accomplished by precisely controlling the magnetic saturation of the transformer core. The control of magnetic saturation is typically accomplished by employing a control nductor winding in addition to the input/output windings. The control winding carries a direct current in order to generate a desired amount of magnetic flux which is added to the magnetic flux generated by the input winding to form a resultant magnetic flux. The control winding can thus "fine tune" the amount of the resultant magnetic flux, and consequently adjust the amount of input current necessary to magnetically saturate the core. Ensuring operation of the transformer while in magnetic saturation ensures that the output voltage is regulated.

(2) Description of Prior Art Ferroresonant transformers have been developed to electronically simulate core saturation. In other words, voltage regulation is achieved without the need for actually magnetically saturating the core.

For example, U.S. Pat. No. 3,573,606, which is hereby incorporated by reference, shows a nonsaturating switching type ferroresonant regulator.

The function of the saturating core is supplied by an inductance and a switch.

A common drawback with prior ferroresonant transformers is that the output voltage of such transformers typically exhibit high total harmonic distortion (THD) sine waves. Expensive harmonic traps normally must be employed to bring the THD down to an acceptable level.

It is therefore an object of the present invention to provide a closed-loop ferroresonant transformer which adjusts the various electromagnetic parameters in order to control the output voltage waveform from a low THD sinewave to a high rise-time quasi square wave which improves performance and does not require the use of expensive filter chokes.

SUMMARY OF INVENTION The present invention resides in a ferroresonant transformer comprising a three-legged magnetic core. The core includes a center leg, and first and second flanking legs. Each of the center and flanking legs have respective first and second longitudinal ends. The first flanking leg is positioned at an opposite side of the center leg relative to the second flanking leg. A first end-connecting portion magnetically couples the first ends of the center and flanking legs, and a second end-connecting portion magnetically couples the second ends of the center and flanking legs. The center leg defines a substantially non-magnetic space, such as an air gap, along a path extending from the first end-connecting portion to the second end-connecting portion via the center leg. An input coil is wound around the center leg, and an output coil is wound around and longitudinally spaced on the center leg relative to the input coil A control coil is also wound around and longitudinally spaced on the center leg relative to the input and output coils for adjusting the flux density through the magnetic core.

The present invention also resides in a three-legged magnetic core. The core includes a center leg, and first and second flanking legs.

Each of the center and flanking legs have respective first and second longitudinal ends. The first flanking leg is positioned at an opposite side of the center leg relative to the second flanking leg. A first end-connecting portion magnetically couples the first ends of the center and flanking legs, and a second end-connecting portion magnetically couples the second ends of the center and flanking legs. The center leg defines a substantially non-magnetic space, such as an air gap, along a path extending from the first end-connecting portion to the second end-connecting portion via the center leg.

One advantage of the present invention is that the total harmonic distortion of the output signal of the ferroresonant transformer can be easily controlled without sacrificing performance or using expensive filter chokes.

Other objects and advantages of the present invention will become apparent in view of the following detailed description and accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS A more complete understanding of the invention and many of the attendant advantages thereto will be readily appreciated as the same becomes better understood by reference to the following detailed description when considered in conjunction with the accompanying drawings wherein: FIG. 1 schematically illustrates a prior art ferroresonant transformer.

FIG. 2 is a schematic circuit associated with the transformer of FIG. 1.

FIG. 3 is a graph illustrating waveforms of signals generated by the transformer of FIG. 1.

FIG. 4 is a graph further illustrating waveforms of signals generated by the transformer of FIG. 1.

FIG. 5 is a graph illustrating waveforms of signals having reduced THD generated by the transformer of FIG. 6.

FIG. 6 schematically illustrates another prior art ferroresonant transformer.

FIG. 7 is a graph illustrating waveforms of signals associated with a fast rise time application of ferroresonant transformers.

FIG. 8 schematically illustrates a first embodiment of a ferroresonant transformer in accordance with the present invention.

FIG. 9 is a schematic circuit associated with the transformer of FIG. 8.

FIG. 10 is a graph illustrating various features of the transformer of FIG. 11.

FIG. 11 schematically illustrates a second embodiment of a ferroresonant transformer in accordance with the present invention.

FIG. 12 is a graph illustrating various features of the ferroresonant transformer of FIG. 13.

FIG. 13 schematically illustrates a third embodiment of a ferroresonant transformer in accordance with the present invention.

BACKGROUND DISCUSSION OF FERRORESONANT TECHNOLOGY FIGS. 1-7 refer to examples of prior ferroresonant transformers in order to better understand the improvements and distinguishing features of the present invention set forth in FIGS. 8-13.

One prior art ferroresonant transformer, as shown in FIG. 1, is generally referenced by the number 10. The transformer 10 typically comprises a magnetic core having three legs: a center leg 12, and first and second flanking legs 14 and 16, respectively. The legs 12, 14 and 16 have respective first longitudinal ends that are magnetically coupled to one another via a first end-connecting portion 18 which, as shown in FIG. 1, is formed integrally with the legs. As can be seen in FIG. 1, the legs and first end-connecting portion cooperate to form an E shaped member. A separately formed second end-connecting portion 20, taking the form of an I shaped member, abuts against respective second longitudinal ends 22 of the legs 12, 14 and 16. The E shaped member and I shaped member therefore cooperate to form the magnetic core of the transformer 10. The transformer 10 further includes a first or series inductance shunt 24 and a second or control inductance shunt 26 each extending outwardly from the center leg 12. The transformer also includes an input coil 28, an output coil 30 and a control inductance coil 32 that are typically employed with ferroresonant transformers. As can be seen in FIG. 1, each of the coils extends outwardly from and circumaxially about the center leg 12.

FIG. 2 schematically illustrates an equivalent electrical circuit 33 of the transformer 10. The equivalent circuit 33 includes an AC input voltage vin at 34, an inductance 36 having reactance Xs representing the leakage inductance of the series inductance shunt 24 of FIG. 1, a resistance R at 38 representing the equivalent resistance of all the windings in FIG. 1, an inductance 40 having reactance XL representing the control inductance of the control inductance shunt 26, an inductance 42 having reactance XM representing the magnetizing inductance, a control circuit switch S at 44 that is opened and closed by a control circuit 46. The magnetizing inductance XM is usually ignored when the control circuit switch S is opened because XM>>XL. An output capacitor at 48 provides the output voltage vO across its terminals.

The output voltage vO of a controlled ferroresonant transformer, such as the transformer described in FIGS. 1-4, is the voltage ve across the output capacitor 48 (shown in FIG. 2), and is defined by the following equations set forth below. All lower case symbols are instantaneous values (i.e., v(t), i(t), etc.), and all upper case symbols are root-mean-square (RMS) values (i.e., V, I, etc.).

(1) (2) (3), when S is open (4), when S is closed (5) where: Vo or vo(t) is the output voltage, Vc or vc(t) is the voltage across the output capacitor, Ic or ic(t) is the current charging/discharging the output capacitor as defined by equation (1), IL or iL(t) is the control inductor current, C is the capacitance of the output capacitor, Iind is the current of the control inductor coil and switch, XM is the magnetizing reactance, L is the control inductance, to is the time the switch S turns on, t1 is the time the switch S turns off, t is any point in time between to and t1, the on/off time being determined by the control circuit 46. FIGS. 3 and 4 are graphs showing the relationship of various signals described and defined in the preceding equations.

With reference to FIG. 2, the leakage inductance 36 cooperates with the output capacitor 48 to generate an electromagnetic resonance such that at low line-full load, the output voltage VO will be qualified.

Closing the control circuit switch 44 will place the inductance 40 having reactance XL in parallel with the output capacitor 48 having reactance Xc, which will reduce the equivalent capacitive reactance: Xc' = Xcii XL. The new reactance Xc' (<Xc) will reduce gain such that at high line-no load, the output voltage VO will remain qualified. The control circuit 46 will sense the output voltage VO and control the triggering of the control circuit switch 44 so that the output voltage vO is always regulated anywhere between the two extreme cases of low line-full load and high line-no load.

For example, if a silicon controlled rectifier (SCR) is part of the control

circuit 46, controlling the SCR trigger phase angle can be employed to regulate the output voltage vO. Yet, the waveform for ic illustrated in FIG.

3 and defined in equation (2) show that discontinuity in iL also causes ic to be discontinuous. Since the instantaneous value of ic is proportional to the slope of the output capacitor voltage vc, as defined in equation (1), then vc will reflect discontinuity in ic in the form of high total harmonic distortion (see FIG. 3). It follows from equations (2), (3) and (4) that discontinuity in ic and total harmonic distortion is reduced if XM is reduced relative to XL which will reduce discontinuity in iC(t) in the neighborhood of t0 and t1 (see FIG. 5). The reduction in total harmonic distortion is one of the advantages of the present invention which is discussed immediately below.

If XM is to reduce in value in order to reduce total harmonic distortion (see FIG. 5), a proportional reduction inXc is required to maintain the same equivalent reactance of the parallel combination of XM and Xc. The current iM will increase in magnitude if the magnetizing reactance XM is reduced.

A ferroresonant transformer 50, as shown in FIG. 6, exhibits a reduced magnetizing reactance XM caused by a substantially non-magnetic space at 52 which is defined by the space between the E and I shaped members. More specifically, a second longitudinal or free end 53 of a center leg 55 is spaced from an adjacent portion of the second end- connecting portion 20 so as to form the non-magnetic space 52 therebetween. The ferroresonant transformer of FIG. 6, however, tends to exhibit higher acoustic noise and lower efficiency relative to the transformer of FIG. 1. The equivalent electrical circuit 33 of FIG. 2 applies also for the ferroresonant transformer 50 with the exception that magnitude of XM is reduced when the control circuit switch S is opened.

When the control circuit switch 44 of FIG. 2 is open, the current through the output capacitor 48 is limited by XM+XL, where XL<<XM. When the control circuit switch 44 is closed, the current through the output capacitor 48 is only limited by XL. The value of XL is selected so that at high line-no load, iL is high enough to maintain a nominal output voltage vo.

With reference to FIG. 7, some ferroresonant transformer applications are best suited by a square wave voltage output. An example is a DC power supply for television cable systems. The slope of the output voltage is defined by dv(t)/dt=dvc(t)/dt=ic/C. The slope of an output

voltage waveform (labelled vc) must be very high (at point 77 of the vc waveform) near the peak of total capacitor current waveform (point 80 of the ic waveform), and very low near the zero crossing of the ic waveform.

But Ic=(Vc/Xc)+iL, where iL is defined by equations (3) and (4).

The control inductor current IIND (see FIG. 2) is normally kept to a minimum to increase efficiency and reduce cost. Increasing the control inductor current beyond its minimum value will result in a lower duty cycle, dictated by the control circuit, and the output voltage required.

A low duty cycle, high amplitude inductor current will result in a high crest factor for the total capacitor current and contribute to a high voltage rise time. The crest factor is the ratio of the peak value to the root-mean- square (RMS) value of the waveform. As for the portion of the voltage waveform near the peak, the slope has to be a minimum which requires the capacitor current to have a very low amplitude which is usually the case since the control inductor switch is off, and the only inductive current through the capacitor is VC/(XM+XL). To further reduce the capacitor current near the zero crossing, the magnetizing reactance XM is increased by interleaving the laminations when the transformer is assembled. A disadvantage of this method is that a high inductor current IIND (see FIG.

2) will reduce the efficiency of the system because of increased I2R losses in the inductor coil and will require a more expensive AC switch to handle the higher current.

DESCRIPTION OF FIRST PREFERRED EMBODIMENT FIGS 8 and 9 refer to a first embodiment of the present invention. As can be seen in FIG. 8, a ferroresonant transformer 56 has a center leg 58 which includes two physically separate longitudinal portions.

A first longitudinal portion 60 of the center leg 58 extends from the first end-connecting portion 18 and terminates at a first free end 62 about midway between the first and second end-connecting portions 18 and 20, respectively. As can be seen in FIG. 8, the first free end 62 of the first longitudinal portion 60 is slightly closer to the first end-connecting portion 18 than to the second end-connecting portion 20. A second longitudinal portion 64 of the center leg 58 extends from the second end- connecting portion 20 and terminates at a second free end 66 about midway between the first and second end-connecting portions 18 and 20,

respectively. As can be seen in FIG. 8, the second free end 66 of the second longitudinal portion 64 is slightly closer to the second end-connecting portion 20 than to the first end-connecting portion 18.

The first free end 62 of the first longitudinal portion 60 of the center leg 58, and the second free end 66 of the second longitudinal portion 64 of the center leg 58 define a substantially non-magnetic space 68 therebetween. As can be seen in FIG. 8, the air gap 68 is located longitudinally midway along the center leg 58 of the E shaped member between the first and second magnetic shunts 24, 26.

FIG. 9 schematically illustrates an equivalent electrical circuit 70 of the ferroresonant transformer 56 of FIG. 8. Like parts of preceding figures are labeled with like reference numbers and symbols. As shown in FIG. 9, an inductance 72 having reactance XL' represents the control inductance which is positioned differently in the circuit 70 as compared to the inductance 40 having reactance XL of the circuit 33 of FIG. 2. Referring to the circuit 70 of FIG. 9, when the control circuit switch 44 is opened by the control circuit 46, the inductive current through the capacitor is limited by XM. When the switch 44 closes, the current is limited by the <BR> <BR> <BR> parallel combination of XM and XL' such that XM|| XL'=XL, whereby the reactance XL' of the circuit 70 of FIG. 9 will have a higher value relative to the reactance XL of the circuit 33 of FIG. 2. As a result, moving the air gap to the middle of the center leg (as shown in FIG. 8) will result in reducing the inductor current iL as well as reducing the size of the magnet wire necessary to handle the current iIND. Consequently, the size of the transformer can be reduced, and power efficiency can be improved over prior ferroresonant transformers.

DESCRIPTION OF SECOND PREFERRED EMBODIMENT FIG. 11 refers to a third embodiment of the present invention where like elements are labeled by like reference numbers. A ferroresonant transformer 74 includes a center leg 76 defining a slot 78 provided about longitudinally midway along the center leg 76 between the first and second end-connecting portions 18 and 20, respectively. The slot 78 extends through the center leg in a direction transverse to the plane of the figure so as to form a bridged substantially non-magnetic space 78 along the center leg 76. In other words, a non-magnetic space can be

introduced in the center leg without dividing the center leg into two physically separate portions. The slot may be formed by a punching process or the like.

Non-magnetic spaces, such as air gaps, are usually associated with acoustic noise because it is very difficult to keep transformer parts close to the non-magnetic space from vibrating. With the bridged non- magnetic space 78 provided about longitudinally midway along the center leg 76, it is possible to punch a slot in the middle of the center leg, as shown in FIG. 11, and interleave laminations of the center leg 76. The slot defined by each lamination will align to produce the same effect as the non-magnetic space 68 illustrated in FIG. 8. The center leg 76 has greater structural strength in the vicinity of the bridged non-magnetic space 78, as compared to the center leg 58 in the vicinity of the non-magnetic space 68 of FIG. 8. As a consequence, the ferroresonant transformer 74 of FIG. 11 will generate less vibration and related noise relative to the ferroresonant transformer 56 of FIG. 8.

The width of a bridge 79 (i.e., narrow portion of the center leg 76 in the vicinity of the non-magnetic space 78) is limited by the effect its magnetic saturation will have on the voltage output waveform. The bridge 79 will saturate at a certain volt-second and the peak total capacitor current will coincide with that of the switched inductor current, and therefore will have the same effect in increasing discontinuity in the total capacitor current. (See the graph of FIG. 10 illustrating the relationships among vc, iL and ic). Keeping the width of the bridge 79 to a minimum will reduce the effect of saturation on the output voltage level. A practical bridge width is the minimum required for proper handling of the laminations.

DESCRIPTION OF THIRD PREFERRED EMBODIMENT Turning now to FIGS. 12 and 13, some applications of a ferroresonant transformer require a fast rise time voltage waveform. A currently used method is to increase the control inductor current IL. This increased current will require a thicker gauge magnet wire for the control inductor coil and a larger current capacity switch. The increased current will also reduce the efficiency due to increased IL2R losses. Another way to achieve the same effect without increasing the inductor current is to slightly saturate the core. By increasing the bridge width of the core (see FIG. 13 illustrating a wide bridge 82 adjacent a gap 84 defined in a center leg

86) the bridge portion of the center leg will saturate near the voltage zero crossing of the waveform vc which will increase the magnetizing current illustrated by waveform ic in FIG. 12). The peak of the saturation magnetizing current will coincide with the switched inductor current and will have the same effect in increasing the slope of the voltage waveform.

As the voltage waveform reverses, the volt-second of the core will reduce which will bring the bridge portion of the center leg out of saturation and the magnetizing current will reduce to its linear low value (see FIG. 12).

The core losses will increase very slightly because saturation is limited to the bridge, a very small portion of the core. Nevertheless, the reduced control inductor current is significant and will increase efficiency. These principles apply not only for EI laminations, but also for any type of magnetic steel.

While the present invention has been described in several preferred embodiments, it will be understood that numerous modifications and substitutions can be made without departing from the spirit or scope of the invention. Accordingly, the present invention has been described in several preferred embodiments by way of illustration, rather than limitation, and the scope of this patent disclosure shall not be determined primarily from the scope of the appended claims.