DESCRIPTION

The present invention relates generally to radiofrequency bidirectional couplers with power divider/combiner functionality, a kind of devices used in the field of radio frequency engineering that split the energy of a radio-frequency signal propagating through one of the M input among a group of N outputs (divider), which also enables to combine the energy propagating through various inputs into a common output (combiner). This invention discloses a directional coupler structure based on dielectric waveguides that is wideband and broadband. Wideband since it operates over a wide frequency range, and broadband, since is suited for modulated signals occupying a wide bandwidth.

Background of the invention

Radiofrequency (RF) systems operate handling signals in the form of guided electromagnetic waves with frequencies starting around 30 MHz and going up into the millimeter (30 GHz to 300 GHz) and Terahertz (300 GHz to 3000 GHz) regions of the spectrum. In the handling of these signals, RF systems might need to split a radiofrequency signal propagating through a waveguide into several copies of itself, with varying amount (split ratio) of energy within each copy, each delivered at different output waveguides. This function constitutes a power divider. The reciprocal function, combining different RF signals into a common output which constitutes a power combiner, is also of interest.

A particular case of such functionality are passive devices known as bidirectional couplers, which are schematically represented in Figure 1. This figure shows a bidirectional coupler (100) with four access ports. The incoming signal arrives to the directional coupler at its input port (P1), coupling a defined amount of energy from this input port to an output transmission port (P2) and another amount to a second output, known as coupled port (P4). The incoming signal power at the input port (P1) is commonly split between the output ports using transmission lines close to each other, allowing the energy passing through one waveguide to couple into the other by evanescent wave leakage. In some devices, any reflection of the signal exiting through the output port (P2) returning to the device enters at this port (P2), being routed to the isolation port (P3). The device is bidirectional as any port can be the input, which will result in the directly connected port being the transmitted port, adjacent port being the isolated port, and the diagonal port being the coupled port.

Directional couplers and power dividers have many applications. These include providing a signal sample for measurement or monitoring, feedback, combining feeds to and from antennas, antenna beam forming, providing taps for cable distributed systems such as cable TV, and separating transmitted and received signals on telephone lines.

Current directional couplers are mostly limited to the microwave frequency range (3 GHz to 30 GHz) using transmission line designs. Hence, there is a demand to obtain an ultra-wideband directional coupler that covers a wide frequency range, including the range of coaxial standards (such as 1-mm connector, up to 110 GHz) and which can operate independently of the different rectangular waveguide standards used by radiofrequency engineering technology for frequencies above (such as, WR8 -90 GHz to 140 GHz-, WR6 -110 GHz to 170 GHz- and WR5 -140 GHz to 220 GHz-). The present invention satisfies this demand, enabling an extremely wide operation frequency range.

Description of the invention

The present invention proposes a new bidirectional coupler with power divider/combiner functionality based on dielectric waveguide structures to extend the operation frequency range of the device covering the micro (3 GHz to 30 GHz), the millimeter (30 GHz to 300 GHz) and terahertz (300 GHz to 3000 GHz) wave ranges.

The coupler comprises a free propagation region substrate with a pair of opposed edges. Furthermore, the coupler comprises a first group of access ports along a first edge of said opposed edges of said free propagation region substrate and a second group of access ports along a second edge of said opposed edges of said free propagation region substrate. The first and second groups of access ports comprise dielectric waveguide structures providing a high-pass filter transfer function operating over a high frequency range starting from a low cut-off frequency fa _. in the microwave or millimeter-wave ranges.

In a first example, the first edge of said opposed edges may be shaped as an arc of a circle with radius rl whose center OA is located closer to the second edge.

In another example, the second edge of said opposed edges may be also shaped as an arc of a second circle with radius r2 whose center OB is located closer to the first edge.

In another example, the arc of the circles can have the same radius (rl= r2), and the centres, OA and OB, are laid in one of the port axes.

In some examples, the ultra-wideband radiofrequency bidirectional further comprises a plurality of arms, wherein the plurality of arms allocates the first group of access ports and the second group of access ports.

In some examples, the dielectric waveguide structures are established on top of the free propagation region substrate.

In some examples, the dielectric waveguide structures are embedded in the free propagation region substrate.

In a preferred example, the dielectric waveguide “DW” structures comprise tapered ends that comprise a DW tapering profile. The first and second group of access ports further comprise tapered slot antennas “TSA” providing a band-pass filter transfer function, operating over a low frequency range up to a high cut-off frequency fc _H in the millimeter/sub-millimeter wave range.

The tapered slot antennas comprise a TSA tapering profile around a first tapered end of the dielectric waveguide structures wherein the TSA tapering profile matches the DW tapering profile.

In some examples, the TSA tapering profile and the DW tapering profile are linear tapered.

In some examples, the TSA tapering profile and the DW tapering profile are exponential tapered.

In some examples, the TSA tapering profile and the DW tapering profile are FERMI tapered.

In another preferred example, the ultra-wideband bidirectional coupler with power divider/combiner functionality comprises a second substrate on which a low- frequency directional coupler is established, configured to operate from DC (or very low frequency, down to few kHz) up to a low cut-off frequency fa _. in the millimeter wave range. In this structure, a plurality of transmission line and waveguide transitions that connects the low-frequency directional coupler and the tapered slot antennas for achieving a directional coupler with an operating frequency range from low frequency

Up tO fcH.

In another preferred example, the ultra-wideband bidirectional coupler with power divider/combiner functionality comprises two first access ports along the at least one edge of said free propagation region substrate comprising two dielectric waveguide structures having tapered ends and two second access ports along the opposite edge of said free propagation region substrate comprising two dielectric waveguide structures having tapered ends. In this example, the four dielectric waveguide structures comprise a DW tapering profile, and the ultra-wideband bidirectional coupler further comprises four tapered slot antennas providing a low-pass characteristic, operating over a low frequency range up to a high cut-off frequency fc _H in the millimeter wave range, wherein the tapered slot antennas comprise a TSA a tapering profile around a first tapered end of the dielectric waveguide structures wherein the TSA tapering profile matches the DW tapering profile.

In other examples, the dielectric waveguide structures can comprise at least one truncated end.

In other examples, the free propagation region substrate can have a different permittivity than dielectric waveguide structures.

In other examples, the free propagation region substrate can comprise absorbers.

In other examples, the free propagation region substrate can comprise dielectric material.

Brief description of the drawings

For a better understanding the above explanation and for the sole purpose of providing an example, some non-limiting drawings are included that schematically depict a practical embodiment.

Figure 1 shows a directional coupler as is commonly understood by experts in the field of radio-frequency engineering.

Figure 2A shows an ultra-wideband bidirectional coupler according to the present invention.

Figure 2B shows the ultra-wideband bidirectional coupler of figure 2A further comprising tapered slot antennas to launch radio-frequency signals into the dielectric waveguides according to the present invention.

Figure 3 shows a particular example of an ultra-wideband bidirectional coupler according to the present invention having four access ports.

Figures 4A shows a 3D view of the particular example of an ultra-wideband bidirectional coupler shown in figure 3, where the dielectric waveguide access structures are located on top of the free propagation region substrate.

Figures 4B to 4D show 3D views of examples of an ultra-wideband bidirectional coupler according to the present invention where the dielectric waveguide access structures are embedded in the free propagation region substrate.

Figures 5A to 5H show the E-field amplitude distributions at different frequencies in the range from 20 GHz to 300 GHz of the particular example of the ultra-wideband bidirectional coupler shown in figure 3 according to the present invention. Figure 6 shows the simulated S parameter amplitude of the particular example of the ultra-wideband bidirectional coupler shown in figure 3 according to the present invention.

Figure 7 shows another example of an ultra-wideband bidirectional coupler according to the present invention with extension of the operating frequency range towards lower frequencies.

Description of a preferred embodiment

Figure 2A shows a proposed dielectric structure as bidirectional coupler with power divider/combiner functionality (200A) according to the present invention. The proposed structure (200A) comprises a first group of M input access ports in the left in the figure from (P-L1) to (P-LM), which are coupled to the second group of N output access ports on the right in the figure from (P-R1) to (P-RN) via a free propagation region substrate (210). The left edge (240A) of the free propagation region, along which the access ports (P-L1) to (P-LM) are laid along, is shaped by the arc of a circle with a radius rl whose centre OA is closer to the opposite edge (240B) of the free propagation region substrate (210). The edge (240B) of the free space propagation region, along which the output access ports (P-R1) to (P-RN) are laid along, is shaped by the arc of a circle with a radius r2 whose centre OB is closer to the opposite edge (240A) of the free space propagation region.

In a preferred implementation, the arc of the circle for input and output waveguides can have the same radius (rl = r2 = r), and the centres, OA and OB, are laid in one of the port axes.

The access ports (PL1-PLM), (PR1-PRM) comprise dielectric waveguide structures, in the left in the figure from (DW-L1) to (DW-LM), and in the right in the figure from (DW-R1) to (DW-RN), all comprising tapered ends for this particular example. Advantageously, the dielectric waveguide structures couple the electromagnetic energy propagating through them into the free propagation region substrate (210), reducing the insertion losses between ports which are now proportional to the distance between the input and output ports instead of being proportional to the square spacing of the DW structures by a distance d between them which is not constrained by the far-field criterion, wherein: d » 2 D ² / where D is the largest waveguide structure dimension and l is the signal wavelength. However, DW structures emit in a specific region that shifts along its axis for varying signal frequency. Since the phase centre is close to the DW tip at high frequencies, any distance between two radiation zones fits the far-field criterion.

The separation between the access ports (PL1-PLM), (PR1-PRM) is advantageously reduced by the confinement of the electromagnetic energy within the dielectric waveguide, enabling a compact configuration without introducing crosstalk between adjacent access ports.

Figure 2B shows a preferred implementation for another structure (200B). The structure (200B) comprises a plurality of arms that allocate the first group of access ports (PL1-PLM) and the second group of access ports (PR1-PRM). The structure (200B) includes launching structures for the radio-frequency signal in the dielectric waveguide DW, which advantageously excite only the fundamental mode of the dielectric waveguide DW for every frequency of the operating frequency range. The launching structure comprises a tapered end of the dielectric waveguide structure and a tapered slot antenna, both having the same tapering profile. In the figure, the launching structure is shown for all access ports, with tapered slot antennas on the left (TSA-L1 to TSA-LM) and on the right (TSA-R1 to TSA-RN) around the tapered end of the dielectric waveguide structures (DWL1-DWLM), (DWR1-DWRN) having a matching linear tapering profile between the tapered slot antennas and the dielectric waveguide structures.

The dielectric waveguide structures (DWL1-DWLM), (DWR1-DWRN) have a high- pass filter characteristic, enabling the electrical interconnection of radio-frequency signals with frequencies above a low cut-off frequency ( f ). The dielectric waveguide structures (DWL1-DWLM), (DWR1-DWRN) can be designed to have a low cut-off frequency (fa) in the microwave range (i.e. between 3 GHz to 30 GHz) or in the millimeter- wave range (i.e. between 30 GHz to 300 GHz), e.g. at an operating frequency of 60 GHz covering a broad frequency range that extends into the Terahertz wave range (i.e. between 300 to 3000 GHz) and beyond. Preferably, the ultra- wideband bidirectional coupler has a low cut-off frequency (fa) of 65 GHz that can be tuned by modifying the structure dimensions.

The tapered slot antennas (TSA-L1-TSA-LM), (TSA-R1-TSA-RN) have a low-pass filter characteristic, enabling the electrical interconnection of signals from low frequencies up to a high cut-off frequency (†CH) in the millimeter- wave range. The tapered slot antenna can be designed as a transmission line with contact tips at its extreme with which establish electrical contact with the access port of the device and can be designed to operate over a range that starts at 0 Hz and extends up into the millimeter- wave range (i.e., between 30 GHz to 300 GHz, e.g., at an operating frequency of 100 GHz).

As shown in the figure, the tapered slot antennas (TSA-L1-TSA-LM), (TSA-R1-TSA- RN) comprise a tapering profile around a first tapered end of the dielectric waveguide structures (DWL1-DWLM), (DWR1-DWRN). Preferably, the TSA tapering profile and the DW tapering is linear tapered. In other examples, different tapering profiles can be implemented, as e.g. fermi or exponential tapering.

In a preferred embodiment for wideband operation, the tapered slot antennas operate over a frequency range that starts at low frequency and extends above the low cut off frequency of the dielectric waveguide structure (fc _H > fa, e.g., above the 60 GHz of previous example). Preferably, the ultra-wideband bidirectional coupler has a higher cut-off frequency (fen) of, at least, 300 GHz. The cut-off frequency can be increased e.g. by reducing the thickness of the components of the ultra-wideband bidirectional coupler and/or by using materials with different electrical permittivity.

Figure 3 shows another bidirectional coupler new bidirectional coupler with power divider/combiner functionality (300) according to the invention. The ultra-wideband bidirectional coupler (300) has four access ports, i.e. two input ports (P-L1 and P-L3) and two output ports (P-R4 and P-R2). The signal enters to the directional coupler through an input port (P-L1), coupling a defined amount of electromagnetic power to an output transmission port (P-R2) and another amount to a second output, known as coupled port (P-R4). There is a second input port, (P-L3), known as the isolated port.

The access ports of the ultra-wideband bidirectional coupler (300) comprise dielectric waveguide structures with tapered ends, dielectric waveguide structure (DW-L1) in the input port (P-L1), dielectric waveguide structure (DW-R2) in the transmission port (P-R2), dielectric waveguide structure (DW-R4) in the coupled port (P-R4) and dielectric waveguide structure (DW-L3) in the isolated port (P-R1). The dielectric waveguides structures provide a high-pass filter transfer function operating over a high frequency range starting from a low cut-off frequency (fa) in the microwave range or in the millimeter-wave range.

Optionally, the access ports (P-L1), (P-R2), (P-L3) and (P-R4) can include launching structures to inject the signals into their corresponding dielectric waveguides. The launching structures comprise a tapered slot antennas and tapered ends of the dielectric waveguide, the tapers of both structures comprise the same tapering profile. The tapered slot antennas (TSA-L1), (TSA-R2), (TSA-L3) and (TSA-R4) provide a low-pass characteristic, operating over a low frequency range up to a high cut-off frequency (†CH) in the millimeter wave range. The tapered slot antennas comprise a matching pattern defining a tapered coupler, preferably a linear tapering profile around the tapered end of the dielectric waveguide at the access port which together with the corresponding tapered end of the dielectric waveguide achieves an ultra- wideband excitation of the directional coupler in a single-mode regime.

A characteristic of this structure is that allows to control the amount of power coupled from one input port to an output port from the relative angle of their respective locations at the opposite edges of the free propagation region. In figure 3 for the ultra- wideband bidirectional coupler (300), the input port (P-L1) electromagnetic energy is divided between the output ports, (P-R2) and (P-R4).

The maximum power coupling occurs when the dielectric waveguides of an input port and an output port are located along the same axis, as shown in figure 3 for the Input Port (P-L1) and the Transmission Port (P-R2) of the ultra-wideband bidirectional coupler (300). In this situation, the transmitted signal level between ports (P-L1) and (P-R2) is controlled by the (DW-L1) antenna tapering angle e and the distance between the tips of the (DW-L1 , DW-R2) antenna tips, d. Both, smaller e and d lead to larger transmitted signal level.

In the example of figure 3, to achieve an ultra-wideband operating frequency range, the structures (DW-R2) and (DW-R4) may point to the phase centre of the structure (DW-L1). By reciprocity, the DW structures (DW-L1) and (DW-L3) may point to the phase centre of the DW structure (DW-R2). Since the phase centre of this structure varies with frequency along the antenna axis, a trade-off can be established to select the frequency for which the phase front is pointed, which sets the upper frequency limit to the bandwidth of the structure.

As shown in figure 3, for this example, the DW structure (DW-L1) points toward the DW structure (DW-R2). The DW structure (DW-R4) points toward the structure (DW- L1) phase centre at 260 GHz. The length c determines the distance between the phase centre and the tip of the DW structure (DW-L1). In this particular example, the DW structure (DW-L3) points towards the DW structure (DW-R2) phase centre at 260 GHz.

The coupling level between an input port and an output port when these are not in the same axis is controlled by the relative angle between their positions, as shown in figure 3 for the input port (P-L1) and the transmission port (P-R4) of the ultra-wideband bidirectional coupler (300). For this example, the angle a controls the coupling ratio between the input port (P-L1) and the transmission port (P-R4). By reciprocity, the angle b controls the coupling ratio between the output port (P-R2) and the isolation port (P-L3). In the figure, both angles are equal, but a and b can be designed independently to achieve different coupling ratios between ports (P-L3) and (P-R2) with respect to (P-L1) and (P-R4). The coupling ratios are reduced when the angle increases.

In this particular embodiment, the tapered slot antenna (TSA-L3) tapering profile (300a) and the DW tapering is linear tapered, as shown in the zoom of figure 3. However, other tapering profiles can be implemented, as e.g. wherein the tapered slot antenna (TSA-L3) tapering profile (300b) and the DW tapering profile are exponential tapered, or wherein the (TSA-L3) tapering profile (300c) and the DW tapering profile are FERMI tapered. Other examples with different trade-offs are possible, that allows to boost the device specifications in a sub-band of interest or to increase the bandwidth. For avoiding reflections in the dielectric material, absorbers can be placed in the end of the free propagation region (310).

Figure 4A shows a 3D view of ultra-wideband bidirectional coupler (300) when the dielectric waveguides and the free propagation region, which can be realized with materials of equal or different permittivity, are stacked, i.e. the free propagation region (310), the four DW structures (DW-L1, DW-R2, DW-L3, DW-R4) and the tapered slot antennas (TSA-L1, TSA-R2, TSA-L3, TSA-R4).

In another example, an alternative single-layer embodiment is obtained when the DW structures (DW-L1 , DW-R2, DW-L3, DW-R4) are embedded within the free propagation region (310) to obtain a more compact system. The embedding implies any fabrication method that achieves to create differences in the permittivity within the free propagation region to define a dielectric waveguide structure, i.e. for example, either by etching porosities for reducing the permittivity around the DW structures or by assembling parts of different permittivity. In this respect, figure 4B shows a 3D view of ultra-wideband bidirectional coupler (400A), wherein the DW structures (DW-L1 , DW-R2, DW-L3, DW-R4) are embedded in the free propagation region (310) and wherein the DW structures (DW-L1 , DW-R2, DW-L3, DW-R4) are truncated.

Figure 4C shows a 3D view of ultra-wideband bidirectional coupler (400B), wherein the DW structures (DW-L1 , DW-R2, DW-L3, DW-R4) are embedded in the free propagation region (310) and include the launching structure tapered slot antennas (TSA-L1, TSA-R2, TSA-L3, TSA-R4) at the access ports, wherein the tapered slot antennas are established on the dielectric material of the free propagation region.

Figure 4D shows a 3D view of ultra-wideband bidirectional coupler (400C), wherein the free propagation region (310) comprises the same dielectric material as the DW structures and wherein the DW structures have a truncated end.

The operating characteristics of the of ultra-wideband bidirectional coupler (300) of figures 3 and 4A have been characterized through full-wave simulations. The obtained E-field amplitude distributions at different frequencies in the range from 20 GHz to 300 GHz are shown in Figures 5A to 5H. As it can be appreciated through these figures, most of the power of an incoming signal at a first port (P-L1) travels through the structure to a second port (P-R2) in a single-mode regime. A small fraction of the incoming power is diverted towards port (P-R4). As shown in the figures, a smaller amount of power of the signal arrives to port (P-L3). Due to the single-mode regime, the phase between ports can be univocally defined, which allows its use for instrumentation purposes as shown in figure 1.

Figure 6 shows the simulated S parameter amplitude (in dB): S11 (601) (incoming port matching), S21 (602) (transmission between port (P-L1) and (P-R2)), S13 (603) (coupling between the incident port P-L1 and the isolated port P-L3), and S14 (604) (coupling to the coupled port (P-R4)).

The transmission between ports (P-L1 , P-R2), i.e. S21 (602) stays flat from 65 GHz to at least 300 GHz. The insertion losses are, approximately 4 dB. The coupling (602) between ports (P-L1, P-R4), i.e. S14 (604) is not constant with frequency, leading to a higher coupler directivity at higher frequencies. Due to the smoothness of the curves, this effect can be easily compensated through a path calibration. The S11 (601) amplitude port matching is lower than -15 dB for the whole band, and lower than -20 dB for frequencies greater than 80 Ghz. The isolation between ports (P-L1 , P-L3), i.e. S13 (603) is greater than 25 dB in the whole band.

The ultra-wideband coupler (300) works as a bidirectional coupler. When port (P2) works as the source of incident signal, por (P-L1) works as the transmission port and port (P-L3) works as the coupled port, while port (P-R4) works as the isolated port. The matching, transmission, coupling and isolation parameters can be the same as in Fig. 6, e.g. S22=S11 (601), S12=S21 (602), S32=S41 (604), and S42=S31 (603).

If a transmitter device is connected to port (P-L1) and two receivers to ports (P-L3, P- R4), wherein (P-L4) is optional, port (P-R2) becomes a bi-directional (input and output) port. In a realistic scenario, port (P-R2) would be connected to an antenna, a waveguide ora connector. In a communication application, an antenna may be placed on port (P-R2). In instrumentation applications, port (P-R2) would be connected to the DUT (device under test). The signal received in port (P-L3) would be proportional to the DUT incident signal and the signal received in port (P-R4) would be proportional to the signal incident in the DUT.

Figure 7 shows a coupler (700) that works with a DC extension, which is a solution to extend the operating frequency range of the bidirectional coupler (300) structure in the low frequency range, towards DC. The coupler (700) comprises the bidirectional coupler (300) and further comprises a low-frequency directional coupler (750). The low-frequency directional coupler (450) works for frequencies up to fo= 65 GHz. The low-frequency directional coupler (450) comprises ports (P-LT, P-R2’, P-L3’, P-R4’) that are connected to the transmission structures (TSA-L1 , TSA-R2, TSA-L3, TSA- R4) ends through metal wires that conforms bifilar lines (720) that can be used for exciting the DW ^' s fundamental mode for all the frequencies above fo and suitable for the integration of the low-frequency directional coupler (700).

For frequencies above fo, the signals are efficiently coupled to the DW structures (DW- L1, DW-R2, DW-L3, DW-R4) from the ports through the structures (TSA-L1, TSA-R2, TSA-L3, and TSA-R4), as illustrated in figures 5 and 6. For frequencies below fo, the signal propagates from the ports through a structures TSA though the bifilar lines (720) without being coupled to the DW structures. One of several transitions can be incorporated from the bifilar line (720) to the low-frequency directional coupler ports (P-LT, P-R2’, P-L3’, and P-R4’). For illustrative purposes, figure 7 shows a low- frequency directional coupler (700) with CPW ports and the transitions between the bifilar lines (720), the CPS and the CPW waveguides.

The transitions and the low-frequency directional coupler (750) are placed in a substrate (710) that can be placed far enough over (or under) the dielectric coupler (300). Since the waves propagates in the 2D plane (in the free propagation region (210)), there is no radiation in the normal direction a compact configuration can be achieved. The distance between both couplers (300, 750) must be big-enough for avoiding near-field coupling between them.