CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims priority of provisional application 63/296,745, filed in the United States on Jan. 5, 2022, the entire contents of which are hereby incorporated by reference.

This application claims priority of provisional application 63/373,983, filed in the United States on Aug. 30, 2022, the entire contents of which are hereby incorporated by reference.

BACKGROUND OF THE INVENTION

1. Field of Invention

This invention relates to electrical field sensing and measurement and, in particular, to an electrometer, a non-invasive voltmeter, and a method of reducing output drift of an amplifier, which method may be employed to improve the performance of an electrometer and a non-invasive voltmeter.

2. Description of Related Art

Electrometers are electrical instruments known to measure electric charge or electrical potential difference. Voltmeters are electrical instruments for measuring electrical voltage. Non-invasive voltmeters do not require direct contact with a conductor exhibiting the voltage being measured, but are typically capacitively coupled to the conductor instead. The sensitivity of an electrometer or voltmeter can be increased by increasing the input impedance of the electrometer or voltmeter. However, excessive input impedance can de-stabilize an electrometer or voltmeter leading to measurement inaccuracy.

United States patent No. 6,922,059 to Zank et al. discloses the use of a large resistance at the input of a sensor for detecting electric field disturbances, as a compromise between high input impedance and stability. Thus, the need for stability reduces the achievable input impedance of the sensor to Zank et al.

United States patent application publication No. 2009/0309605 to Prance et al. and Patent Cooperation Treaty (PCT) international patent application publication No. WO 03/048789 to Clark et al. discloses several other known techniques for increasing the input impedance of an electrometer, namely, a guard circuit, a bootstrapping circuit, a neutralisation circuit, supply rail drift correction, supply modulation, and offset correction.

The guard circuit of Prance et al. and Clark et al. alleviates stray capacitance by positive feedback maintaining the same potential on a guard or shield physically surrounding input circuitry, including an input detection electrode, as on the input detection electrode. However, the presence of the guard or shield physically impedes access to the input detection electrode.

The bootstrapping circuit of Prance et al. and Clark et al. also employs positive feedback to an amplifier by splitting an input bias resistor into two resistors and adds a capacitor connected from between the two resistors and the amplifier output. However, the bootstrapping circuit requires use of the input bias resistor, which lowers input impedance, and acts as a highpass filter that prevents the amplifier from operating below a frequency associated with the filter's time constant.

The neutralisation circuit of Prance et al. and Clark et al. also employs positive feedback to an amplifier by connecting a capacitance directly between the output and input of the amplifier. Further resistors and a potentiometer are used to set the neutralisation to a desired level. However, the neutralisation circuit of Prance et al. and Clark et al. requires multiple operational amplifiers for the amplifier, which increases complexity of the overall amplifier. Clark et al. discloses that a low-leakage photodiode or a neon lamp coupled to a source of illumination (to control the resistance presented by the photodiode or neon lamp) may be employed in place of the input resistor used in guarding, bootstrapping, and neutralisation circuits.

However, all forms of positive feedback have the inherent disadvantage that introducing positive feedback to an amplifier lowers the stability of that amplifier.

Clark et al. discloses that supply rail drift correction involves incorporating into the power supply a de feedback loop that responds to the output drift at the output of an amplifier, such that the supply rails are moved in order to counteract the drift. However, the supply rail drift correction requires customization of the power supply used to produce the supply rails for the amplifier.

Clark et al. discloses that supply modulation (i.e. the bootstrapping of device capacitance) may be used as an alternative to neutralisation to reduce input capacitance of an amplifier. Thus, supply modulation carries similar or analogous disadvantages to those of neutralisation in addition to requiring customization of the power supply.

Clark et al. discloses that the offset correction technique presents an offset correction signal via an offset voltage adjustment pin of the integrated circuit (IC) chip constituting the amplifier. However, this technique is only applicable to IC amplifiers having an offset voltage adjustment pin.

Measuring devices such as DC voltmeters are widely used in the automotive, marine and aerospace service industry to measure and indicate the electrical potential or voltage present on various electrical conductors that are part of electrical or electronic circuits. These voltages can span from less than 1 volt to as much as 1000 volts or more. A typical measurement accuracy lies in the range of 1 % to 3%.

Conventional DC voltmeters require that ohmic contact be established with the electrical conductor whose potential is to be measured. Unfortunately, this requires that the conductor is either exposed by stripping back its insulation so that contact can be made, or the conductor is contacted by the piercing of its insulation by a sharp probe. This is undesirable as it leads to ingress of moisture and can lead to the failure of the conductor. In fact, it is well known in the electrical service industry that insulation should not be pierced for measurements, yet hitherto that has been the necessary method.

Accordingly, there is a great need to measure the DC voltage on electrical conductors through the intervening insulation without making ohmic contact with the electrical conductor. Achieving this objective has hitherto been elusive, but continues to be needed today in that it would greatly enhance the servicing and reliability of electronic and electrical circuits in the automotive, marine, aerospace and other industries where a preponderance of DC circuits exist.

An object of the invention is to address the above shortcomings.

SUMMARY

The above shortcomings may be addressed by providing, in accordance with one aspect of the invention, an electrometer that includes an amplifier exhibiting output drift. The electrometer includes: (a) the amplifier that includes a floating input terminal; (b) a high-impedance current source electrically connected to the floating input terminal to provide an electrical current from the current source to the floating input terminal; and (c) a controller operable to control the electrical current so as to counteract the output drift.

The high-impedance current source may include a first current source for providing the electrical current at a first polarity. The high-impedance current source may include a second current source for providing the electrical current at a second polarity opposite to the first polarity. The first current source and the second current source may be connected in an anti-parallel configuration. The first and second current sources may include first and second phototubes, respectively. A first optical source may be in optical communication with the first phototube. A second optical source may be in optical communication with the second phototube. The first phototube may be operable to apply the electrical current at the first polarity in response to the first light source being actuated. The second phototube may be operable to apply the electrical current at the second polarity in response to the second light source being actuated. The controller may be a microcontroller. The microcontroller may be operable to selectively actuate the first and second light sources in response to the output drift. The first and second light sources may be first and second light-emitting diodes, respectively. The electrometer may further include a first light shield encompassing the first phototube and the first light source. The electrometer may further include a second light shield encompassing the second phototube and the second light source. The amplifier may be an operational amplifier. Each of the first and second current sources may be selected from the group consisting of: (i) a photodiode; (ii) a gas-filled chamber containing a first electrode and a second electrode distal from the first electrode, the first electrode being an emitter of ionizing radiation; (iii) a neon bulb; and (iv) a high-vacuum reed switch. At least one of the first and second current sources may be a photodiode. At least one of the first and second current sources may be a gas- filled chamber. The gas-filled chamber may contain a first electrode and a second electrode distal from the first electrode. The first electrode may be an emitter of ionizing radiation. At least one of the first and second current sources may be a neon bulb. At least one of the first and second current sources may be a high-vacuum reed switch.

It is an object of the present invention to provide for an electrometer capable of measuring electric field gradients without the need for a ground reference.

It is a further object of the present invention to provide for an electrometer that exhibits ultra-high input resistance and impedance. It is another object of the present invention to provide for an electrometer that presents minimal self-disturbance to the electric field to be probed.

It is also an object of the present invention to provide for an electrometer that is able to cancel out (balance) the various leakage currents present at its sensing input.

It is still a further object of the present invention to provide for an electrometer that exhibits minimal drift over time.

And it is another object of the present invention to provide for an electrometer that exhibits minimal input capacitance.

In accordance with another aspect of the invention, there is provided a voltmeter for non-invasively measuring voltage on a conductor. The voltmeter includes: (a) the aforementioned electrometer; (b) an excitation source operable to apply an excitation signal to an excitation terminal, the excitation signal being a step excitation, the controller of the electrometer being operable to determine, in response to the step excitation, a step change in the output of the amplifier; and (c) a relay for connecting and disconnecting the voltmeter to and from an electrical ground associated with the conductor.

The excitation terminal and the floating input terminal may be displaced a fixed distance from each other. The voltmeter may further include a first dielectric shroud attached to the floating input terminal. The voltmeter may further include a second dielectric shroud attached to the excitation terminal.

In accordance with another aspect of the invention, there is provided a computerized method of correcting output drift of an amplifier having a floating input terminal. The method involves: (a) applying a predetermined non-floating input to the floating input terminal of the amplifier; (b) disconnecting the predetermined non-floating input from the floating input terminal; (c) determining by a computerized controller a deviation of the output voltage of the amplifier from a predetermined output value associated with the predetermined nonfloating input; and (d) in response to the deviation, applying an electrical current from a high-impedance current source to the floating input terminal so as to minimize the deviation.

The method may further involve applying a step excitation at an excitation terminal of a non-invasive voltmeter that includes the electrometer. The method may further involve determining the output voltage of the amplifier in response to the step excitation. Determining the output voltage of the amplifier in response to the step excitation may involve determining the output voltage when a relay of the non-invasive voltmeter is open. The method may further involve determining a loss factor in response to a magnitude of the step excitation and the output voltage. Determining the loss factor may involve dividing the magnitude of the step excitation by the output voltage. Determining the output voltage of the amplifier in response to the step excitation may involve determining the output voltage when the relay is closed. The method may further involve determining a scaled voltage in response to the output voltage and the loss factor. Determining the scaled voltage may involve multiplying the output voltage by the square-root of the loss factor. The method may involve correcting the output drift when the amplifier is an operational amplifier. Step (a) of the method may involve applying the predetermined non-floating input until the output of the amplifier settles to a predetermined output value.

In accordance with another aspect of the invention, there is provided an electrometer, which includes: (a) means for applying a predetermined nonfloating input to the floating input terminal of the amplifier; (b) means for disconnecting the predetermined non-floating input from the floating input terminal; (c) means for determining by a computerized controller a deviation of the output voltage of the amplifier from a predetermined output value associated with the predetermined non-floating input; (d) means for applying, in response to the deviation, an electrical current from a high-impedance current source to the floating input terminal so as to minimize the deviation; and (e) means for determining the output voltage of the amplifier. In accordance with another aspect of the invention, there is provided a voltmeter for non-invasively measuring voltage on a conductor. The voltmeter includes: (f) the aforementioned electrometer; (g) means for applying a step excitation at an excitation terminal of the voltmeter, wherein the means for determining the output voltage of the amplifier is operable to determine a response to the step excitation; and (h) means for connecting and disconnecting the non-invasive voltmeter from an electrical ground associated with the conductor.

In accordance with another aspect of the invention, there is provided a voltmeter that can ascertain the direct current (DC) voltage on an electrical conductor located within a dielectric sheath (insulation), wherein firstly the voltmeter is electrically floating (ungrounded with respect to the electrical conductor) and wherein two capacitive structures abut the electrical conductor, and wherein their respective capacitances are a function of capacitive structure size, their distance to the conductor, conductor diameter and the insulator dielectric constant, and wherein by coupling a reference voltage waveform from the first capacitive structure to the second capacitive structure the overall capacitance of the structure will be determinable thereby providing for the derivation or other determination of a correction factor independent of any electrical potential on the conductor, and wherein secondly the present invention is then grounded with respect to the conductor and the reference voltage waveform amplitude coupled across the two capacitive structures will be offset due to any DC potential on the conductor, and wherein by applying the derived correction factor, the true DC potential or voltage on the electrical conductor will be measured without requiring ohmic contact.lt is an object of the present invention to provide for a DC voltmeter that can measure DC electrical potentials on an electrical conductor without requiring ohmic contact thereto. It is another object of the present invention to provide for a non-contact DC voltmeter that drastically reduces the electrical circuit measurement interval by not requiring conductor exposure or insulation piercing.

It is a further object of the present invention to provide for a non-contact DC voltmeter that greatly increases electrical circuit reliability by not requiring conductor exposure or insulation piercing.

It is another object of the present invention to provide for a non-contact DC voltmeter that greatly increases operator safety in higher voltage electrical circuits by not requiring conductor exposure or insulation piercing.

It is yet another object of the present invention to provide for a non-contact DC voltmeter that minimizes the loading or influence on delicate electrical circuits by not requiring ohmic contact with the conductor.

The foregoing summary is illustrative only and is not intended to be in any way limiting. Other aspects and features of the present invention will become apparent to those of ordinary skill in the art upon review of the following description of embodiments of the invention in conjunction with the accompanying figures and claims.

BRIEF DESCRIPTION OF THE DRAWINGS

In drawings which illustrate by way of example only embodiments of the invention:

Figure 1 is an electrical circuit schematic of an electrometer in accordance with a first embodiment of the invention, showing a floating input terminal and anti-drift circuitry;

Figure 2 is an electrical circuit schematic of a variation of the electrometer of Figure 1 , showing anti-drift circuitry being separated into two different parts; Figure 3 is an electrical circuit schematic of a variation of the electrometer of Figure 2, showing a high-vacuum reed switch;

Figure 4 is an electrical circuit schematic of a variation of the electrometer of Figure 3, showing a simplification of one part of the anti-drift circuitry;

Figure 5 is an electrical circuit schematic of a variation of the electrometer of Figure 4, showing a pair of high-vacuum reed switches;

Figure 6 is a block diagram of one electrometer of one of Figures 1 to 5 embedded in a system operable to wirelessly transmit measurement results of the one electrometer;

Figure 7 is a screenshot of an output display showing measurements, by the one electrometer of Figure 6, of a 3Hz, 10V square wave excitation stimulus;

Figure 8 is a screenshot of an output display showing measurements by a prior art oscilloscope under conditions similar to those of Figure 7, showing a capacitive response;

Figure 9 is a screenshot of an output display showing measurements, by the one electrometer of Figure 6, of human heart activity as the excitation stimulus;

Figure 10 is a screenshot of an output display showing measurements, by the one electrometer of Figure 6, of human body movement as the excitation stimulus;

Figure 11 is an electrical circuit schematic of a non-invasive voltmeter employing principles of the electrometer of Figure 1 , showing an excitation terminal;

Figure 12 is a flow diagram showing steps of a method of correcting output drift of an amplifier in accordance with embodiments of the invention; Figure 13A is a flow diagram of a first portion of a method of non-invasively measuring voltage of a conductor in accordance with embodiments of the invention;

Figure 13B is a flow diagram of a second portion of the method of Figure 13A; and

Figure 13C is a flow diagram of a third portion of the method of Figures 13A and 13B.

DETAILED DESCRIPTION

An electrometer includes: (a) means for applying a predetermined nonfloating input to the floating input terminal of the amplifier; (b) means for disconnecting the predetermined non-floating input from the floating input terminal; (c) means for determining by a computerized controller a deviation of the output voltage of the amplifier from a predetermined output value associated with the predetermined non-floating input; (d) means for applying, in response to the deviation, an electrical current from a high-impedance current source to the floating input terminal so as to minimize the deviation; and (e) means for determining changes in the output of the amplifier separate from the deviation.

A non-invasive voltmeter includes the aforementioned electrometer and means for applying a step excitation at an output terminal of the non-invasive voltmeter, wherein the means for determining changes in the output of the amplifier separate from the deviation is operable to determine a response to the step excitation.

Referring to Fig. 1 , the electrometer according to a first and preferred embodiment of the invention is shown generally at 10. The electrometer 10 is operable to detect and measure electric field potential and disturbances thereto. Also, the electrometer 10 has a front end capable of nullifying leakage currents and charge creepage, and exhibiting an ultra-high input impedance, thereby allowing for the detection and measurement of electric field potentials without reference to ground.

Fig. 1 shows a vastly improved electric potential sensor for the non- invasive and non-contact measurement of electric field (E-field) potentials and disturbances thereto (events) and distortion therein/thereof. The sensor comprises at least one detection electrode arranged for probing the electric field and for generating a signal which is fed to a novel sensor amplifier adapted to receive the signal at its input and to supply a detection signal at its output, while providing an inherent ultra-high input impedance (>1-5PQ), minimal input capacitance, minimal drift and offset, all without the use of bootstrapping (positive feedback) and the negative effects common thereto (i.e. , it is a true impedance converter.)

The electrometer 10 of Fig. 1 is applicable to non-contact I stand-off electric field potential sensing and measurement, such as, but not limited to, earth electric field measurements, the detection of distortions in and disturbances to natural and artificial reference electric fields, bioelectrical signal detection, neurological electrical signal detection, materials detection and classification and discrimination, movement detection, the detection of leakage and creep currents, ionization product detection, ion currents and electric field gradients in gasses etc.

The electrometer 10 of Fig. 1 is a highly sensitive electrometer without a ground reference. This electrometer 10 enables long-range detection of metallic objects, plastics, and other semi-conductive materials. This electrometer 10 also enables the measurement of electric potentials and potential gradients with no reference to ground. The electrometer 10 uses an internal reference point, and thus is no longer bound to a reference point in the system. This enables the electrometer 10 to actively measure the state of a system, as well as time dependant changes therein. With no ground reference the electrometer 10, which may be referred to herein as a Floating Electric Potential Sensor (FEPS), is able to actively measure any electric field gradient. Due to the properties of conductors, semi-conductors and insulators they produce disturbances in the immediate field around them. As a highly sensitive field sensor, the FEPS is able to detect these minute variations in the electric field. The FEPS is also able to detect sources of electricity which cause distortions to the ambient field. This enables the FEPS to act as an energy potential seeker. If two FEPS are utilized (differential mode) in combination, an electric field gradient can be measured, without reference to earth ground, and the corresponding gradient vector can be ascertained.

The electrometer 10 of Fig. 1 includes a microcontroller 560 which is powered from the output 550 of voltage regulator 540. The regulator 540 is fed via line 530 from battery 510, whose negative is shown tied to system ground in this case.

Also shown is an analog output 600 from microcontroller 560 via line 570, allowing for the connection of a multimeter, oscilloscope, data logger or the like across terminals 580 and 590.

Further shown, is an RF transceiver 620 that can communicate with the microcontroller 560 via line 610. The transceiver 620 connects to antenna 640 via line 630 and may be Bluetooth, BLE, Zigbee or any other type suitable for a particular application.

A sensing electrode 650 is connected to the “floating” high-impedance positive input 660 of op-amp 670. Op-amp 670 is of a suitable type of operational amplifier such as the Texas Instruments LMP7721 . The output 680 is tied to the negative input to create a voltage follower and is also fed into an analog input on microcontroller 560 for digitization.

The embodiment of Fig. 1 includes anti-drift circuitry to prevent the output of the op-amp from drifting over time until it railed either positively or negatively.

As shown the positive input of the op-amp 670 is also connected via line 690 to two back-to-back connected phototubes 700 and 750. The phototubes may be of a miniature Hammamatsu R6800U type or of a custom construct, being optically and electrostatically shielded in any event.

Each of these two phototubes 700 and 750 has a light-emitting diode associated with it, 720 driven by line 710, and 740 driven via line 730. These two light-emitting diodes 720 and 740 are situated within a light-tight enclosure (not visible in Fig. 1 ) such that the respective light-emitting diodes can illuminate their respective photocathodes of the phototubes 700 and 750 without interference from ambient or other stray light.

Line 760 connects the phototubes other common connection to a midpoint reference created via resistors 770 and 780. Typically, the resistors 770 and 780 have the same value such that the voltage on line 760 is the mid-point voltage of the op amp 670, which is midway in value between the supply rail voltages of the op amp 670 (e.g. midway between V+ and ground).

As can be seen, via appropriate pulsation, modulation or switching of the light emitting diodes via control action from microcontroller 560, electrons may be precisely injected or removed from the positive input 660 of op-amp 670. This across/through a virtually infinite impedance vacuum space within the phototubes cavity, hence also minimizing leakage and creepage.

Variations of the embodiment of Fig. 1 are shown in Figs. 2 to 5. In particular, Fig. 2 shows, within block 10, a schematic diagram of a second embodiment of a single-channel electric field sensor or electrometer.

Furthermore, Fig. 2 shows an electric field probe/electrode 11 , connected via 12 to the positive input of a suitable amplifier (i.e. , electrometer grade amplifier) 13, such as the commercially available National Semiconductor LMP7721 for example. This type of amplifier exhibits an ultra-high input impedance (>100 teraohms) by virtue of having a well-designed MOSFET front end. Not shown are various guarding arrangements and leakage/creep path reduction means that are well-known in the art. The amplifier 13 is typically configured as a simple non-inverter for the specific purpose of presenting as high a possible impedance at its positive input, yet providing a low-impedance output at 17.

As shown, the amplifier 13 has its output 17 fed back to its negative input

14 via feedback resistor (Rf) 16. Also connected to the negative input 14 is an input resistor (Rin) 15. The selection of resistance values for the resistors 16 and

15 provide selectable gain. Input resistor (Rin) 15 also is connected to a variable voltage source that includes a digital potentiometer such as the EPOT 18 shown in Fig. 2 under microcontroller command 19 (microcontroller not fully shown for ease of illustration). In variations, the EPOT 18 may be replaced by a digital-to- analog converter (DAC) or the input resistor (Rin) 15 may be directly connected to a DAC output of the microcontroller.

In one embodiment, the feedback resistor 16 and the input resistor 15 have the same value, thereby providing a double gain amplifier similar to a unitygain voltage follower. Also shown in Fig. 2 is an analog-to-digital converter 20, which serves to digitize the analog output 17 from the amplifier 13, and provide digitized data to a microcontroller (microcontroller again not fully shown.)

Further, as so far described, one can see that a varying offset voltage can be imposed on negative input 14, allowing for intelligent, i.e. , software controlled displacement of the amplifier’s null point. This, without affecting the positive input’s mean impedance.

Still referring to Fig. 2, it is assumed that the amplifier 13 is subject to charge creep and leakage paths in or out of the ultra-high impedance positive input, resulting in at least some uncontrollable drift of the potential at the positive input, which with careful design and physical execution can be made to occur very slowly, however, the timescale to the amplifier 13 railing (seconds to minutes) will still be of too short a duration as to render it practical. In addition to the above drift due to the creep and leakage, and summed with the creep and leakage induced drift, are device internal bias current leakages and MOSFET gate leakage currents.

These are substantially higher than the leakage and creep currents, and especially in the case of the MOSFET gate leakage, are highly temperature dependant. For example, parametric data shows the example NS amplifier 13 (LMP7721 ) to have +/- 3fA at 20 degrees C, rising to as much as +/- 900fA at 85 degrees C.

Connecting a costly high-value resistor between the positive input and system ground (as is the usual prior-art practice) in order to counteract the sum total of drift inducing mechanisms, unfortunately lowers the input impedance that might have otherwise been attained.

If just as much charge was to be injected or withdrawn from the positive input of amplifier 13, as was occurring via the sum of the various drift inducing mechanisms mentioned so far, one could negate amplifier drift entirely.

The present invention performs the same by, in the embodiment shown in Fig. 2, utilizing a small, low-cost inert gas-filled chamber 22 containing two electrodes (similar to a small neon bulb in construct.) One of the electrodes 24, is treated to become an energetic emitter of ionizing radiation (e.g., thoriated), thereby ensuring a sufficient number of gas ionization events per unit time.

Further, the chamber 22 is shielded against light (electromagnetic radiation) ingress via a shield 23 (and also against ambient radiation ingress by using lead as the shielding material), hence ensuring that the only remaining ionization events in the gas-filled chamber 22 are due to the action of the energetic electrode (and the occasional cosmic ray.)

The chamber 22 can be theoretically modelled as a very high-value (but also variable) resistor. As shown in Fig. 2, a drift balancing charge flow can be injected through the chamber 22, via the application of a variable potential 25, from an EPOT 26 (or DAC) under microcontroller command 27 (microcontroller again not fully shown.)

If the voltage at 25 is very close to the potential at 12 (i.e., <2V), then the fairly high resistance of the chamber 22 (i.e., @500 gigaohms @20V) will seem much, much larger, i.e., on the order of petaohms, due to the highly non-linear resistance behaviour thereof.

And it is in this state that the sensor spends the majority of its time, with the quiescent leakage, creep and bias currents being nullified through the abovementioned action. This presents an almost virtually infinite input impedance/resistance (>1-5PQ), at the positive input 12 of amplifier 13.

With reference to Fig. 3, there is shown a schematic diagram of a further embodiment of the present invention. It is similar to the preferred embodiment described in Fig. 3 above with the addition a high-vacuum reed switch 28 (virtually infinite impedance when open) that is controlled by a microcontroller via 29 (microcontroller not fully shown.)

Closing the switch 28 serves to short circuit the connection/path normally provided through chamber 22, thereby allowing for a much rapid bias adjustment at the positive input of the amplifier.

Now with reference to Fig. 4, there is shown a schematic diagram of another embodiment of the present invention. It is similar to the preferred embodiment described in Fig. 3 above, except that the simple gas-filled chamber seen in previous embodiments has been replaced by a specially constructed gas-filled reed switch 28, having one of its leads being treated at a suitable location to be an energetic emitter of ionizing radiation (as in the gas-filled chambers described above.)

The single reed switch 28 thereby behaves like, and replaces the previously described combination use of a gas-filled chamber and a separate high-vacuum reed switch. This serves to lower circuit complexity and more importantly, further reduces leakage/creep charge flow paths in some critical applications.

With reference now to Fig. 5, there is shown a schematic diagram of yet another embodiment of the present invention. It is similar to the preferred embodiment described in Fig. 4 above, but with the addition of a high-vacuum reed switch 32, which is controlled by a microcontroller (again not fully shown) via 34, and which serves to connect a variable voltage from an EPOT 33 (or DAC) under microcontroller command 35 (microcontroller not fully shown) to the positive input 12 of the amplifier 13 when desired, in order to force a rapid potential change on the sensing electrode. Further, Rf 30 is shown to be of an EPOT type and serves to provide variable gain by a microcontroller (again not fully shown) via 31 .

Referring now to Fig. 6, there is shown a schematic block diagram of the present invention as utilized in a typical application. The sensor FEPS (Floating Electrical Potential Sensor) 400 is representative of the electrometer 10 of Fig. 1 and the alternative embodiments of Figs. 2 to 5. The FEPS 400 is shown connected to microcontroller 407 via lines 406 for control and transfer of data representative of the electric field potential in which probe electrode 401 is immersed. The probe electrode 401 of Fig. 6 is representative of the sensing electrode 650 of Fig. 1 and the electric field probe/electrode 11 of Figs. 2 to 5.

Also shown in Fig. 6 is a suitable power supply 403, which is shown here deriving power from a battery 404 and providing power to the FEPS 400 via lines 402 and to the microcontroller 407 via lines 405.

Shown lastly, is a suitable RF transceiver 409 connected to the microcontroller 407 via lines 408, which serves to communicate the FEPS sensed data to a remote location via EM radiation 411 .

With reference to Fig. 7, there is shown a screenshot illustrating the sensitivity and response of an embodiment of the present invention to an electrical stimuli/excitation signal (in this case a 3Hz, 10 volt square wave) coupled through bone dry sand (note the general preservation of waveform due to the FEPS ultra-high input impedance associated with the op-amp 670 of Fig. 1 and the amplifier 13 of Figs. 2 to 5.)

Now shown in Fig. 8 is a screenshot illustrating the response of a reference oscilloscope under the same conditions (3Hz, 10 volt square wave coupled through bone dry sand) as described in Fig. 7 above (note the pure capacitive response shown in Fig. 8.) The reference oscilloscope of Fig. 8 has impaired performance relative to the superior performance of the embodiments of the present invention of Figs. 1 to 5.

With reference to Fig. 9, there is shown a screenshot illustrating the sensitivity and response of embodiments of the present invention to minor electrical stimuli (in this case, the human heart, ECG at a stand-off distance of @25mm from the FEPS sensor probe 401 of Fig. 6.)

Now with reference to Fig. 10, there is shown a screenshot illustrating the sensitivity and response of the present invention to minor physical stimuli (in this case, the movement of the big toe in a leather shoe at 1 meter from the FEPS sensor probe.) In Fig. 10, the lower (red) waveform results from unity gain measurements, and the upper (blue) waveform results from measurements at a gain of 20 V/V.

Thus, there is provided an electrometer that includes an amplifier exhibiting output drift. The electrometer includes: (a) the amplifier that includes a floating input terminal; (b) a high-impedance current source electrically connected to the floating input terminal to provide an electrical current from the current source to the floating input terminal; and (c) a controller operable to control the electrical current so as to counteract the output drift.

Non-invasive Voltmeter

The principles of the electrometer 10 of Fig. 1 and the alternative embodiments of Figs. 2 to 5 may be suitably employed in a non-invasive voltmeter, such as the non-invasive DC (Direct Current) voltmeter 1100 shown in Fig. 11 . Shown in FIG. 11 is a portable handheld embodiment of the present invention. The voltmeter 1100 of Fig. 11 is operable to non-invasively measure the voltage, if any, present on the conductor 240 without making direct ohmic contact with the conductor 240. Thus, the voltmeter 1100 can advantageously determine the voltage on conductor 240 without any need to pierce, cut or otherwise impair the insulation 230 that surrounds the conductor 240.

The voltmeter 1100 of Fig. 11 is operable to determine the electrical potential or voltage on electrical conductors without the requirement for ohmic contact to the electrical conductor. In particular, the voltmeter 1100 can ascertain the direct current (DC) voltage on an electrical conductor located within a dielectric sheath (insulation), wherein at first the voltmeter 1100 is electrically floating (ungrounded with respect to the electrical conductor) and wherein two capacitive structures are created incorporating the electrical conductor, and wherein their respective capacitances are a function of capacitive structure size, their distance to the conductor, conductor diameter and the insulator dielectric constant.

By coupling a reference voltage waveform from the first capacitive structure to the second capacitive structure the overall capacitance of the structure will be determined thereby providing for the derivation of a correction factor independent of any electrical potential on the conductor.

When secondly the voltmeter 1100 is then grounded with respect to the conductor, the reference voltage waveform amplitude coupled across the two capacitive structures will be offset due to any DC potential on the conductor, and whereby applying the derived correction factor, the true DC potential or voltage on the electrical conductor will be measured without requiring ohmic contact.

Referring to Fig. 11 , the battery 10 can be seen to have its negative tied to system ground via 20. Its positive terminal feeds voltage regulator 40 via 30. The out put 50 of voltage regulator 40 provides the regulated system positive rail. Further shown is microcontroller 60 which can be any one of numerous available. uC 60 has a digital output 70 that feeds a relay drive circuit for relay 80. Relay 80 has a normally open contact 90 which is connected to system ground and when closed, can be seen to allow for the ohmic connection of system ground to an external circuit ground via 110 as will be described later.

While Fig. 11 shows a relay 80, any component or electronic circuitry operable to electrically connect and disconnect may be suitably employed.

When open, contact 90 contributes to the systems overall parasitic capacitance Cp to an external circuit that can be minimized by the use of a large gap contact arrangement form of relay or the like.

As shown, line 120 shows the demarcation between the system HHM 130 and the external circuit AUTO 140 for illustrative purposes. In this case AUTO 140 depicts a typical automotive vehicle measurement scenario.

Illustrated is automotive battery 150 whose negative terminal 160 is connected to vehicle chassis ground 180 via 170. As shown ground probe 190 is attached or clamped onto ohmic contact with the negative chassis.

Positive terminal 200 can be seen to supply ECU 220 via 210 and shown is an example electrical conductor 230 from ECU 220. The present invention will now be described in its operation to measure the electrical potential or voltage on 230.

Shown in exaggerated view is conductor 230, with its electrical conductor 240 coaxial to its insulating sheath 250. uC 60 has an analog output 260 that drives buffer amplifier 270, whose output 280 is connected to excitation electrode 290. Electrode 290 is in physical contact with the outer insulation surface 250 with an intervening dielectric shroud 300. It can be seen that there is formed a capacitance Cl 310 between electrode 290 and conductor 240.

Also shown is sensing electrode 320 which is in physical contact with the outer insulation surface 250 with an intervening dielectric shroud 330. It can be seen that there is formed a capacitance C2 340 between electrode 320 and conductor 240.

It should be obvious that capacitance Cl 310 and capacitance C2 340 are unknowns, dependant on the electrode sizes and dielectric constants of their shrouds, their distance to the surface of the conductor, and the conductors diameter and its insulation dielectric constant.

And it can be seen that there is a small mutual interelectrode capacitance Cie 380. This capacitance Cie 380 is known a priori (i.e. can be determined by calculation or measurement during manufacturing).

Both electrodes assemblies are kept in contact with conductor insulation 250 via suitable spring loaded clamping means or the like.

Now as illustrated sensing electrode 320 is connected to the input of amplifier 360 via line 350. Amplifier 360 is of the very high input type such as an electrometer amplifier. Its output is fed via 370 to an analog input on uC 60.

Now as shown, the circuit comprised of electrode 320 and the positive input of amplifier 360 are virtually open circuit and will suffer from inherent drifting caused by unavoidable bias and offset currents assumed to exist in the amplifier 360 is. In order to bias electrode 320 and the positive input of amplifier 360 around a reference midpoint, line 390 is shown connected to antiparallel arranged ultra-high vacuum photoemissive (UHVPE) valves 400 and 480. These UHVPE valves 400 and 480 are also shown connected via line 430 to a midpoint reference created by resistors 440 and 450.

UHVPE valve 400 has its photocathode controllably illuminated via LED 420 which is controlled via uC 60 via analog output 410. LED 420 is of suitable wavelength to match the work function of the photocathode. As shown, UHVPE valve 400 can remove electrons from the electrode 320 circuit. Likewise shown is UHVPE valve 480 which has its photocathode controllably illuminated via LED 470 which is controlled via uC 60 via analog output 460. LED 470 is again of suitable wavelength to match the work function of the photocathode. And as shown, UHVPE valve 480 can inject electrons into the electrode 320 circuit.

Operation of the total system will now be explained in greater detail.

In a first or capacitance calibration instance, contact 90 is left open, even with lead 110 being connected to vehicle ground 180. The system imposes a reference voltage waveform onto conductor 240 via excitation electrode 290. Irrespective of any electrical potential on conductor 240 with reference to its own ground 180, because the system is floating, the imposed voltage will be primarily due to coupling through capacitance Cl 310. The imposed voltage on conductor 240 is then coupled to sensing electrode 320 primarily through capacitance C2 340.

As it is presumed that physical aspects of the capacitive structures remain static as well as dielectrics constant, and since the amplitude of the excitation reference voltage waveform is known, it is easy to calculate C1 =C2 after removing the Cie capacitance 380 and Cp capacitance 100, both of which are known. One can then calculate a correction or gain or scale factor therefrom.

Thereafter, in a second or potential measurement instance, contact 90 is closed thereby connecting system ground to the vehicle ground 180. Again, the system imposes a reference voltage waveform onto conductor 240 via excitation electrode 290. And in this instance any voltage on conductor 240 with reference to its own ground 180 will offset the imposed reference voltage waveform sensed by electrode 320. By applying the correction factor to the sensed voltage, the actual voltage on conductor 240 can be deduced.

Also during the calibration interval, as everything is floating and symmetric, the UHVPE valves can be utilized to maintain midpoint bias and to remove the effects of drift on sensing electrode 320.

The calibration sequence can be repeated quickly and at occasional intervals to ensure no changes in the capacitive structure have occurred. Also, the calibration sequence can be triggered by uC 60 when it detects that electrode 320 has drifted too far from its midpoint reference.

Now further shown in FIG. 11 are connector jack 520 tied to system ground and connector jack 510 which is fed an analog output voltage from uC 60 via line 500. There is a connection interface MMOUT 490 to allow the connection of a normal voltmeter to display the measured voltage if the HHM system is not a standalone unit (display components have not been shown for clarity.) Further shown is transceiver 540 which may be of any suitable type, which has its RF output 550 connected to antenna 560. The transceiver 540 has a bidirectional data link 530 in order that it can communicate the systems measurements to a remote handheld device such as a smartphone, tablet or the like.

Thus, there is provided a non-invasive voltmeter that includes the previously described and illustrated electrometer, and an excitation source operable to apply an excitation signal to an excitation terminal, the excitation signal being a step excitation, a controller of the electrometer being operable to determine, in response to the step excitation, a step change in the output of the amplifier separate from the output drift.

Method of Correcting Output Drift of an Amplifier

The electrometer 10 of Fig. 1 , the electrometers 10 of Figs. 2 to 5, the system shown in Fig. 6, and the non-invasive voltmeter 1100 of Fig. 11 includes a controller, such as microcontroller 560 of Fig. 1 , the uC or uP 407 of Fig. 7, and/or the uC 60 of Fig. 11. As is well known in the art, a controller such as a microcontroller or microprocessor typically includes a processing circuit, such as a processor which may be a Central Processing Unit (CPU), and a memory circuit typically operable to store digital representations of data or other information, including measurement results and/or control information, and to store digital representations of program data or other information, including program code for directing operations of the processor.

Referring to Fig. 12, the memory circuit in accordance with embodiments of the invention and variations thereof contains blocks of code comprising computer executable instructions for directing the processor to perform the steps of a method shown generally at 1200. Additionally or alternatively, such blocks of code may form part of a computer program product comprising computer executable instructions embodied in a signal bearing medium, which may be a recordable computer readable medium or a signal transmission type medium, for example.

Referring to Figs. 1 and 12, when electrical power is being supplied to the processor and the memory circuit, the processor is directed to begin executing the instructions of block 1202. Block 1202 then directs the processor to cause anti-drift circuitry of the electrometer 10 to apply a non-floating input to a floating input terminal of an amplifier to set its amplifier output to a predetermined value. In the first embodiment of Fig. 1 , executing block 1202 may involve causing both the light-emitting diodes 720 and 740 to be fully actuated such that maximum current flows through the phototubes 700 and 750 and the phototubes 700 and 750 present minimal series resistance. Under such conditions, the non-floating voltage between the resistors 770 and 780 is applied via line 760 at minimal series resistance via the phototubes 700 and 750 along line 690 to the floating high-impedance positive input 660 of the op-amp 670. Given that the nonfloating input between the resistors 770 and 780 has a predetermined value and the gain of the op-amp 670 is known, the result is that a predetermined value appears at the output 680 of the op-amp 670. Executing block 1202 typically involves waiting a specifiable amount of time, such as waiting until the amplifier output reaches the predetermined value, before ending execution of block 1202.

After block 1202 has been executed, block 1204 then directs the processor to remove the non-floating input from the floating input terminal. Block 1204 typically involves ceasing to actuate the light-emitting diodes 720 and 740 such that the phototubes 700 and 750 appear as an extremely high-impedance open circuit to the positive input 660 of the op-amp 670.

Blocks 1202 and 1204 may be considered to provide the op-amp 670 a hard reset by forcing its output to the predetermined value, which is typically selected as being midway between the supply rail voltages for the op-amp 670. In this manner, a hard reset of the op-amp 670 maximizes the extent of output drift that can occur without the saturating the op-amp 670 at one of its supply rail voltages.

After block 1204 has been executed, block 1206 then directs the processor to determine a deviation of the amplifier output from the predetermined value. Executing block 1206 typically involves the microcontroller 560 digitizing the output voltage of the op-amp 670, comparing the output voltage value to the predetermined value, and determining a deviation, or difference, between the output voltage value and the predetermined value. Typically, the deviation immediately after executing blocks 1202 and 1204 is zero or negligibly small, and then increases either positively or negatively over time at a typically predictable rate. When the deviation becomes sufficiently notable, block 1206 ends and block 1208 is executed.

Block 1208 directs the processor to apply an electrical current from a high- impedance current source to the floating input terminal so a to minimize the deviation. In the embodiment of Fig. 1 , the microcontroller 560 determines the magnitude and polarity (positive or negative) of the deviation and produces a actuating voltage signal at line 710 or line 730, selected according to the polarity of the deviation. The magnitude of the actuating voltage signal at the line 710 or 730 is selected by the microcontroller 560 in response to the magnitude of the deviation, according to a predetermined scale factor, lookup table, or the like. The actuating voltage signal actuates the light-emitting diode 720 or the lightemitting diode 740 so as to cause the electrical current to flow from the phototube 700 or the phototube 750 such that the electrical current applied at the floating input terminal 660 of the op-amp 670 counteracts the generation of the output drift of the op-amp 670, thereby minimizing the deviation as determined by the microcontroller 560. Ideally, the actuation of the light-emitting diode 720 or 740 is selected to eliminate the deviation and return the output voltage of the opamp 670 to the predetermined value (of block 1206).

The process of blocks 1202, 1204, 1206, and 1208 can continue indefinitely as long as the electrometer 10 is being powered.

After block 1208 has been executed, block 1210 directs the processor to determine whether the electrometer 10 is being powered down.

If the electrometer 10 is not being powered down, then block 1212 directs the processor to determine whether the deviation is greater than a threshold. The value of the threshold may be predetermined such as during manufacturing and may be hard-coded into firmware of the electrometer 10, for example.

If the deviation is greater than the threshold, then the method returns to block 1202 for a hard reset of the op-amp 670.

If the deviation is less than the threshold, then the method returns to block 1206 to update the deviation and then by block 1208 apply a counter-deviation electrical current to the floating input terminal 660.

If by executing block 1210 it is determined that the electrometer 10 is being powered down, then the method 1200 ends.

While the method 1200 of Fig. 12 has been described herein above with reference to the electrometer 10 of Fig. 1 , the method 1200 is applicable to correcting the output drift of any suitable amplifier, including various operational amplifiers and particularly operational amplifiers configured to have a floating input.

Thus, there is provided a computerized method of correcting output drift of an amplifier having a floating input terminal, the method comprising: (a) applying a predetermined non-floating input to the floating input terminal of the amplifier; (b) disconnecting the predetermined non-floating input from the floating input terminal; (c) determining by a computerized controller a deviation of the output voltage of the amplifier from a predetermined output value associated with the predetermined non-floating input; and (d) in response to the deviation, applying an electrical current from a high-impedance current source to the floating input terminal so as to minimize the deviation.

Method of Non-lnvasively Measuring Voltage on a Conductor

Referring to the non-invasive voltmeter 1100 of Fig. 11 and the method 1300 of non-invasively measuring voltage on a conductor shown in Figs. 13A, 13B, and 13C, the method 1300 begins execution at block 1302.

When electrical power is being supplied to the non-invasive voltmeter 1100 of Fig. 11 , block 1302 directs the processor (e.g. CPU) of the microcontroller 60 to initialize the non-invasive voltmeter 1100 such that a relay is open. In the embodiment of Fig. 11 , executing block 1302 may involve the microcontroller 60 opening the relay 80 by applying an appropriate digital output value at pin 70 of the microcontroller 60 such that the negative (ground) terminal 20 of the battery 10 is disconnected from the external (chassis) electrical ground 190.

After block 1302 has been executed, block 1304 directs the processor to perform the same or analogous steps described herein above in respect of Fig. 12. For example, with reference to the embodiment of Fig. 11 , executing block 1304 of Fig. 13A may involve causing both the light-emitting diodes 420 and 470 to be fully actuated such that maximum current flows through the phototubes 400 and 480 and the phototubes 400 and 480 present minimal series resistance. Under such conditions, the non-floating voltage between the resistors 440 and 450 is applied via line 430 at minimal series resistance via the phototubes 400 and 480 along line 390 to the floating high-impedance positive input 350 of the amplifier 360. Given that the non-floating input between the resistors 440 and 450 has a predetermined value and the gain of the amplifier 360 is known, the result is that a predetermined value appears at the output 370 of the amplifier 360. Executing block 1304 typically involves waiting a specifiable amount of time, such as waiting until the amplifier output reaches the predetermined value, before ending execution of block 1304.

After block 1304 has been executed, block 1306 then directs the processor to perform the same or analogous steps described herein above in respect of Fig. 12. For example, with reference to the embodiment of Fig. 11 , executing block 1306 of Fig. 13A typically involves ceasing to actuate the lightemitting diodes 420 and 470 such that the phototubes 400 and 480 appear as an extremely high-impedance open circuit to the positive input 350 of the amplifier 360.

Blocks 1304 and 1306 may be considered to provide the amplifier 360 a hard reset by forcing its output to the predetermined value, which is typically selected as being midway between the supply rail voltages for the amplifier 360. In this manner, a hard reset of the amplifier 360 maximizes the extent of output drift that can occur without the saturating the amplifier 360 at one of its supply rail voltages.

After block 1306 has been executed, block 1308 then directs the processor to perform the same or analogous steps described herein above in respect of Fig. 12. For example, with reference to the embodiment of Fig. 11 , executing block 1308 of Fig. 13A typically involves the microcontroller 60 digitizing the output voltage 370 of the amplifier 360, comparing the output voltage value to the predetermined value, and determining a deviation, or difference, between the output voltage value and the predetermined value. Typically, the deviation immediately after executing blocks 1304 and 1306 is zero or negligibly small, and then increases either positively or negatively over time at a typically predictable rate. When the deviation becomes sufficiently notable, block 1308 ends and block 1309 is executed. Block 1309 directs the processor to perform the same or analogous steps described herein above in respect of Fig. 12. For example, with reference to the embodiment of Fig. 11 , in accordance with block 1308 of Fig. 13A, the microcontroller 60 determines the magnitude and polarity (positive or negative) of the deviation and produces an actuating voltage signal at line 410 or line 460, selected according to the polarity of the deviation. The magnitude of the actuating voltage signal at the line 410 or 460 is selected by the microcontroller 60 in response to the magnitude of the deviation, according to a predetermined scale factor, lookup table, or the like. The actuating voltage signal actuates the light-emitting diode 420 or the light-emitting diode 470 so as to cause the electrical current to flow from the phototube 400 or the phototube 480 such that the electrical current applied at the floating input terminal 350 of the amplifier 360 counteracts the generation of the output drift of the amplifier 360, thereby minimizing the deviation as determined by the microcontroller 60. Ideally, the actuation of the light-emitting diode 420 or 470 is selected to eliminate the deviation and return the output voltage of the amplifier 360 to the predetermined value (of block 1308).

The execution of blocks 1304, 1306, 1308, and 1309 can loop indefinitely within the context of the method 1300.

However, after block 1309 has been executed, block 1310 directs the processor to apply an excitation voltage to an output terminal of the non-invasive voltmeter. Executing block 1310 may involve the microcontroller 60 producing an analog output 260 that drives buffer amplifier 270, whose output 280 is connected to excitation electrode 290. The analog output 260 may be a squarewave, for example, of a selectable duty cycle and/or amplitude.

After block 1310 has been executed, block 1312 directs the processor to determine a measured voltage of the non-invasive voltmeter. Executing block 1312 may involve the microcontroller 60 digitizing the output voltage 370 of the amplifier 360, for example. After block 1312 has been executed, block 1314 directs the processor to determine whether the relay is closed. Block 1314 may involve the microcontroller 60 determining whether the relay 80 is closed.

If the relay 80 is not closed, then block 1316 directs the processor to calculate a loss factor in response to the excitation voltage and the measured voltage. Block 1316 may involve calculating the loss factor to be equal to the peak-to-peak value of the squarewave excitation voltage, divided by the measured voltage determined by block 1312. The loss factor provides an indication of the attenuation of the excitation after passing through the conductor 240 when it is capacitively coupled to the excitation electrode 290 and to the sensing electrode 320. Such attenuation is assumed to occur as a result of noninfinite input impedance of the amplifier 360.

After block 1316 has been executed, block 1318 directs the processor to remove the excitation signal. Executing block 1318 may involve removing the analog output 260, such as by allowing the duty cycle of the squarewave to transition to an off-portion of the duty cycle. Typically, the off-portion of the duty cycle is at zero volts.

Block 1320 then directs the processor to close the relay, such as the controller 60 causing the relay 80 to close so that the negative terminal of the battery 10 becomes connected to the (chassis) electrical ground 190.

After block 1320 of Fig. 13B has been executed, the method returns to block 1312 of Fig. 13A.

Block 1312 is described herein above and results in a measured voltage being determined.

Referring to block 1314 of Fig. 13A, if the relay is closed, then block 1322 of Fig. 13C directs the processor to calculate a scaled measured voltage in response to the measured voltage and the loss factor. Executing block 1322 may involve calculating the scaled measured voltage to be equal to the measured voltage (previously determined by block 1312) multipled by the square root of the loss factor. By using the loss factor to scale the measured voltage, the scaled measured voltage represents the voltage on the conductor 240. In the described embodiment, this accurate determination of the voltage on the conductor 240 is made possible by advantageously timing the measurements with a step change in conditions caused by the closing of the relay 80 so as to suddenly "expose" the voltage of the conductor 240 to the amplifier 360 via the capacitive coupling of the sensing electrode 320.

In a variation of embodiments, block 1318 described above need not be executed such that the excitation signal is not removed. In such embodiments, the measured voltage of block 1312 is taken to be the change in output voltage 370 of the amplifier 360. That is, the change in the output voltage 370 from when the relay was open to the output voltage 370 when the relay becomes closed is calculated to produce the measured voltage of block 1312, which in turn is used by block 1322 to calculate the scaled measured voltage.

After block 1322 has been executed, block 1324 directs the processor to open the relay, such as relay 80 of Fig. 11 . Opening the relay 80 resets the steps of the method so that another measurement can be taken. In this manner, multiple measurements can be taken in sequence at a sufficiently high rate for averaging, removing spurious results, extracting from the measurements the effect of a relatively slow-moving output drift with correction, etc., thereby advantageously improving accuracy.

After block 1324 has been executed, block 1326 directs the processor to determine whether the non-invasive voltmeter 1100 is being powered down.

If the voltmeter 1100 is not being powered down, then block 1328 directs the processor to determine whether the deviation is greater than a threshold. The value of the threshold may be predetermined such as during manufacturing and may be hard-coded into firmware of the voltmeter 1110, for example.

If the deviation is greater than the threshold, then the method returns to block 1304 of Fig. 13A for a hard reset of the amplifier 360. If the deviation is less than the threshold, then the method returns to block 1308 of Fig. 13A to update the deviation and then by block 1309 apply a counterdeviation electrical current to the floating input terminal 350.

If by executing block 1326 of Fig. 13C it is determined that the voltmeter 1110 is being powered down, then the method 1300 ends.

Thus, there is provided a method of non-invasively measuring voltage on a conductor, the method comprising: (a) applying a predetermined non-floating input to the floating input terminal of the amplifier; (b) disconnecting the predetermined non-floating input from the floating input terminal; (c) determining by a computerized controller a deviation of the output voltage of the amplifier from a predetermined output value associated with the predetermined nonfloating input; (d) in response to the deviation, applying an electrical current from a high-impedance current source to the floating input terminal so as to minimize the deviation; (e) applying a step excitation at an output terminal of a non- invasive voltmeter comprising the electrometer and determining a measured voltage when a relay of the non-invasive voltmeter is open; (f) calculating a loss factor in response to a magnitude of the step excitation and the measured voltage; (g) removing the step excitation and closing the relay; (h) determining a second measured voltage when the relay is closed; and (i) calculating a scaled measured voltage in response to the second measured voltage and the loss factor.

While embodiments of the invention have been described and illustrated, such embodiments should be considered illustrative of the invention only. The invention may include variants not described or illustrated herein in detail. Thus, the embodiments described and illustrated herein should not be considered to limit the invention as construed in accordance with the accompanying claims.