TECHNICAL FIELD OF THE INVENTION

This invention relates generally to the field of switching power conversion such as DC-DC and DC-AC power conversion, and more specifically to high definition switching audio power amplification.

BACKGROUND

Switching Class D audio amplifiers have found increasing use in the industry in recent years, due to the improvements in output stage switching devices and equally modulation and feedback control methods. The classical switching power amplifier system consists of the pulse modulator, converting an analog or digital source into a pulse-modulated signal, following amplified by a switching power stage. A passive demodulation filter reproduces the power modulated power signal.

Most switching class D amplifiers are based on variants of Pulse Width

Modulation (PWM).

The switching power stage consists of switching devices or elements, such as, but not limited to, MOSFETs (metal-oxide-semiconductor field-effect transistors).

When the Class D audio amplifier is outputting high output current the output devices in the switching power stage is often the hotspot. For protection against overheating of the switching devices it is desired to have a precise measurement of the temperature in the switching devices.

In an integrated power stage, it is easy to include a temperature sensor on the die, to get a good measurement, which also tracks fast temperature raises well. This is more difficult when using discrete output power devices. In case of a fast temperature increase of the switching device, the

preciseness of the measured temperature of the switching device depends among other things on the thermal resistance from the switching device to the sensor that measures the temperature, and of the thermal capacitance of the area where the temperature sensor is mounted. On the above background it is desirable to have access to systems, devices and methods that provide fast temperature protection of a switching device or element, such as MOSFETs. SUMMARY OF THE INVENTION

It is therefore an object of the present invention to provide a

temperature measuring system, a gate driver and a method for making temperature measurements in switching power conversion systems and devices which alleviate all or at least some of the above-discussed drawbacks of presently known system.

The above and further objects and advantages are attained by the system, device and method according to the appended claims. The term exemplary is in the present context to be interpreted as serving as an example, instance or illustration.

According to a first aspect of the present invention there is provided a precise and fast temperature measuring system for a switching power conversion system, the power conversion system comprising a switching device, wherein the temperature measuring system comprises:

a temperature sensor configured to provide a temperature signal (corresponding to a sensor temperature, T);

means for differentiating the temperature signal from the temperature sensor, such that a temperature of the switching device is determined as a sum of the sensor temperature signal, and a constant, p, times the time derivative of the sensor temperature signal.

Thus, by means of such a system it is suggested to generate a time derivative of the sensor temperature signal and to add this value multiplied by a constant, p, to the sensor temperature signal in order to achieve a faster and more precise temperature estimation/determination of a switching device in a switching power conversion system. Moreover, in audio application where a converted signal is an audio signal and the load is reasonably symmetric, the heating of both switching devices will be very similar and sensing temperature on one will represent the other, thus in audio applications the present invention allows for improved temperature measurements and consequently improved protection for overheating of both of the switching devices.

In other words, the aforementioned temperature measuring system has differentiating circuitry configured to differentiate the temperature signal in order to provide a time derivative, of a sensor temperature signal, and whereby the temperature measuring system is configured to determine a temperature of the switching element (or of internal structures of the switching element), as a sum of the sensor temperature signal, and the time derivative of the sensor temperature, times a constant, p. More specifically, a

representation of a temperature of the switching device is determined as the above described sum. The representation is a voltage signal which is indicative of the switching device temperature.

According to an exemplary embodiment the constant, p, is a product of is a product of:

a resistance representing a parallel connection of a thermal

resistances from a die of the switching device to the temperature sensor and a thermal loss resistance from the sensor to the environment, and

a thermal capacitance of the sensor and an area on which it is mounted.

In other words, the constant, p, is a product of the parallel connection of the thermal resistance from the die of the switching device to the

temperature sensor together with the thermal loss resistance from the sensor to the environment, and the thermal capacitance of the sensor and the area on which it is mounted.

In more detail, if the temperature of the switching device is represented by:

VCOMP = V+p ^* dV/dt

Where p is a constant depending on the mechanical design, V is the sensor temperature signal and dV/dt is the sensor temperature signal gradient (i.e. time derivative of the sensor temperature signal, V). Then the constant, p, is defined as the product of a resistance, R, and a thermal capacitance, C, i.e. p=RC. Here, R is to be understood as R=R2 ^* R3/(R2+R3), where R2 is a thermal resistance from a die of the switching device to the temperature sensor and R3 is a thermal loss resistance from the temperature sensor to the environment/ambient. The thermal capacitance C is defined as a thermal capacitance of the sensor and an area on which it is mounted. This will however further described in the detailed description in reference to the appended drawings.

The power dissipated in the power device is also dissipated in the power device die, making it hot. Ideally this temperature should be sensed, but the sensor is normally placed outside the device and will be (slightly) colder than the die. The sensor has the temperature T, which in an electrical equivalent schematic of the thermal system has been provided the symbol V. The sensor system voltage, described as NTC_in in the illustrations is an equivalent to T and V, but will have a gain and offset compared to V, and will have a linearity error.

In more detail, VCOMP is a representation of the switching device temperature and is furthermore a compensated version of an electrical temperature equivalent of the sensor temperature signal, V; and it is compensated by the addition of a fraction of the time derivative of V, in order to compensate for the thermal mass around the sensor, and thereby a much better representation of the dice temperature (and consequently the switching device temperature) can be provided in applications having dynamic power dissipation such as e.g. in amplifiers and other dynamic systems.

In the present context, the term die (or dice) is a small block of semiconducting material on which a given functional circuit is fabricated, e.g. a switching device.

According to an embodiment of the first aspect of the present invention the switching device is a MOSFET.

According to an embodiment of the first aspect of the present invention the temperature is measured at the drain terminal of a high-side MOSFET in the power stage by means of a temperature sensor. The high-side MOSFET is in the present context to be interpreted as the MOSFET having a terminal connected to positive voltage supply. According to another embodiment of the first aspect of the present invention the temperature sensor is electrically connected to the drain of the high side MOSFET.

According to yet another embodiment of the first aspect of the present invention the temperature sensor is a NTC resistor.

According to yet another embodiment of the first aspect of the present invention the system comprises means configured for level shifting said temperature signal to an input signal level.

According to a first embodiment of the first aspect of the present invention the means configured for level shifting are integrated into an integrated circuit.

According to yet another embodiment of the first aspect of the present invention the means for differentiating the sensor temperature signal or a part of the means for differentiating the sensor temperature signal are integrated into an integrated circuit.

According to yet another embodiment of the first aspect of the present invention the means for differentiating are configured to provide a limited differentiator function that can only increase the measured temperature.

According to yet another embodiment of the first aspect of the present invention the increase of the measured temperature has a limited level.

According to a second aspect of the present invention there is provided a gate driver for a Class D amplifier, where the gate driver is provided with a precise fast temperature measuring system according to the first aspect of the invention and any embodiments hereof.

According to the second aspect of the present invention the gate driver is provided with a precise fast temperature measuring system according to the first aspect of the present invention for measuring temperature at a sensor that is electrically connected to the one terminal of an output switching element (such as the drain terminal of a MOSFET).

In one exemplary embodiment, the temperature is measured at the high-side MOSFET drain in order to minimize the thermal resistance from the power device die to the sensor, where the power device driver includes a circuit that increases the temperature measurement or correspondingly lowers the action thresholds, with which the measurement is compared relative to the temperature rising rate dV/dt, where the power device driver is provided with internal level shifters to transfer the measured signal to the potential where the temperature information is needed.

With this aspect of the invention, similar advantages and preferred features are present as in the previously discussed first aspect of the invention.

According to a third aspect of the present invention there is provided a method for making or performing temperature measurements in switching power conversion systems and devices, such as e.g. DC-DC, AC-DC, DC-AC power conversion systems and switching audio power amplification, such as e.g. Class D amplifiers. More specifically in systems or devices that comprise switching elements, such as e.g. MOSFETs. The method for making temperature measurements comprises:

providing a temperature sensor in thermal connection with at least one of said switching devices;

determining a sensor temperature;

determining a time derivative of the sensor temperature;

estimating a temperature of the switching device or of internal structures of the switching device as a linear combination of the sensor temperature and the time derivative of the sensor temperature. Hereby a method for fast and precise temperature protection of switching devices of a switching power conversion system is provided.

According to an embodiment of the third aspect of the present invention the linear combination is given by:

VCOMP = V + p(dV/dt)

where VCOMP is the sensor temperature (equivalent signal) in the stationary state, i.e. in the state where dV/dt is equal to zero. In this state, VCOMP is equal to Die_temp ^* k. V is the sensor temperature and p is a constant that characterizes the specific layout of the switching element and its

surroundings.

According to another embodiment of the third aspect of the present invention the internal structures is the die of the switching element. According to yet another embodiment of the third aspect of the present invention the constant p is a product of:

a resistance representing a parallel connection of a thermal resistance from a die of the switching device to the temperature sensor and a thermal loss resistance from the temperature sensor to the environment, and

a thermal capacitance of the sensor and an area on which it is mounted.

According to yet another embodiment of the third aspect of the present invention the temperature is measured at the drain terminal of the high-side MOSFET in the power stage of the power conversion system. In other words, in one exemplary embodiment, the step of providing a temperature sensor further comprises providing a temperature sensor in thermal connection with a drain terminal of a high-side MOSFET of the power conversion system. Thus, the step of determining a sensor temperature then comprises measuring a temperature at the drain terminal of the high-side MOSFET.

According to yet another embodiment of the third aspect of the present invention the temperature sensor is electrically connected to the drain of the high-side MOSFET. In more detail, the step of providing a temperature sensor, further includes providing a temperature sensor in thermal and electrical connection with the drain terminal of the high-side MOSFET.

According to yet another embodiment of the third aspect of the present invention the temperature sensor is a NTC resistor.

Also, with this aspect of the invention, similar advantages and preferred features are present as in the previously discussed first aspect and second aspect of the invention, and vice versa.

The application of the precise fast temperature measuring system, the gate driver, and/or the method according to the three aspects of the present invention result in more precise temperature monitoring at rapid temperature rises resulting in better protection and/or price reduction on power devices and cooling system in dynamic systems, such as in switching power conversion systems, such as DC-DC and DC-AC power conversion, and more specifically in high definition switching audio power amplification systems. These and other features and advantages of the present invention will in the following be further clarified with reference to the embodiments described hereinafter BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be better understood by reading the following detailed description of an embodiment of the invention in conjunction with the drawings, where:

Figs. 1 (a) and 1 (b) show a simplified thermal equivalent circuits that may be used to determine relation between temperature and the power dissipated in an output switching device;

Fig. 2 shows a schematic block diagram of a half bridge gate driver according to an embodiment of the invention used for driving Class D power amplifiers;

Fig. 3 shows an embodiment of a NTC branch of a voltage dividing circuit that is used in an embodiment of the invention; and

Fig. 4 shows a circuit diagram of an embodiment of a fast temperature measuring system according to the invention.

DETAILED DESCRIPTION

In the following detailed description, preferred embodiments of the present invention will be described. However, it is to be understood that features of the different embodiments are exchangeable between the embodiments and may be combined in different ways, unless anything else is specifically indicated. Even though in the following description, numerous specific details are set forth to provide a more thorough understanding of the present invention, it will be apparent to one skilled in the art that the present invention may be practiced without these specific details. In other instances, well known constructions or functions are not described in detail, so as not to obscure the present invention

With reference to figures 1 (a) and 1 (b) the basic theory necessary for understanding the principles of the invention is given. Referring to figure 1 (a) there is shown a simplified thermal equivalent schematic, where the power dissipated in an output device is represented by an equivalent current source 1. The outer resistor 2 having a resistance R1 represents thermal resistances from the power device die to surrounding environment. The resistor 3 has a resistance R2 and represents a thermal resistance from the die to the temperature sensor, while the capacitor 4, having a capacitance C, represents the thermal capacitance of the thermal sensor and the area it is mounted to. The last resistor 5, having a resistance R3 is a loss resistance from the thermal sensor area to surrounding environment, indicating that the thermal sensor will never fully reach the temperature of the power device die.

To make the measurement good (and accurate), the resistance R3 should be much higher than the resistance R2. When the power device is a MOSFET, the best thermal connection to the die will normally be at the drain terminal of the MOSFET. As standard electrical conductors generally are much better thermal conductors than standard electrical insulators, the lowest value of R2 will be achieved when one terminal of the thermal sensor is electrically connected to the MOSFET drain. In a half bridge topology, one MOSFET drain is connected to the switching node, electrically moving up and down, the other drain is connected to the high positive supply. In the case the converted signal is an audio signal, and the load is reasonably symmetric, the heating of the high and low MOSFET will be very similar and sensing temperature on one will represent the other. Due to noise in the switching node, the more convenient place to measure of the two is at the high positive supply. Measuring temperature precisely at the high positive supply would require some kind of local supply and level shifting circuits to level shift it to signal input level to be able to react on the temperature information, all adding to component count and cost, when made in discrete electronics.

Also, to make the measurement good, the sensor should be on a small thermal mass to make the capacitance C (and the capacitor 4) small, and the sensor should not be cooled too much, to make R3 large.

A further problem relates to the thermal capacitance C. When the thermal sensor is PCB mounted, possibly SMT mounted, it will be situated on a thermal mass. If the thermal mass is very cold, and power dissipation suddenly becomes very big, the sensor can for a while be far colder than the power device die.

Further, if we choose k= R3/(R2+R3) and R=R2 ^* R3/(R2+R3), the equivalent schematic shown in figure 1 (a) is simplified to the schematic shown in figure 1 (b). In this schematic, V _C oMP=Die _TEMP ^* k is in the electrical equivalent schematic the temperature the sensor will finally get in a stationary situation, a small amount less than the die temperature. This is just a matter of the chosen gain, and can be compensated for. It is however better that k is as close as possible to unity, i.e. if R2«R3.

If the die temperature rises very quickly, the sensor temperature will be lacking/lagging behind the die temperature development, being much colder than the die. To protect the switching devices in the fast rising case, there is needed a rather low temperature limit. In more normal use, when the temperature rises slowly, the sensor will be at almost the same temperature as the power device, but the temperature limit will still be low, and the power device is not used to its full temperature potential, reducing performance from what the power stage would be able to do in the slow rising case, if the sensing was more correct.

The formula for the equivalent voltages (where voltages represent temperatures) is as follows:

VcoMP=V+R ^* C ^* dV/dt

As R and C are constant for a given mechanical design, the above expression can be re-written as:

VCOMP = V+p ^* dV/dt, where p is a constant depending on the

mechanical design, V is the sensor temperature, VCOMP is the temperature that would be obtained if the above described problems relating to the capacitance were not present and dV/dt is a representation of the sensor temperature gradient.

In other words, when the sensor temperature and the temperature gradient for a system is found, and where the constants p and k have been found, it is possible to determine/calculate the temperature of the power device die and thereby make a better protection and/or a cheaper design. Referring to figure 2 there is shown a schematic block diagram of a half bridge gate driver 8 used for driving Class D power amplifiers. The output stage of the power amplifier comprises the two MOSFETs 13 and 14, where the source terminal of MOSFET 14 is connected to one polarity PVSS 17 of a dual polarity voltage supply and the drain terminal of the MOSFET 14 is connected via the output switching node 15 to the source terminal of the other MOSFET 13, the drain terminal of which is connected to the other polarity PVDD 16 of the dual polarity voltage supply. The gate driver 8 provides an output signal 24 from the controls and indicator block 23 and the output gate driver signals from the high side driver block 32 and the low side driver block 29, respectively, for the respective output power MOSFET 13 and 14. These driver output signals are based in the input PWM signal 25 provided to the input block 26 as schematically illustrated in the block diagram in figure 2.

Integrated into the half bridge gate driver 8 there is provided an embodiment 10 of the temperature sensing system according to the inventive concept, which embodiment comprises the temperature sensor 18 in electrical and thermal connection with the drain terminal of the high-side MOSFET 13 and a temperature sensing circuit comprising the temperature sense block 19, a differentiator 20 and a level shifter 21.

The half bridge gate driver 8 comprises a plurality of level shifters 22,

27, 28, 30 and 31 configured to ensure the correct level at different internal interfaces in the gate driver 8. The output signal provided by the sensing device 10 is provided to the controls and indicator block 23.

A self-supplying IC-well below the high power supply rail for the power stage contains the temperature sensor device 10, to allow the sensor to be electrically connected to the drain of the high side MOSFET 13 as indicated by the heavy line 33 in figure 2. The signal from the temperature sensing block is level shifted to the input well. For high precision at the important decision thresholds, given the bad absolute matching of internal chip components from chip to chip and the high electrical noise level in a power stage, the comparison circuit is made as a local circuit. The output voltage or voltages provided by the external resistor to NTC (negative temperature coefficient resistor) voltage divider (that is schematically illustrated in Fig. 3 3) is compared to an internal resistor voltage divider.

The NTC branch of the external voltage dividing circuit shown in Fig. 3 comprises the temperature sensor as such, i.e. the NTC resistor 34

connected in parallel with a resistor R _P 35. In series with the parallel connection of the NTC and Rp 34, 35 is the series connection of the resistors Rs and RD 36 and 37 and to the node formed between these resistors Rs and RD there is connected to capacitor CDIF 38. The two resistors 35, 36

connected to the NTC 34, may be referred to as helper resistors used to modify and linearize a resistor curve of the NTC 34. The parallel resistor R _P 35 is dominant when the NTC has high impedance, and Rs 36 is dominant when the NTC has low impedance.

The top of the external voltage divider is at the potential PVDD and the bottom of the external voltage divider is at the potential NTC_gnd. The external voltage divider provides two intermediate voltages i.e. (i) the voltage NTC_diff at the terminal of the capacitor CDIF opposite the node between RS and RD and (ii) the voltage NTC_in at the node between RS and RD, where NTC_in represents the sensor temperature.

The output voltage NTC_in(t) of a NTC voltage divider is not linear versus temperature. This means that adding a fraction of the differentiated sensor signal dV/dt to the sensed voltage an approximation to adding a fraction of dT/dt to the temperature. A temperature sensing system should have a temperature readout and an over-temperature protection, with the highest focus on precision in the high temperature range, when approaching temperature shutdown. To achieve this goal with a reasonable complexity and precision, the NTC (top) and resistor (bottom) voltage divider voltage is differentiated, and the result is subtracted from an internal voltage divider top voltage. The designed differentiator (54, 55, 56, 57, 58 in Fig. 4) is inverting and configured to output 84% to 100% of the reference voltage, PVDD, to the resistor ladder. This way the differentiator will not compensate for sensor dV/dt at falling temperatures since the OP-amp 54 will saturate and the output signal, Ref, will be same as the signal at the non-inverting input, PVDD. At a constant temperature, its output will be at the reference since there will be no signal at Diff_in. At rising temperatures, the high temperature thresholds will be reduced by up to 16% in voltage depending on the temperature gradient, corresponding to max 30K in temperature. The value of the differentiator resistor 55 is chosen high enough for the external differentiator capacitor 58 to be of acceptable size while not too big area-wise on the chip.

The resistor ladder voltages are compared to the sensor input voltage, NTC_in, by 7 comparators, the 7 outputs (giving 8 possible states) are converted to 3 bits, level shifted to the GND reference block as seen in the block schematic and used inter alia for protection, to provide user information and to provide temperature compensation of potential current limiting means that are used in the power stage.

Further, referring to Fig. 4 there is shown a circuit diagram of an example embodiment of a fast temperature protection and switching device according to the invention. In this embodiment the temperature-dependent branch of the voltage divider, i.e. the NTC resistor 59 with its parallel resistor 60 and the series resistors 61 and 62 are provided outside the integrated circuit (termed "inside IC" in Fig. 4) as well as the differentiator capacitor 58. The temperature-dependent branch of the voltage divider provides the voltages NTC_in and Diffjn to the circuit in the IC 37. The voltage Diff_in results in a current flowing into virtual GND of the operational amplifier 54.

In the IC 37 there is provided the inverting-differentiating circuit that, apart from capacitor 58, in this embodiment comprises an operational amplifier 54 (although other amplifying elements than an operational amplifier might alternatively be applied), a feedback resistor 55 (sometimes referred to as differentiating resistor), a filtering capacitor 56 connected in parallel with the feedback resistor 55 and a pair of diodes 57. The output from the differentiator is connected to the Ref node 65 in the resistive voltage divider 45-53. The circuit input, "Diffjn" is a trans-impedance input, converting the current from the differentiator capacitor 58 into a voltage over the feedback resistor 55, slightly filtered by the filtering capacitor 56.

A series of comparators 38-44 is also provided within the IC 37. One input terminal of each respective comparator (in the figure the non-inverting input terminal) is connected to the voltage NTC_in that is not differentiated, whereas the other input terminal (in the figure the inverting input terminal) of each respective comparator is connected to respective voltage divider nodes 66-72 of the resistive voltage divider 45-53.

The output signals from each respective comparator 38-44 are coupled to a 8 to 3 coder and 3 bit level shifter 63 that provides the output signals (3 bits) 64 to the GND block (ref. 9 in figure 2).

The embodiments of the present invention can be implemented with discrete components. Further, the fast temperature protection device according to embodiments of the invention can be partly or completely integrated in the driver device of the amplifier. The differentiator can be implemented as an analog circuit or as a software algorithm.

The device according to the invention can be used not only with the MOSFET switches as illustrated in this description, but also with different other types of output switches, such as GaN, IGBT and other power switching elements.

The temperature measure could be measured at the low side MOSFET drain but it is a more complex solution, since the pin is moving with the PWM signal.

The fast temperature protection device according to the invention can be used in single ended amplifiers and in connection with BTL dual supplies, BTL single supply, systems with constant switch frequency and systems with variable switch frequency such as self-oscillating systems. Apart from use in power amplifiers, such as audio power amplifiers, the fast temperature protection device according to the invention can also be used in motor drivers and power supplies.

It is understood that the embodiment of the invention that is described in detail above only constitute a non-limiting example of an implementation of the principles of the present invention and that the scope of the invention is defined by the claims.