FIELD This invention relates generally to a converter, particularly though not solely, to a converter for an inductive power transmitter.

BACKGROUND Electrical converters are found in many different types of electrical systems.

Generally speaking, a converter converts a supply of a first type to an output of a second type. Such conversion can include DC-DC, AC-AC and DC-AC electrical conversions. In some configurations a converter may have any number of DC and AC 'parts', for example a DC-DC converter might incorporate an AC-AC converter stage in the form of a transformer.

One example of the use of converters is in inductive power transfer (IPT) systems. IPT systems are a well-known area of established technology (for example, wireless charging of electric toothbrushes) and developing technology (for example, wireless charging of handheld devices on a 'charging mat' or for power transfer in an industrial or commercial environment, such as in wind turbines).

IPT systems will typically include an inductive power transmitter and an inductive power receiver. The inductive power transmitter includes a transmitting coil or coils, which are driven by a suitable transmitting circuit to generate an alternating magnetic field. The alternating magnetic field will induce a current in a receiving coil or coils of the inductive power receiver. SUMMARY

The present invention may provide an improved inductive power transmitter or which provides the public with a useful choice.

According to an example embodiment there is provided an inductive power transmitter comprising: an LCL circuit which includes a transmitter coil; and a power inverter including a plurality of control devices configured to drive the LCL circuit; wherein the frequency of a plurality of drive signals for the respective control devices is determined based on a desired power factor for the LCL circuit.

It is acknowledged that the terms "comprise", "comprises" and "comprising" may, under varying jurisdictions, be attributed with either an exclusive or an inclusive meaning. For the purpose of this specification, and unless otherwise noted, these terms are intended to have an inclusive meaning - i.e. they will be taken to mean an inclusion of the listed components which the use directly references, and possibly also of other non-specified components or elements.

Reference to any documents in this specification does not constitute an admission that those documents are prior art or form part of the common general knowledge. BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings which are incorporated in and constitute part of the specification, illustrate embodiments of the invention and, together with the general description of the invention given above, and the detailed description of embodiments given below, serve to explain the principles of the invention, in which:

Figure 1 is a block diagram of an inductive power transfer system;

Figure 2 is a circuit diagram of an example transmitter;

Figure 3 is a further circuit diagram of an example transmitter;

Figure 4 is a graph of example voltage waveforms;

Figure 5 is graph of a transmitter with a lagging power factor;

Figure 6 is graph of a transmitter with zero voltage switching;

Figure 7 is graph of a transmitter with a leading power factor;

Figure 8a is a circuit diagram of a system with 0.26 coupling;

Figure 8b is a circuit diagram of a system with 0.5 coupling;

Figure 9a is a graph of a system with 0.3 coupling;

Figure 9b is a graph of a system with 0.6 coupling;

Figure 10 is a schematic of a first control strategy;

Figure 11 is a schematic of a second control strategy; and

Figure 12 is a schematic of a third control strategy. DETAILED DESCRIPTION

An IPT system 1 is shown generally in Figure 1. The IPT system includes an inductive power transmitter 2 and an inductive power receiver 3. The inductive power transmitter 2 is connected to an appropriate power supply 4 (such as mains power or a battery). The inductive power transmitter 2 may include transmitter circuitry having one or more of a converter 5, e.g., an AC-DC converter (depending on the type of power supply used) and an inverter 6, e.g., connected to the converter 5 (if present). The inverter 6 supplies a transmitting coil or coils 7 with an AC signal so that the transmitting coil or coils 7 generate an alternating magnetic field. In some configurations, the transmitting coil(s) 7 may also be considered to be separate from the inverter 5. The transmitting coil or coils 7 may be connected to capacitors (not shown) either in parallel or series to create a resonant circuit. Additional coils may be provided, for example in an LCL configuration.

A controller 8 may be connected to each part of the inductive power transmitter 2. The controller 8 may be adapted to receive inputs from each part of the inductive power transmitter 2 and produce outputs that control the operation of each part. The controller 8 may be implemented as a single unit or separate units, configured to control various aspects of the inductive power transmitter 2 depending on its capabilities, including for example: power flow, tuning, selectively energising transmitting coils, inductive power receiver detection and/or communications.

The inductive power receiver 3 includes a power pick up stage 9 connected to power conditioning circuitry 10 that in turn supplies power to a load 11. The power pick up stage 9 includes inductive power receiving coil or coils. When the coils of the inductive power transmitter 2 and the inductive power receiver 3 are suitably coupled, the alternating magnetic field generated by the transmitting coil or coils 7 induces an alternating current in the receiving coil or coils. The receiving coil or coils may be connected to capacitors (not shown) either in parallel or series to create a resonant circuit. Additional coils may be provided, for example in an LCL configuration.

In some inductive power receivers, the receiver may include a controller 12 which may control tuning of the receiving coil or coils, operation of the power conditioning circuitry 10 and/or communications.

The term "coil" may include an electrically conductive structure where an electrical current generates a magnetic field. For example inductive "coils" may be electrically conductive wire in three dimensional shapes or two dimensional planar shapes, electrically conductive material fabricated using printed circuit board (PCB) techniques into three dimensional shapes over plural PCB 'layers', and other coil-like shapes. Other configurations may be used depending on the application. The use of the term "coil", in either singular or plural, is not meant to be restrictive in this sense.

Various different topologies for inductive power transfer (IPT) are used depending on the application. For example the inductive power transmitter 2 and/or the inductive power receiver 3 may be resonant or non-resonant. Resonant power transfer has the advantage that the range of power transfer can be increased, compared to non-resonant transfer.

Resonant topologies may include series resonant circuits or parallel resonant circuits. Another topology is LCL, which is a combination of both series and parallel tuning that uses at least two inductors and at least one capacitor. Each option has pros and cons. From the point of view of the inverter 6, a series tuned transmitting coil or coils 7 can appear as a low impedance load at its tuned frequency and a high impedance load at higher frequencies due to the series inductive element. Because of this the series tuned transmitting coil or coils 7 can draw significant current from the inverter 6 when being driven at the resonant frequency, even when a low inverter 6 output voltage is used. A consequence of this is that, when driven at a fixed voltage, for example from a voltage source inverter 6 such as a full bridge inverter, a series tuned transmitting coil or coils 7 can build up a large and uncontrolled resonant voltage and current, which risks damaging the inverter 6 and the inductive power receiver 3. Furthermore, all of the current which flows in the transmitting coil or coils 7 also flows through the inverter 6, which can lead to lower efficiency of the inductive power transmitter 2.

On the other hand from the point of view of the inverter 6, a parallel tuned transmitting coil or coils 7 can appear as a high impedance load at its tuned frequency and a low impedance load at higher frequencies due to the parallel capacitive element. This is inherently not well suited for use with voltage source inverters 6 such as a full bridge inverter, because the voltage source inverter will typically try to force its output voltage to rapidly switch from one value to another. This rapidly changing voltage in parallel with the parallel tuning capacitor of the transmitting coil or coils 7 can result in large transient currents which can damage the inverter 6.

An LCL tuned transmitting coil or coils 7 has a combination of the desirable input and output characteristics of the parallel and series tuned transmitting coil or coils 7 and also has fewer downsides. For example, with an LCL tuned transmitting coil or coils 7 when an inductive power receiver 3 is not present, the inductive power transmitter 2 can maintain a current flowing in the transmitting coil or coils 7 without needing to switch high currents through the switches of the inverter 6, even when the inverter 6 is a voltage source inverter. An LCL tuned transmitting coil or coils 7 may allow power transfer at unity power factor when driven with a voltage source inverter. That is the real power flowing into the LCL tuned transmitting coil or coils 7 network from the inverter 6 can be substantially equal to the apparent power.

According to an example embodiment, an IPT system 1 can be implemented with an LCL tuned power transmitting coil or coils 7 and an LCL tuned power pick up stage 9. When applied to IPT, in particular to a rotating IPT coupling for industrial applications eg: a wind turbine, it may be desirable to operate the transmitter and receiver in a tuned manner. This may reduce the reactive current flowing in the inverter 6 circuitry by operating at close to unity power factor, which may increase efficiency and allow for lower component ratings. It may also enable zero voltage switching in relation to the inverter 6 and/or any rectifiers within the power conditioning circuitry 10, which may further increase efficiency and/or reduce component stress.

In order to maintain close to unity power factor and/or minimise reactive current flow, it may be necessary to make adjustments to keep the LCL circuits of the inductive power transmitter 2 and inductive power receiver 3 tuned in spite of changes in the circuit or environment. One example of a change is a change in the coupling coefficient between the transmitter coil or coils 7 and the receiver coil or coils. This may happen due to a change in distance between the inductive power transmitter 2 and the inductive power receiver 3. Retuning of the tuned networks can be done in a numbers of ways; the particular methodology can be chosen according to the requirements of the application. One example for retuning is to modify the IPT frequency of the transmitter to the new "resonant" frequency, or to another frequency somewhat above or below the resonant frequency as required by the application. Changes in the transmitting frequency can be used to tune the transmitting coil or coils 7, the power pick up stage 9 or to tune some combination of the two.

Figure 2 shows an example of an inverter 6 which uses LCL tuned circuit 201. The inverter includes switches SI 202, S2 203, S3 204 and S4 205. A DC source 206 is shown here for simplicity but would in practice be provided by the converter 5 or from a power supply 4. A transmitting coil or coils 7 is connected first in series with a series tuning capacitor 207, the combination of which is then connected in parallel with a parallel tuning capacitor 208. This is then connected to a series tuning inductor 209 which is in turn connected to switches S2 203 and S4 205 of the inverter 6. Switches SI 202 and S3 204 drive the first leg 210 of the LCL tuned circuit 201 and switches S2 203 and S4 205 drive the second leg 211. The driving voltage 212 to the LCL tuned circuit 201 is the voltage difference between the first leg 210 and the second leg 211. The current flowing into the LCL tuned circuit 201 flows through the series tuning inductor 209 and is denoted the driving current 213. The body diodes 214 of the switches SI 202, S2 203, S3 204 and S4 205 are explicitly shown however in practice these will typically be a part of the switches, which will typically be MOSFETs. However, it is also possible to use other device types such as but not limited to bipolar junction transistors, silicon carbide FETs, gallium nitride FETs or IGBTs.

An LCL tuned circuit 201 should comprise a transmitting coil or coils 7, a parallel tuning capacitor 208 and a series tuning inductor 209. The values of the components should be chosen for a resonant frequency. This tuned frequency may for example be the designated IPT frequency, or frequency range, or it may be shifted from that depending on the application requirements. In order for the LCL circuit to be tuned at the resonant frequency, the reactance of each branch should be matched. For example the series tuning inductor 209 is the same reactance as parallel tuning capacitor 208 which must be the same reactance as the combination of transmitting coil 7 and the series tuning capacitor 207. It is also possible to split one or more of the reactive components to create a more symmetrical circuit, for example, splitting the series tuning inductor 209 into two tuning inductors each of half the inductance value, one connected to SI 202 and S3 204 and the other in the present position of the series tuning inductor 209. Further another capacitor could be placed in series with series tuning inductor 209 and/or the tuning of the LCL circuit may be detuned, depending on the application requirements.

It is advantageous to use a full bridge inverter or a half bridge inverter to drive an LCL tuned circuit 201, because it allows for an inductive power transmitter 2 that is as simple as possible and has a low component count. If DC power is fed directly into the inductive power transmitter 2, a separate converter 5 may not be required because the effective duty cycle of the inverter 6 can be changed in order to control the amplitude of the current flowing in the transmitting coil or coils 7. In this way, high efficiency of the IPT system 1 can be achieved, irrespective of input voltage or changes in the coupling coefficient between the inductive power transmitter 2 and the inductive power receiver 3. In practice, efficiencies of better than 90% have been achieved even with 10% input voltage variation and with coupling coefficients that vary between 0.27 and 0.6 Furthermore, this full bridge inverter implementation of the inductive power transmitter 2 does not require a bulky DC inductor as is common in other DC converter 5 or inverter 6 designs. While a series tuning inductor 209 is still required, this is typically much smaller than a DC inductor would be.

Figure 3 shows an implementation of an LCL tuned transmitting coil or coils 7 with a full bridge inverter 6. Waveforms for a possible mode of operation of this circuit are shown in Figure 4. In this implementation all of switches SI 202, S2 203, S3 204 and S4 205 run at approximately 50% duty cycle with a small dead- time. The switches that are connected to a given leg of the LCL tuned circuit 201, for example SI 202 and S3 204 for the first leg 210, are operated at 180 degrees out of phase from each other. The effective duty cycle of the inverter 6 is controlled by changing the phase angle between drive voltages applied to the switches on one side of the bridge (202 and 204) versus the other side of the bridge (203 and 205). In other words controlling the phase of the input to the MOSFETs which controls the duty cycle. When the voltages are applied at 180 degrees out of phase from each other the inverter 6 is then running at a 50% effective duty cycle. When they are in phase then full bridge is running at 0% effective duty cycle and both ends of LCL will be simultaneously either at 24V or at 0V. By varying the effective duty cycle the inductive power transmitter 2 can control the power being transmitted. For example, a low effective duty cycle can be ideal for light loads and/or when the transmitting coil or coils 7 and the power pick up stage 9 are tightly coupled. The driving voltage 212 can be approximated as the effective duty cycle multiplied by the voltage of the DC source 206.

The power factor of the LCL tuned circuit 201 is the phase angle between the driving voltage 212 and the driving current 213. A leading power factor means the current leads the voltage by some phase and implies a capacitive load, while a lagging power factor means current lags the voltage and implies an inductive load.

Designing the LCL tuned circuit 201 to operate at a certain power factor can be advantageous. For example, it can be beneficial to operate the LCL tuned circuit 201 with a unity power factor. That is, the voltage and current seen from the inverter 6 across or through the first leg 210 and the second leg 211 are in-phase. Maintaining the power factor at or close to unity is desirable because it means that the current through the series tuning inductor 209 is minimised, which helps to reduce power losses on this inductor. Furthermore, the inverter 6 is able to use zero current switching for SI 202, S2 203, S3 204 and S4 205 which helps to reduce switching losses. Still further, a unity power factor helps minimize the amount of VARs that the inverter 6 needs to provide or absorb, since the amount of reactive power provided by the inverter 6 into the LCL tuned circuit 201 is minimized. These factors help to improve the efficiency of the inverter 6 and the LCL tuned circuit 201, while minimizing the size, the voltage ratings and the current ratings of their constituent components.

Making the LCL tuned circuit 201 run at a desired power factor can be difficult to achieve in practice, for a number of reasons. It can be difficult and expensive to source capacitances and inductances with narrow tolerances for the reactive components within the LCL tuned circuit 201. This means that the combination of components that make up the LCL tuned circuit 201 can easily be out of tune. This mis-tuning effect can be made worse by changes in temperature or aging. A further problem observed by the inventor with designing the LCL tuned circuit 201 to operate at a specific power factor was that many system parameters, which often vary, can affect the power factor of the LCL tuned circuit 201. For example the value of the load 11, the coupling coefficient between the transmitting coil or coils 7 and the power pick up stage 9 and the introduction of foreign metallic objects can all affect the power factor of the LCL tuned circuit 201. Also the inductance of the transmitter coil 7 will change under different conditions.

A possible solution to achieving the desired power factor for the LCL tuned circuit 201 in spite of component variation and system changes is to adjust the values of the reactive components in the LCL tuned circuit 201 during operation of the IPT system 1. For example, additional inductance or capacitance can be switched into or out of the LCL tuned circuit 201 so that it operates with unity power factor. The use of saturable inductors can also allow the LCL tuned circuit 201 to be re-tuned during use. The drawback of these solutions is that they add additional cost, complexity and bulk to the inductive power transmitter 2.

Typically, with a series tuned or a parallel tuned transmitting coil or coils 7, increasing the driving frequency above the resonant frequency of the transmitting coil or coils 7 while holding the driving voltage constant will result in more reactive power flowing in the transmitting coil or coils 7 and less real power. However, in the case of an LCL tuned circuit 201, increasing frequency of the driving voltage 212 to a value that is somewhat above the resonant value for the LCL tuned circuit 201 can cause the reactive power flowing in the transmitting coil or coils 7 to increase, and the real power increases as well. By taking advantage of this relationship between the driving frequency and the reactive power flowing in the transmitting coil or coils 7 it is possible to account for component variation or changes in system parameters so that high power transfer efficiency and reliability can be achieved, with a minimum of additional components.

It may also not always desirable to operate the LCL tuned circuit 201 at unity power factor. It also may be desirable to change the operating power factor during operation. In some cases it may be desirable to run the LCL tuned circuit 201 at a lagging power factor as shown in Figure 5. For example, by running the LCL tuned circuit 201 at a lagging power factor, zero voltage switching (ZVS) in the full bridge can be achieved. Soft switching is important in order to both reducing the power loss in the switches SI 202, S2 203, S3 204 and S4 205 and in reducing the high frequency electromagnetic noise that would introduced by hard switching. For example in the embodiment shown in Figure 2, the body diodes 214 in the full bridge inverter 6 can be made to conduct before the switches SI 202, S2 203, S3 204 and S4 205 are turned on which prevents the overlap of high current and voltage in the switches. This reduces switching losses. Switching losses can be approximately halved when ZVS is used compared with when no ZVS is used. ZVS will only occur under specific conditions for the circuit topology shown in Figure 2, ZVS can be obtained by controlling the power factor of the LCL to be slightly lagging.

In an example embodiment, shown in Figure 6, zero voltage switching can occur when D ~ 0.5. Before t = to, switch S2 and S3 were conducting. At t = to, S2 and S3 are turned off and a gate pulse is applied to SI and S4 . Since iL is negative at this instant, it flows through Dl and D4 . At t = tl , Dl and D4 turn off naturally at zero current and iL now flows through SI and S4 . Similarly in the next half cycle, SI and S4 are turned off at t = t2 and a gate pulse is applied to S2 and S3 . Since iL is positive at this instant, it flows through D2 and D3. At t = t3 , D2 and D3 turn off naturally at zero current and iL flows through S2 and S3 . At t = t4 , S2 and S3 is turned off and SI and S4 is turned on once again marking the beginning of the next cycle. Thus, in this mode, the device conduction sequence is such that the antiparallel diodes conduct prior to the switch conduction resulting in ZVS turn- on for both the switches.

The controller 8 will typically be able to determine the driving voltage 212 from the combination of the gate drive signals it creates as well as the value of the voltage coming from the DC source 206. However the phase and magnitude of the driving current 213 may be unknown to the controller 8 and this information is useful in measuring the power factor of the LCL tuned circuit 201. A current transformer can be used for current detection, placed for example in series with the series tuning inductor 209.

The inverter 6 and the LCL tuned circuit 201 can also be run with so that the LCL tuned circuit 201 has a leading power factor as shown in figure 7. While a slightly lagging power factor can be used to achieve maximum efficiency, higher power can be obtained when operating at a leading power factor and a higher frequency. An increase in frequency will act to decrease the overall impedance seen by the inverter 6 looking into the LCL tuned circuit 201 and increase the current in the transmitting coil or coils 7, thus providing a higher potential maximum output power. For example, in one test it was observed that at an operating frequency of 200kHz the current flowing in the transmitting coil 7 was 5.02A while at 210kHz the current flowing was 5.52A. Because the maximum possible power transfer is proportional to the amount of reactive power flowing in the transmitting coil or coils 7, according to VARs = I ² CJL, and because both the angular frequency and the current have increased, this corresponds to an increase in the reactive power and in the maximum possible power transfer of about 27%.

Mathematically, at higher frequency, the parallel combination of C2 and Ll/Cl will present a more capacitive reactance but a lower resistance based on equation 1:

1/(1/(R+ XC1+XL1)+1/XC2) (1)

Overall the impedance of the LCL can then be given by equation 2:

It can be shown that an increase in frequency will act to decrease the overall impedance seen by the supply and increase the current in the Tx coil, thus providing a higher maximum power. At a frequency of 210kHz the calculated current in Lp_tx is 5.52A. This means that a higher frequency, may achieve a higher power transfer. However, efficiency decreases to achieve this power boost due to operating at a higher Q (higher loss in the Tx and Rx power coils). A leading power factor also prevents ZVS and so the bridge may become noisier and less efficient.

It is possible that this method may be used to provide a temporary power boost (or allow the system to operate at a lower k) for a short period of time.

Preliminary tests show that this method can be used to allow a prototype system to operate at 20mm (k=0.15) at full power (100W). Efficiency at this operating point is much lower at about 80%.

The ideal power factor or phase difference varies depending on the distance between the transmitting coil or coils 7 and the power pick up stage 9. For example, in one test, at 15mm separation a leading power factor is ideal in order to allow enough power transfer to the receiver. At smaller separations, a unity or lagging power factor may be more efficient.

Figures 8a and 8b show the changes to the effective inductances of the transmitting coil or coils 7 and a receiving coil or coils 801 that occur when the distance between the transmitting coil or coils 7 and the receiving coil or coils 801 is changed, or alternately when the coupling coefficient is changed. In this example, an LCL tuned circuit is used for both the inductive power transmitter 2 and the inductive power receiver 3, which also includes a series tuning capacitor 802, a parallel tuning capacitor 803 and a series tuning inductor 804. The values shown are for coupling coefficient changing from 0.26(Figure 8a) to 0.5(Figure 8b). When the reactance of each branch is matched, the LCL is correctly tuned and will operate at unity power factor. It can also be tuned to operate at slight lagging power factor in order to achieve ZVS. However, when coupling coefficient between the transmitting coil or coils 7 and the receiving coil or coils 801 changes, the reactance of the transmitting coil or coil 7 also changes, so the system can no longer operate at the correct power factor if it is operating at a fixed frequency and in the absence of other corrective circuit changes.

In one embodiment the LCL tuned circuit becomes detuned when the coupling between the transmitting coil or coils 7 and the receiving coil or coils 801 is high. This causes large currents to flow through the series tuning inductor 209 of the inductive power transmitter 2 as well as the equivalent component which may be present in the inductive power receiver 3. This can cause damage to the IPT system 1 and can result in low efficiencies. Sometimes this effect can even be present in the case of moderate coupling coefficients between the transmitting coil or coils 7 and the receiving coil or coils 801.

In the detuned state a large reactive power flowing through the full bridge switches into the de-tuned LCL which means a high switching current in MOSFETs resulting in high loss, poor efficiency and possible damage. In order to avoid this, the frequency response of the LCL tuned circuit 201 can be examined at different operating points to see how changing the switching frequency will affect the system.

The frequency response of the system changes with changing coupling and changing load. Also, the load reflected by the receiver is not constant and is likely to be non-linear given how the shorting control regulates the Tx output voltage.

The frequency response of the current in the Tx LCL changes with changing coupling between 0.3 in Figures 9a and 0.6 in Figure 9b. In particular, the frequency 902 where unity power factor can be achieved decreases with increased coupling eg: 200kHz @ 0.3 and 190kHz @ 0.6. It may be desirable to control the power factor to be slightly lagging under all conditions of load and coupling. This achieves zero voltage switching as well as keeping the reactive power provided by the supply to a minimum. The

The controller 8 can derive the phase of the driving voltage 212 because the controller 8 produces the gate drive signals controlling the inverter 6. The controller can obtain the phase of the driving current 213 by using a current transformer placed in series with the series tuning inductor 209 and comparing the secondary side output of the current transformer to ground with a fast comparator for example. By then making changes to the switching frequency the controller can maintain a constant phase angle or power factor angle between the driving voltage 212 and the driving current 213.

A control strategy for the circuit shown in Figure 2 is shown in Figure 10. In this case the controller simply controls the frequency up or down to achieve a desired power factor or phase between the voltage and current.

Figure 11 shows an alternative controller 8 that controls the power transfer as well as the power factor. In this case the transmitter is independent in that it does not need to communicate directly with the inductive power receiver 3. The controller 8 is able to control the duty cycle based on measured values within the inductive power transmitter 2. For example, as the input voltage varies from 21.6V to 26.4V the duty cycle can be decreased linearly from 50% to 35%.

The controller 8 may control the frequency to maintain a given power factor such as a certain lagging power factor for the LCL tuned circuit 201. In this case, the controller 8 can estimate the approximate separation between the transmitting coil or coils 7 and the receiving coil or coils 801 from the frequency that is required to maintain the specific power factor. If the operating frequency is low, it means the inductive power receiver 3 is closer and so the duty cycle can be reduced. For example the maximum duty cycle at 200kHz (at 21.6V) could be 50% while the maximum duty cycle at 180kHz could be 35%. The opposite applies if the operating frequency increases.

The duty cycle of the inverter 6 can be chosen by estimating the load 11 by monitoring the input power from the power supply 4 or through the converter 5. For example, for an input power of HOW the duty cycle can be set to the 50% for maximum output power while when the input power is 30W the duty cycle can be set to 30% for reduced output power. Under some circumstances the inductive power receiver 3 may fall out of regulation due to insufficient received power but the input power to the inductive power transmitter 2 may remain be quite high.

Figure 12 shows a further alternative controller 8 wherein the receiver communicates information about output voltage, for example, the quality of the regulation and/or the receiver series tuning inductor current. Having information from the receiver would allow better control including duty cycle control and more versatile frequency control. The Transmitter then adjusts frequency and duty cycle to provide the most efficient operating point.

Using the method of control in Figure 11 power efficiency of above 90% over the full coupling range (4-14mm) was able to be achieved in a prototype, with peak efficiency of 92.5% occurring at 10mm. Without any frequency control the system didn't work closer than about 8mm and was much less efficient at this distance. The operating frequency of the transmitter currently varies between 204kHz and 180kHz to maintain the desired output power and phase angle. This control method comes at very little extra cost with just one extra comparator and the CT being moved from the Tx power coil, compared to dynamically switching in capacitors or using a saturable inductor. While the present invention has been illustrated by the description of the embodiments thereof, and while the embodiments have been described in detail, it is not the intention of the Applicant to restrict or in any way limit the scope of the appended claims to such detail. Additional advantages and modifications will readily appear to those skilled in the art. Therefore, the invention in its broader aspects is not limited to the specific details, representative apparatus and method, and illustrative examples shown and described. Accordingly, departures may be made from such details without departure from the spirit or scope of the Applicant's general inventive concept.