The invention also relates to an arrangement defined in the preamble of claim 5 for stabilizing the operation of a switched-mode power supply.

A switched-mode power supply, or switcher, is a converter in which an input voltage or current is converted, or"chopped", by means of switches into an output voltage or current the magnitude and quality of which are suitable for the load.

The"flyback"-type switched-mode power supply, or switcher, shown in Fig. 1 re- presents the prior art. Fig. 2 shows signal waveforms found in the switcher depicted in Fig. 1. The switcher comprises a transformer 101 and switch 102 which is coupled to the primary winding 101 a of the transformer 101. The switcher addition- ally comprises a diode 106 and a capacitor 107 functioning as an energy store, which both are coupled to the secondary winding 101b of the transformer 101.

Electric energy is supplied to the circuit of the primary winding 101a of the trans- former, and said electric energy is consumed by the load 108 coupled to the secondary winding 101b of the transformer. The switcher also comprises a control unit 109 to control the switch 102. The control unit in turn comprises a controller 104, which controls the switch 102. and an oscillator 103 the output of which is coupled to the controller 109. The switcher further comprises a feedback circuit 105 from the circuit of the secondary winding 101b of the transformer 101 to the circuit of the primary winding 10 la.

The switcher in Fig. 1 operates as follows. A direct voltage Vi is supplied to the switcher input IN, and a second direct voltage Vo. the value of which is lower than that of the input voltage Vi. is taken from the switcher output OUT. The input voltage Vi is generated e. g. by rectifying an alternating voltage. When switch 102 conducts, the current il through the primary winding 101a of the transformer 101 rises linearly (Fig. 2), and energy is stored in the transformer 101. Diode 106 is at that point reverse-biased in relation to the voltage induced in the secondary winding 101b. When switch 102 is made non-conductive, current i\ drops to zero. The direction of the change in the magnetic field of the transformer 101 is now such that diode 106 is forward-biased in relation to the voltage induced in the secondary winding 101b. and the energy accumulated in the transformer is discharged into the circuit coupled to the secondarv winding lOlb. At first. energy is mainly transferred

to a capacitor 107 coupled in parallel with the secondary winding 101b so that the charge of the capacitor grows and the voltage across it, or the output voltage Vo, increases (Fig. 2). A load 108 at the output OUT, coupled in parallel with capacitor 107, results in a steady decrease in voltage Vo; the bigger the load, the lower the voltage of the load between the charge cycles of capacitor 107. The switcher operates at a fixed frequency; switching frequency is determined by oscillator 103.

Switch 102 is made conductive at the beginning of each cycle as shown in Fig. 2 (voltage Vc into state"1", or high). A switch controller 104 monitors, through a feedback circuit 105, how the output voltage Vo changes and makes switch 102 non-conductive (Vc into state"0", or low). The smaller the output voltage Vo on average, the longer the time that switch 102 will be closed, or conductive, at a time.

Thus, the closed position of switch 102 is controlled according to the width of its control pulse. The converter thus applies pulse width modulation. The feedback circuit 105 provides galvanic isolation between the circuit of the transformer's secondary winding 101b and switch 102 and, in general, its control unit 109.

Switchers of the type described above may experience functional problems at small loads. Because of the delays of switch 102 and control unit 109, current pulses generated by the switch have a certain minimum length even while the switcher is idling. If load 108 is missing or is very small, the output voltage Vo rises. In a typical construction this results in that at times switch 102 gets no control; cycles are randomly missed, and the output voltage fluctuates more than allowed.

One way of avoiding the instability described above is to have the switcher operate at a variable frequency: As the load gets smaller, so will the switching frequency.

This way, fluctuation of the output voltage Vo will be under control. A disadvantage of this technique is that as the switching frequency varies, so will the noise spectrum, and noise components cannot be attenuated with filters as is done in a fixed-frequency switcher.

Another, more widely used, method of avoiding switcher instability at small loads is to add loading such that the total load is never too small for stable operation of the switcher. In a simple case, the additional load comprises a resistor permanently coupled in parallel with the load proper. A disadvantage of this method is the continuous power dissipation caused by the additional load which degrades the efficiency of the switcher.

In some switchers the additional load comprises, instead of a single resistor. a circuit including active components as well, i. e. an active load with variable resist-

ance and, hence, variable dissipation power. The active load comes on when the load proper reaches a certain small enough value and grows while the load proper further decreases. The magnitude of the additional load in idle state depends on the type of the switcher. It is at its greatest with flyback-type switchers (up to 10% of maximum load). Control for the active load requires knowledge of the magnitude of the load current proper. In known switchers that knowledge is obtained using a current-measuring resistor coupled in series with the load. A disadvantage of a current-measuring resistor is the power dissipation caused by it. The power dissi- pation can be minimized by choosing as small as possible a resistance for the current-measuring resistor. However, current-measuring resistors with very small resistances have the disadvantage that they require precise tuning and that the control for the active load is more sensitive to spurious effects than when using greater resistances. Control reliability with small current-measuring resistances is thus poorer. In principle, losses caused by current measuring could be minimized by using a current-measuring transformer, but this solution would mean more com- plicated circuit technology and added costs.

An object of the invention is to eliminate the disadvantages mentioned above. The invention enables stable operation of a switcher while idling and at small loads without increasing dissipation power in a normal load situation and without increasing ambient noise levels.

The method according to the invention is characterized by what is expressed in claim 1. The arrangement according to the invention is characterized by what is expressed in claim 5. Other claims define preferred embodiments of the invention.

The basic idea of the invention is as follows. The active load control uses a signal dependent on the magnitude of the load, which signal is primarily generated for controlling the switch in the switcher. A simple controller converts this signal into an active load control signal such that the dissipation power of the active load is at its greatest when the switcher is idling and the dissipation power advantageously drops down substantially to zero when the load proper in the switcher reaches e. g.

5% of the maximum load. The controller is advantageously slowed down, lest changes in the feedback signal during a switching cycle affect the active load.

An advantage of the invention is that stable operation of the switcher is achieved without the additional load causing power dissipation in a normal load situation.

Another advantage of the invention is that the controller of the active load, which provides the additional load, is simple, reliable and has low manufacturing costs. A

further advantage of the invention is that the noise spectrum of the switcher is relatively stable so that spurious frequency components can be attenuated using narrow-band filtering.

The invention will now be described in more detail. Reference will be made to the accompanying drawings wherein Fig. 1 shows a schematic of a switcher according to the prior art, Fig. 2 shows waveforms generated by the structure depicted in Fig. 1, Fig. 3 shows in the form of circuit diagram a switcher according to the invention, Fig. 4 shows in the form of circuit diagram an additional load applied in the switcher of Fig. 3, and Fig. 5 graphically illustrates characteristics of the additional load according to Fig. 4.

Figs. 1 and 2 were already discussed in connection with the description of the prior art.

Fig. 3 shows in the form of circuit diagram an implementation of the switched-mode power supply according to the invention. In its basic structure it largely corresponds to the switched-mode power supply depicted in Fig. 1. The circuit diagram does not show components, such as overcurrent protection and soft start circuits, which are of no significance to the invention, and those parts that are shown, are shown in a simplified manner.

The switcher comprises a transformer 301 and switch 302 which is placed in the circuit of the primary winding 301a of the transformer 301. The switcher also includes a control unit 309 to control switch 302. The control unit 309 in turn comprises a controller 304 for driving switch 302, and an oscillator 303 the output of which is coupled to the controller 304. The switcher further comprises a feedback circuit 305 to provide feedback from the circuit of the secondary winding 301b of the transformer 301 to the circuit of the primary winding 301a. The control unit 309 also includes a current-measuring unit 340 for the primary winding 301a. The switcher includes an additional load 400 placed in the circuit of the primary winding 301a. In addition, the switcher includes a diode 306 and an energy-storing capacitor 307, both of which are placed in the circuit of the secondary winding 301b of the transformer 301. The purpose of components 311 at the switcher input, i. e. coil L2 and capacitor C6, is to suppress noise in the input voltage Vj. In addition. the circuit

of the primary winding 301a of the transformer 301 includes a circuit 312, comprising a diode D2, capacitor C3 and resistor R1, for damping voltage spikes in the primary. The circuit of the secondary winding 301b includes a filter 313 realized with a coil LI and capacitor C2. Electric energy is supplied to the transformer's primary winding 301a, and said electric energy is consumed by the load 308 coupled to the transformer's secondary winding 301b.

In addition to said two windings, i. e. primary and secondary windings 301a, 301b, the transformer includes a third, additional winding 301c. Power supply unit 310 for the control unit 309 comprises the third winding 303 and diode D3, resistor R13 and capacitor C7. Power supply unit 310 generates the operating voltage Vcc for the control unit 309. Resistor R12 helps start the switcher when the supply voltage Vi is turned on.

The feedback circuit 305 comprises a regulator 332 and an opto-coupler 333. The input of the feedback circuit 305 is coupled to the circuit of the secondary winding 301b of the transformer 301 before filter 313. Resistors R15, R16 realize a voltage divider to decrease the feedback voltage vl obtained from the secondary circuit of the transformer 301 in such a manner that the regulator 332 gets from the switcher output a suitable current to control the opto-coupler 333. Thus the voltage across resistor R2 at the other side of the opto-coupler 333, i. e. the feedback circuit output voltage v2, is proportional to voltage vl.

Controller 304 comprises an amplifier unit 334, comparator 335 and a memory unit 336. The amplifier unit 334 and comparator 335 are advantageously operational amplifiers. The output voltage v2 of feedback circuit 305 is brought to a first input of amplifier unit 334, in this particular application to the inverting input. In connection with the first input of amplifier unit 334 there is a matching network comprising resistors R3, R2 and feedback resistor R5 to amplify and invert voltage v2. To the resulting voltage it is added a suitable constant voltage which depends on a fixed reference voltage Vref brought to a second input, in this particular case the non-inverting input, of amplifier unit 334. Feedback circuit 339 of amplifier unit 334 further includes series-connected resistor R4 and capacitor C4, coupled in parallel with feedback resistor R5. The capacitance for capacitor C4 is chosen such that the gain of amplifier unit 334 starts to drop at frequencies which are an order of magnitude smaller than the switching frequency of switch 302 (which corresponds to the operating frequency of oscillator 303). Output voltage v3 from amplifier unit 334 is taken to a shaping circuit 338 where it is shaped such that output voltage V4 from the shaping circuit 338 is near zero when the load of the switcher is small, and

increases as the load increases (because vl decreases) at a suitable rate. In this case the shaping circuit 338 comprises a diode D4 and resistor R6, connected in series, and a resistor R7 coupled to ground. The output of the shaping circuit 338 between resistors R6, R7 is coupled to one input of comparator 335.

The output of oscillator 303 is coupled to the setting input S of a memory unit 336, which advantageously is a two-state memory circuit. According to its operating frequency the oscillator 303 sends at regular intervals a setting pulse to the memory unit 336 so that output voltage v7 from output Q of the memory unit 336 is set to logical"1", or high in this application, at the beginning of each duty cycle, as was mentioned above. Thereby, via a buffer 337 and resistors R10 and Rll, it also drives the field-effect transistor (FET) functioning as a switch 302 into conduction at the beginning of each duty cycle. The current i 1 through the primary winding 301a of the transformer 301 starts to flow through switch 302 and grows linearly.

The current il is measured by a current-measuring unit 340 comprising a resistor R8 which advantageously has a small resistance. Resistor R8 is connected in series with switch 302. The current-measuring unit 340 is followed by a low-pass filter 341 the output of which is coupled to one input of comparator 335. Resistor R8 is connected in series with switch 302. The voltage across resistor R8 is filtered by a low-pass filter 341 which comprises a resistor R9 and capacitor C5. They produce a voltage V5 proportional to the current il through the primary winding.

To a first input, specifically the inverting input, of comparator 335 and to a second, specifically the non-inverting input of comparator 335 it is brought, respectively, output voltage V4 from the shaping circuit 338 and output voltage V5 from the current-measuring unit 340 which is proportional to the current il through the primary winding 301a. Comparator 335 is used to determine the ratio of voltage V5, which is proportional to the current il flowing through the primary winding, and voltage V4, which depends on the switcher's load level. At the beginning of each duty cycle voltage V4 is greater than voltage V5, and output voltage v6 of comparator 335 is high. When during the duty cycle voltage V5 exceeds voltage V4, output voltage v6 of comparator 335 drops low. The output of comparator 335 is coupled to the resetting input R of the memory unit 336. When output voltage v6 changes states, the memory unit 336 is set to logical"0", i. e. output voltage V7 drops low. As a result, switch 302 is made non-conductive by output voltage V7 from the memory unit 336 via buffer 337. The smaller the resistance of load 308, the longer the time that will be spent in the duty cycle before voltage V5, which is proportional to the current il in the primary winding 301a, reaches output voltage V4 of the

shaping circuit 336, and the greater the amount of energy that is transferred through the switcher's primary winding 301a to the secondary winding 301b and further to load 308 in each duty cycle. The duration of the logical"1"state, or state"high", of the memory unit's 336 output voltage V7 during each duty cycle, i. e. the length of the control pulse for switch 302, is thus controlled, and the control is realized using the voltage ramp v5 proportional to the current il flowing through the primary winding 301a. Here, too, pulse width modulation is applied to control switch 302. In addition, we refer to The Electronic Engineering Handbook, ed. Richard C. Dorf, CRC Press, 1993, pp. 711-713.

The basic operation of a switcher as described above does not include the use of an additional load 400. The additional load 400 comprises a controller 402 and the variable active load 401 proper. In this exemplary case the additional load 400 is coupled to the additional winding 301c of the transformer 301. The additional load 400 is controlled by the control unit 309, as shown in Fig. 3. Fig. 4 shows an advantageous implementation of the additional load 400, and its more detailed structure and operation are described below.

The controller 402 in the additional load 400 includes an amplifier 403. The amplifier 403 has two inputs, i. e. a first and a second input. To the first, inverting, input it is coupled a feedback circuit from the active load 401, comprising two resistors R19, R18, and a constant voltage Vs via resistor R17. To the second, non- inverting, input it is coupled an output voltage v3 from controller 304 of switch 302.

Amplifier 403 is advantageously a linear amplifier such as an operational amplifier.

The active load 401 proper, which in this case comprises a variable resistance, includes a current-measuring element 401a and current generator 401b, connected in series. The current-measuring element 401a comprises a resistor R20, and the current generator 401b comprises a transistor Ql. Voltage-reducing diodes D5, D6 are coupled between resistor R20 and transistor Ql. The active load 401 thus comprises a resistively loaded current generator. The feedback circuit of amplifier 403 is coupled between resistor R20 and the diode pair D5, D6 to provide the feedback voltage vg. Controller 402 regulates the current ia flowing through the active load 401 as well as its resistance, which depends on the output voltage V3 from controller 304 and on constant voltage Vs. Voltage v3 in turn depends on the switcher's load 308 in such a manner that it grows linearly as the resistance of the load decreases. In amplifier 402 voltage V3 is amplified and a certain constant voltage, which depends on voltage Vs, is subtracted from it. Voltage Vs is a reference voltage generated from the operating voltage Vcc. Thus the resulting

output voltage v9 of controller 402 increases as the load increases. Values for resistors R17, R18 and R19 of amplifier 403 are chosen such that output voltage vg changes e. g. from value 0.5Vcc to value Vcc when load 308 grows from zero to a certain fraction of the maximum load, say, to 7% of the maximum load value (0.07 times the maximum load) in accordance with the voltage curve in Fig. 5a. In that case, the active load current ia changes according to curve 411 in Fig. 5b from a certain value, set by means of resistor R20, to zero when the load changes as described. The active load power P = ia'Vcc, so its change is similar to the change in current ia. Output voltage vg of amplifier 403 cannot become as large as the amplifier's operating voltage Vcc. The series-connected voltage-reducing diodes D5 and D6 are included for the purpose that the dynamic range of voltage vg be sufficient to control transistor Q 1 regulating the load current ia.

Above it was described an example of the implementation of an active load in a switched-mode supply. The invention is not limited to the circuit arrangements described above, nor to the isolated current-mode"flyback"switcher described in the example. The inventional idea can be applied in all cases where there is need to provide a switched-mode power supply with an additional load that depends on the load proper.