TECHNICAL FIELD The present invention relates generally to ranging systems and in particular to ranging systems utilizing complex modulation techniques.

BACKGROUND

GNSS satellites transmit signals to be received by a universal receiver and to be able to estimate the Position Velocity and Time (PVT) solution. The communication method is the Direct-Sequence-Spread-Spectrum (DS-SS) with the Radio-Frequency (RF) carriers located in the L bands of the RF spectrum. Each satellite transmits the signal with a unique pseudo-random-noise (PRN) code, which is modulated onto the carrier. The Galileo E5 signal employs a carrier frequency of 1191.795 MHz (= ^■ >£* ) and a modulation type known as AItBOC. In an AItBOC modulation that uses the in-phase and quadrature phase of a complex subcarrier, up to four codes can be combined and modulated onto the carrier. The E5 signal uses this type of modulation for four codes called E5a-I, E5a-Q, E5b-I, and E5b-Q, each PRN code having a length 10230 and chipping rate 10.23 MHz. The sub-carrier frequency is 15.345 MHz. The result of this type of modulation is that the spectrum of the transmitted signal is split into two parts whose main lobes have approximately 20 MHz first null-null bandwidth and are centered around 1176.45 MHz (=^ ^5α ) and 1207.14 MHz (=/«*) frequencies respectively. These two lower and upper main lobes are referred to as E5a and E5b signals, respectively. A receiver using the Galileo E5 signals for PVT estimation can demodulate the

E5 signal received through a RF front-end having a two sided bandwidth of at least 51 MHz, so as to pass the two main lobes. A local replica of the AItBOC signal is generated and multiplied with the incoming signal to continuously estimate the Doppler frequency and code delay and track the satellite signal. Also, the receiver can independently process the E5a and E5b signals either through side-band translation or filtering techniques, to find the Doppler frequency and code delay. Consider a case where the incoming signal from the satellite in a GNSS receiver comprises a mix of the direct and reflected signals and where the direct signal is the desired signal and reflected signals are the non-desired signals. This phenomenon is known as 'Multipath fading'. A receiver cannot completely distinguish between the direct and reflected signal and hence the processing of the combined signal results in erroneous measurement of the code delay and the carrier Doppler frequency estimates, which in turn results in an erroneous PVT solution. The magnitude of the possible error due to the reflected signal can be obtained by analyzing the code phase and carrier phase measurements for different numbers, delays and magnitude of the reflected signals that get combined with the direct signal. New signal structures proposed to be used in the GNSS modernization focus on this issue and are designed to counteract multipath fading.

Multipath fading affects the performance of the GNSS receiver. Mitigation of the multipath has been a research focus for several decades and the problem still persists. Efforts to combat multipath have progressed in two major directions: the signal structure design and the receiver design. For signal structure design, reduction of the effects of multipath fading has been one of the main design criteria for the proposed signals in the GNSS modernization process. For receiver design, a number of techniques focus on different stages of the receiver signal processing chain to resolve the effect of multipath fading on the measurements.

The Galileo E5 signal transmitted at a carrier frequency centered around 1191.795 MHz is the most sophisticated signal among all the current modernized GNSS signals. A special modulation belonging to the class of offset-carrier modulations, known as Constant-Envelope AltBOC(15, 10), is used for the E5 signal. With a sub- carrier frequency of 15 x 1.023 MHz and code chipping rate of 1 Ox 1.023 MHz represented as AItBOC(15, 10), the E5 signal offers unprecedented performance with code tracking jitter less than 5 cm, even at a signal strength of 35 dB-Hz. Due to the code chipping rate and higher signal bandwidth, the AItBOC(15, 10) also helps in eliminating the long-range multipath effects on code phase measurements. However, with the standard Delay Locked Loop (DLL) and code discriminator architectures, the short-range multipath effects remain. SUMMARY

In accordance with an aspect of the invention, there is provided a method of processing in a receiver a signal generated using a complex modulation technique involving a spreading code and a subcarrier for a ranging system. The modulated signal comprising an upper sideband component, a lower sideband component, and a main component is received. Signal-to-noise-ratios of the upper sideband component, the lower sideband component, and the main component of the modulated signal received by the receiver are determined. Errors in carrier phases of the upper sideband component, the lower sideband component, and the main component of the modulated signal are determined. The presence of multipath is identified dependent upon the determined signal-to-noise ratios and the determined errors in the carrier phases. hi accordance with another aspect of the invention, there is provided an apparatus for processing in a receiver a signal generated using a complex modulation technique involving a spreading code and a subcarrier for a ranging system. The apparatus comprises: a module for receiving the modulated signal comprising an upper sideband component, a lower sideband component, and a main component; a module for determining signal-to-noise-ratios of the upper sideband component, the lower sideband component, and the main component of the modulated signal received by the receiver; a module for determining errors in carrier phases of the upper sideband component, the lower sideband component, and the main component of the modulated signal; and a module for identifying the presence of multipath dependent upon the determined signal- to-noise ratios and the determined errors in the carrier phases.

In accordance with yet another aspect of the invention, there is provided an apparatus for processing in a receiver a signal generated using a complex modulation technique involving a spreading code and a subcarrier for a ranging system. The apparatus comprises: a memory for storing data and a computer program; a processing unit coupled to the memory for executing a computer program. The computer program comprises: computer program code for receiving the modulated signal comprising an upper sideband component, a lower sideband component, and a main component; computer program code for determining signal-to-noise-ratios of the upper sideband component, the lower sideband component, and the main component of the modulated signal received by the receiver; computer program code for determining errors in carrier phases of the upper sideband component, the lower sideband component, and the main component of the modulated signal; and computer program code for identifying the presence of multipath dependent upon the determined signal-to-noise ratios and the determined errors in the carrier phases.

In accordance with a further aspect of the invention, there is provided a ranging system, comprising: an antenna; a receiver coupled to the antenna; an apparatus for processing in the receiver a signal generated using a complex modulation technique involving a spreading code and a subcarrier for the ranging system; and a position, velocity, and time solution module coupled to the receiver and the apparatus for receiving multipath mitigated code delay estimates τ _cm (t) for sources of the modulated signals tracked by the receiver. The apparatus comprises: a module for receiving the modulated signal comprising an upper sideband component, a lower sideband component, and a main component; a module for determining signal-to-noise-ratios of the upper sideband component, the lower sideband component, and the main component of the modulated signal received by the receiver; a module for determining errors in carrier phases of the upper sideband component, the lower sideband component, and the main component of the modulated signal; and a module for identifying the presence of multipath dependent upon the determined signal-to-noise ratios and the determined errors in the carrier phases.

In accordance with a still further aspect of the invention, there is provided a computer program product comprising a computer readable medium having recorded thereon a computer program executable by a processing unit for processing in a receiver a signal generated using a complex modulation technique involving a spreading code and a subcarrier for a ranging system. The computer program comprises: a computer program code module for receiving the modulated signal comprising an upper sideband component, a lower sideband component, and a main component; a computer program code module for determining signal-to-noise-ratios of the upper sideband component, the lower sideband component, and the main component of the modulated signal received by the receiver; a computer program code module for determining errors in carrier phases of the upper sideband component, the lower sideband component, and the main component of the modulated signal; and a computer program code module for identifying the presence of multipath dependent upon the determined signal-to-noise ratios and the determined errors in the carrier phases. hi each of the foregoing aspects, code phase error in the receiver due to the identified multipath may be mitigated.

The signal generated using a complex modulation technique may be an Alternate Binary Offset Carrier (AItBOC) modulated signal and the ranging system may be a Global Navigation Satellite System (GNSS). The upper sideband component, the lower sideband component, and the main component may comprise Galileo E5b, E5a, and E5 AItBOC modulated signals, respectively.

BRIEF DESCRIPTION OF DRAWINGS

Embodiments of the invention are described hereinafter with reference to the drawings, in which: Fig. l is a block diagram of a generalized, standard tracking loop architecture for

Galileo E5 signal;

Fig. 2 is a plot providing a code multipath error comparison for El BOC(I ₅ I) (4 MHz), El BOC (1,1) (32 MHz), E5a (20 MHz), and E5 (51 MHz);

Fig. 3 is a plot showing carrier phase multipath envelope for E5a (20 MHz) and E5 (51 MHz);

Fig. 4 comprises plots of minimum and maximum attenuation for E5a, E5b, and E5;

Fig. 5 comprises plots of composite phases for different multipath delays for E5a, E5b, and E5; Fig. 6 comprises plots of code phase errors due to multipath for E5a, E5b, and

E5;

Fig. 7 comprises plots of differences in composite carrier phases for E5- E5a, E5b -E5, and E5b- E5a;

Fig. 8 comprises plots of differences of composite carrier phase and code multipath error for of K ₂ (φ _c . φ _ca )-τ _c , τ _bc = K ₂ (φ _cb - φ _c )-? _c and Ki(φ _cb . φ _ca )-τ _c ; Fig. 9 comprises plots of multipath affected SNR for E5a, E5b, and E5; Fig. 10 comprises plots of differences in SNRs of received signals E5- E5a, E5b - E5, and E5b- E5a;

Fig. 11 is a block diagram of an architecture for implementing code phase multipath mitigation in an Galileo E5 AItBOC receivers in accordance with an embodiment of the invention;

Fig. 12 is a block diagram of a combiner block 1200 having inputs of composite carrier phases φ _c (t), φ _ca (i) _^ ^an & Φ _cb (t) _> *h ^e code delay τ{t) , the signal strength estimates b _c (t), b _cα (t), and b _cb (t);

Fig. 13 is a flow diagram illustrating a method of processing in a receiver a signal generated using a complex modulation technique involving a spreading code and a subcarrier for a ranging system in accordance with another embodiment of the invention; and

Fig. 14 is a block diagram of a ranging system comprising a receiver in accordance with an embodiment of the invention and a PVT solution module.

DETAILED DESCRIPTION

Methods, apparatuses, receivers, ranging systems and computer program products are disclosed for processing in a receiver a signal generated using a complex modulation technique involving a spreading code and a subcarrier for a ranging system. In the following description, numerous specific details, including particular lowpass filters, and the like are set forth. However, from this disclosure, it will be apparent to those skilled in the art that modifications and/or substitutions may be made without departing from the scope and spirit of the invention, hi other circumstances, specific details may be omitted so as not to obscure the invention. Where reference is made in any one or more of the accompanying drawings/figures to steps and/or features, which have the same reference numerals, those steps and/or features have for the purposes of this description the same function(s) or operation(s), unless the contrary intention appears.

In heavy multipath scenarios like urban canyons, the user code-phase pseudorange error due to multipath can be as high as two meters even with the most sophisticated signals like Galileo E5 AItBOC (15, 10). In the embodiments of the invention, methods are provided to mitigate the code phase multipath by exploiting the frequency diversity inherent to the AItBOC modulation used in Galileo E5 satellite navigation signals. The method is called Sideband Carrier Phase Combination (SCPC). Using the SCPC method, the instantaneous code phase multipath error can be reduced to less than half a meter. The method involves identifying the presence of multipath, estimating the magnitude of the multipath, and mitigating the multipath. The method estimates the code phase error due to multipath by using the carrier phase of the E5 signal (main component of AItBOC modulated signal) and the carrier phases of the E5a and E5b components (lower and upper sideband components) of the E5 signal. The carrier phase error is a measure of the phase delay of the received signal and code phase error is a measure of the group delay of the received signal. Therefore, from another perspective, the method estimates the group delay due to multipath by using the phase delay of the E5 signal (the main component of AItBOC modulated signal) and the phase delays of the E5a and E5b components (lower and upper sideband components) of the E5 signal. The apparatus described hereinafter implements the method of estimating the code phase multipath and correcting the code phase for multipath error, in a Galileo E5 receiver.

The SCPC method disclosed herein exploits the Frequency Diversity feature of the Galileo E5 AItBOC modulated signal. In wireless communication systems other than the ranging systems, frequency diversity has been used to address the problem of multipath but in a different context, hi these systems, transmitting and receiving multiple frequencies effectively carrying the same information are used to combine the energies in multipath channels via some special techniques like Maximal Ratio Combining (MRC) and Equal Gain Combining (EGC). The main aim in these techniques is to increase the channel capacity by reducing the Liter Symbol Interference (ISI). However the focus of ranging systems is estimating the code delay and carrier phase of the direct signal, excluding all of the superimposed multipath components at the receiving antenna.

Broadly speaking, the embodiments of the invention relate to a method of processing in a receiver a signal generated using a complex modulation technique involving a spreading code and a subcarrier for a ranging system 1300 as shown in Fig. 13. In step 1310, the modulated signal comprising an upper sideband component, a lower sideband component, and a main component is received. In step 1312, signal-to- noise-ratios of the upper sideband component, the lower sideband component, and the main component of the modulated signal received by the receiver are determined. In step 1314, errors in carrier phases of the upper sideband component, the lower sideband component, and the main component of the modulated signal are determined. In step 1316, the presence of multipath is identified dependent upon the determined signal-to- noise ratios and the determined errors in the carrier phases.

The method may further comprise the step of mitigating code phase error in the receiver due to the identified multipath. The mitigating step may comprise combining errors in the carrier phases of the upper sideband component, the lower sideband component, and the main component of the modulated signal. The combining step may comprise: determining differences in carrier phases of the upper and lower sideband components referenced to the main component of the modulated signal; and scaling the determined differences in carrier phases. The mitigating step may utilize frequency diversity in the modulated signal. The method may further comprise independently processing the upper sideband component, the lower sideband component, and the main component of the modulated signal. The method may further comprise utilizing different shapes of correlation waveforms that are obtained by the independently processing step.

The method may further comprise the step of measuring code phase and carrier phase from the upper sideband component, the lower sideband component, and the main component of the modulated signal using a single numerically controlled oscillator. The method may further comprise the step of obtaining a multipath mitigated code-phase measurement.

Preferably, the signal generated using a complex modulation technique is an Alternate Binary Offset Carrier (AItBOC) modulated signal and the ranging system is a Global Navigation Satellite System (GNSS). The upper sideband component, the lower sideband component, and the main component comprise Galileo E5b, E5a, and E5 AItBOC modulated signals, respectively.

These and other details are described in greater detail hereinafter. Galileo E5 Signal

The Galileo E5 signal encompasses a number of features. The two sideband components of the Galileo E5 signal can be considered as carrying the same ranging information. The two sidebands when considered separately can be treated as PSK-R(IO) and therefore differ in their modulation type to that of the main signal, which can be treated as 8-PSK AItBOC (this feature can be argued as modulation diversity). Since the effect of multipath is frequency selective and also depends on the type of modulation, the embodiments of the invention exploit these features in reducing the multipath effect. The received E5 AItBOC signal of any one satellite in the presence of a direct signal and N reflected components can be represented as: where ^s (0 ^{= s} _c (0 + J ^s _s (0 is the complex baseband signal, O ⁾ _[F and O ⁾ _d are the

carrier and Doppler frequencies respectively, Q is the phase, ^ is the delay and a, is the attenuation of the ι ^h signal, P is the received signal power, n(t) is the additive white Gaussian noise and Ji is the real argument function. For most applications, the difference in Doppler between the direct signal and the reflected signals can be considered to be negligible (especially for delays less than one chip of the E5 spreading code). The attenuation, time delay and phase of each of the reflected signals can be written relative to that of the direct component as:

Letting ^o ^~~ ®IF ^~ * ^~ ®d , Equation (1) can be written as: Signal tracking in Galileo E5 receivers can be achieved in several ways, including: wide-band signal tracking, and side-band tracking. The wide-band signal tracking involves a received signal being passed through a wideband filter (at least 51 MHz bandwidth so as to pass the first two side lobes) centered around 1191.795 MHz. This type of tracking allows full utilization of the shape of the AItBOC correlation function and the received power. This method is also referred to as 8-PSK AhBOC tracking. In side-band tracking, the E5a and E5b sidebands are extracted from the received signal by multiplying the received signal with an appropriate complex sub- carrier (equivalent to sideband translation). Since this operation results in a PSK-R(IO)- like correlation function, this operation is also referred to as the PSK-R(IO) tracking.

Fig. 1 shows a generalized, standard tracking loop architecture 100 for the Galileo E5 signal. Without loss of generality, analog representation of the signals is used at different points in the tracking loop up until an integrate-and-dump stage. Also an infinite bandwidth is assumed for the received signal rjp(t) unless otherwise specified. All lines with x, s andy as the label carry complex signals.

In the architecture 100 of Fig. 1, the received E5 AItBOC(15, 10) modulated signal ri _F (t) is mixed by a carrier mixer 102 with the output of a complex carrier module

104 (the locally generated complex carrier is ^x "' ^{~ e} ). The output y(t) of the mixer 102 is provided as input to three mixers 106, 114, and 120. The code mixer 106 mixes y(t) with the local replica of the 8-PSK AItBOC (code with subcarrier is

* i ^Λ \

¹ ^ ' ) of a reference baseband signal generator 130 to produce yi(t). Three versions of the local replica are generated: Early (E), Prompt (P), and Late (L). The mixer 106 produces six different correlation values corresponding to IE, EP, IL, QE, QP, and QL. The complex signal y _t (t) is provided to a lowpass filter comprising an integrator 108 and a sample and hold device 110 that samples at period T ₁ , to provide an integrate and dump circuit, producing y _λm as input to a code discriminator 112. The

* ( "Λ mixer 114 mixes y(t) with the output ^ ² ^ ' of the reference baseband signal generator 130 to produce y2(t). The output yj(t) is lowpass filtered, being provided to an integrator 116 coupled to a sample and hold device 118 that samples also at duration T ₁ , to provide an integrate and dump circuit, producing y _2m as another input to the code discriminator 112. The typical integration duration is one symbol period of the E5 signal, which is four milliseconds. However, the algorithm presented herein does not depend on the integration duration, or on the knowledge of the secondary code that might be present with the primary spreading code, and hence the algorithm is equally valid for other integration durations.

The code discriminator 112 outputs the code phase error to a code loop filter 126, the output of which is coupled to a code numerically controlled oscillator (NCO) 128. The code phase error is converted to code frequency updates. The output of the code NCO 128 is provided to the reference baseband signal generator 130. The code NCO 128 controls the timing offset of the reference base band signal generator 130, which is a complex signal generator comprising the sub-carrier and the spreading code. This is unlike the case of PSK-R(IO), where the reference signal generator only generates the spreading code. s ' (t - τ) Referring again to the mixer 120, the output ⁰ ^ ' of the reference baseband signal generator 130 is input to the mixer 120, which produces the output yo(t) provided to the integrator 122. The integrator 122 is coupled to the sample and hold device 124 that samples at duration T ₂ to perform an integrate-and-dump function producing output y _Ql provided to a carrier discriminator 132. The carrier lock loop discriminator 132 obtains the phase error between the incoming signal and the locally generated carrier. The output of the carrier discriminator 132 is provided to a carrier loop filter 134. The output of the carrier loop filter 134 is provided to a carrier NCO 136. This carrier phase error is filtered by the carrier lock loop filter 134 and converted to an appropriate frequency generated by the carrier NCO 136 for the complex carrier module 104. The updated frequency is mixed by mixer 102 with the input signal rj _f (t). The code phase output is obtained from the code loop filter 126 and the carrier phase output is obtained from the carrier loop filter 134.

The received signal r _/f (t) is multiplied by the mixer 102 with the complex carrier generated at the estimated intermediate frequency plus Doppler ( ^ ₀ ) and the estimated phase ( Φ ) to obtain >v) . The output yif) of the carrier mixer 102 is then multiplied by the reference baseband signal. The reference baseband signal comprises both the code and the sub-carrier, hi its simplest form, the reference signals from the generator 130 are early, late and prompt versions of the 8-PSK AItBOC signal with a chip spacing of 2δ between early and late samples:

_Sl ^' (t-τ) = s ^' (t-τ + δT _e ) _(4a)

s ₂ ^' (t-τ) = s'(t-τ-δT _c ) _m

s ₀ (t-τ) = s (t-τ) _(4c)

where i is the estimate of the code delay and ^ ₀ is the chip duration. Since the signal comprises pilot components that can be used for efficient carrier tracking, the prompt arm can carry the combined E5a and E5b pilot channels:

_* ( _t -i\- J f ^c «e('- ^f )-* ^c ('- ^f ) ⁺ )

(5)

where sc (t) = sc _s (t) + j ^■ sc _s (t - T _s /4) , SC ₅ [ _s the sum-sub-carrier, ^■ * ^■ _s is the sub-carrier period and ^c _a Q and c _bQ are the E5a and E5b quadrature spreading codes (including secondary codes), respectively.

In the case of PSK-R(IO) tracking, say with the E5a component, assuming that both the pilot and data channels are used for code tracking and only the pilot signal is used for carrier tracking, the reference signals are:

s;(t-τ) = ^c;{t-τ ₊ δT _c ).sc(t-τ + δT _c ) _(6a)

s ₂ {t-τ) = ^c _a ^* (t-τ-δT _c )- _S c(t-τ-δT _c ) _(6b)

so ( ^t -^) = ^-c _aQ (t-τ)-sc(t-τ) _(6c) where ^c _a ^{~ c} _a i ^~ ^ J ^c aQ is the E5a code (including secondary codes). A similar approach can be used for the PSK-R(IO) tracking with the E5b component.

The outputs obtained by multiplying with the different versions of the reference signal are integrated over a specified duration T ₁ seconds to use in the code tracking loop and T ₂ seconds to use in the carrier tracking. The code discriminator in this architecture uses the time shifted versions of the reference signal (e.g. early minus late) and the carrier tracking uses the prompt reference signal (e.g. arctan). The functionality of different blocks of the code and carrier tracking loops is similar to the ones used for other spread spectrum ranging signals.

Code tracking error with noise in the absence of multipath

Due to the presence of complex signals and the special sub-carrier waveforms, the derivation of the code tracking jitter is quite involved. For the non-coherent early minus late power code discriminator, Equation (7) provides the final result. σ _ε ² denotes the error variance in chips:

^ ₀ is the one-sided noise spectral density, B _L is the one-sided close loop noise bandwidth of the code lock loop, and the reference signals for this case are as given in Equations (4a,4b):

*(0 = * _c O + A ( ^• ) = *(') * */( ^• ), ^ = 1,2 ₍ 8) represents the correlation function between the input signal and the reference signal,

Ki-) = K (0 + JK (0 = *χ (0 x S _x ^* (0, * = 1 ,2 (9) represents the auto-correlation function of the reference signal, is the slope of the S-curve with the discriminator function given by

Code tracking error in the presence of multipath, without noise

Again due to the reason that the equations get more involved with the complex signals in place and also in accordance with the customary method used, only a single reflected signal and without noise is considered to derive an equation for the discriminator function and hence to analyze the effect of multipath on the code tracking error. The discriminator function in the presence of a single reflected signal for the early-minus-late-power (EMLP) discriminator is given by:

Interestingly, this equation is similar to the ones derived for the GPS Ll C/A

case, except for the use of complex correlation functions. The function ¹ ^ in the last term can be thought of as a 'correlation coefficient of the correlation functions' . The

JV function -** ^■ represents the correlation between the correlation of the direct signal with the reference signal and the correlation of reflected signal with the reference signal. Fig 2 shows a code multipath error comparison involving the multipath error envelope of the 8-PSK AItBOC and PSK-R(IO) tracking architectures of the E5 signal. The plot illustrates carrier phase error in the presence of multipath and code phase multipath error, without noise. The concepts developed for carrier phase error for the other signals hold good for the E5 AItBOC tracking architectures. This is because as far as the carrier is concerned, there is no change in the signal structure except the carriers are at different frequencies. The carrier phase multipath error for the AItBOC modulated signal comprising a direct signal component and a single reflected signal component can be derived and is given by:

where ™ ^c is the phase error and again ^■ ** ^■ is the complex autocorrelation function between the input signal and the reference signal.

Fig. 3 shows the carrier phase multipath error envelope for E5a and E5 and provides a carrier phase multipath error comparison. Unlike the PSK-R tracking, both the code and carrier phase errors have nulls within a single chip. This very nature quite similar to any BOC(m,n) signal is due to the phenomenon of negative correlation between frequency components of the signal. The presence of subcarrier alters the shape of the correlation function of the spreading code, which is the major influencing factor for the shape of the multipath error envelope.

Signal-to-Noise-Ratio (SNR) in the presence of multipath If the composite signal received at the IF (single reflector case) is expressed as:

where the subscript c denotes the 'composite' signal, then the strength of the composite signal at the output of the correlator can be derived equivalently as: b _c = + r, ) costø )

If the correlator output strength for the direct signal is assumed to be ° ^~ l ' , then the equivalent attenuation of the composite signal with respect to the direct signal from (2) is:

β _c = -^ = l + α,A +2Ja^ ₁ costø)

(16) where ^ ¹ is the attenuation of the reflected signal with respect to the direct signal at the correlator output.

Fig. 4 illustrates the minimum and maximum attenuation in the case of E5a, E5b and E5 signals. Again, the shape of the attenuation for E5, E5a and E5b is due to the shape of the respective correlation function.

Sideband carrier phase combination method and architecture E5a and E5b Carrier Phases under Multipath λ λ

The E5a, E5 and E5b signals have wavelengths of ^E5a = 25.48 cm, ^E5 = 25.15 cm, and λ ^E5b =24.83 cm, respectively, due to their corresponding carrier frequencies. When a signal travels from a source (satellite) to a destination (receiver), certain cycles and a certain fraction of a cycle elapse. This fraction of a cycle converted

to radians is the phase of the received signal, i.e., ^ . In the absence of reflected signals, the phase of the received signal depends on the distance between the satellite and the receiver plus any associated errors in estimating the pseudorange. The phases of

_ 2πd 2nd the received signals in this case for E5, E5a and E5b are denoted as ^{~ ~~} λ — ^£5 , ^~ i — ^£5 ° , ft - ^lπd and ^Eih , respectively. In the presence of multipath, there can be any number of reflected signals. For ease of description, a single reflection case is used herein. The method and apparatus described herein equally apply to the case where there is more than one reflected signal combined at the receiver antenna. The reflected signal always arrives after the direct signal. This means that the reflected signal always travels a longer

distance than the direct signal. The phases of the reflected signals are then ^E5 , . The multipath delay which is the difference in time , _ ⁽ <W ⁾ between the reflected signal and direct signal is ^c , where C is the velocity of light. hi the case of E5a and E5b processing, the composite phases are given by: ) (17a)

where *» ; is the autocorrelation function of the PSK-R(IO) baseband signal and

υ is the corresponding code estimate error.

The time estimates for the E5a and E5b code lock loops are provided by the E5

code tracking loop, which makes ε ^{a =} ε ^{b =} ε . In addition, the reflector can be considered to be frequency independent within the 30 MHz band around 1191.795 MHz. au _ — «i* _ — «10 _ — ₍ _J ₁

Hence, " ^Oα " ⁰⁶ " ⁰⁰ . Therefore the composite carrier phases for the E5a and E5b components of the signal can be expressed as:

Fig. 5 illustrates the composite carrier phases plotted for E5, E5a, and E5b (top, middle, and bottom plots) for different multipath delays.

Fig. 6 shows the code phase error for E5a, E5b, and E5 (top, middle and bottom plots) due to multipath for different multipath delays. In these simulations, without loss of generality, the phase of the reflected signal is assumed to be only due to the multipath delay. If the composite carrier phases are each subtracted one from another, then the obtained results are as shown in Fig.7, which shows the differences in composite carrier phases for E5-E5a, E5b-E5, and E5b-E5a (top, middle, and bottom plots). As shown in Fig. 7, the shapes of the differences in composite carrier phases resemble the shape of the code phase multipath error of the E5 signal (see Fig. 6, top plot). If the code phase error is again subtracted with this difference in carrier phases, then the difference of composite carrier phase and code multipath error is obtained as shown in Fig 8 (τ _c is the code multipath error for E5). The subtraction is formulated as follows:

(19)

Fig. 8 shows plots of resulting values T _ca ,τ _bc and τ _ba . The top, middle and bottom plots in Fig. 8 are of K2{φ _c - φ _ca )~^ _c , t _bc = K2(φ _cb . φ _c )— T _c , and Kj(φ _cb -φ _ca )—T _c , respectively. The code phase multipath errors are reduced to a great extent, if these are used for the measurements. The optimum values of the constants ^ ¹ and ² depend on the receiver bandwidth. In addition, ^{2 ~} ' . Typically, the value of 1 is found to be around 2.0.

E5a and E5b Signal to Noise Ratios under Multipath

The received signal consisting of a single reflected signal as described above is considered. In multipath, a receiver processing the signal affected by multipath is known to estimate the signal to noise ratio as the composite signal to noise ratio instead of the signal to noise ratio of the direct signal. In the case of E5, E5a and E5b processing, this signal to noise ratio is given by:

b _ca = R _P ² {ε) + a _x R _P ² {ε + τ _x ) + 2^R _P {ε)R _p (ε + τ _x )cos(φ _Xa ) b _Λ = R _P ¹ (ε) + a _x R _p ¹ (ε + τ _{ ) + 2 _y [a _x R _P (ε)R _P (ε + τ _x )cas{φ _{b ) Equations (20) show that the composite signal strengths differ among the three signals, and the difference is only related to the phase of the multipath signal. Fig. 9 shows plots of the composite SNRs for different multipath delays. The top, middle and bottom plots of Fig. 9 show the multipath affected SNR for E5, E5a, and E5b, respectively.

If the composite SNRs are each subtracted from another, then the result as shown in Fig.10 is obtained. The top, middle, and bottom plots of Fig. 10 show the SNR difference for E5-E5a, E5b-E5, and E5b-E5a, respectively. The shapes of the differences in composite SNRs resemble the shape of the code phase multipath error of the E5 signal (see Fig. 6, bottom plot). This can be used in conjunction along with the composite phase difference method. As the SNR measurement is less robust than the phase measurement, the SNR method can be used to verify the detection of multipath. The difference equation is formulated as follows. 1 if iK _h -b _ca ) > η

™ab = 1 0 otherwise

1 if (b _cb - b _c ) > η mbc =

0 otherwise

1 if Φ _c -K) > ri mca = 0 otherwise

(21) where ^xy are the multipath indication flags and ' is the threshold optimized to trigger the difference. Depending on the SNR estimation method used and the tracking loop parameters, ' is chosen. Typical values of ^ range from 0.5-1.0 (in dB).

Architecture for the SCPC method Fig. 11 illustrates an architecture 1100 for implementing the SCPC method. This is a special case of the generalized architecture 100 shown in Fig. 1. The received E5 AltBOC(15, 10) modulated signal rιp(t) is mixed by the carrier mixer 1102 with the

locally generated complex carrier of the complex carrier module 1104 ( ^{w ~ e} ). The output y(t) of the carrier mixer 1102 is provided as input to three code mixers 1106, 1114, 1120 and an E5a/b band translator 1140. The E5a/b band translator 1140 is

described hereinafter. The mixer 1106 mixes y(t) with the output ( ^ES ^ ^c ^ ) of an AItBOC reference signal generator 1130 to produce yi(t). The AItBOC generator 130 produces E, P, and L versions of the replica. The complex signal yι(t) is lowpass filtered using an integrator 1108 and sample and hold device 1110 that samples at duration T ₁ , to provide an integrate and dump circuit, producing y _λm as input to a code discriminator

1112. The mixer 1114 mixes y(t) with the output ^s ^ K ^{~ τ ~} J _o f the AItBOC reference signal generator 1130 to produce yiit). The output y2(t) is provided to an integrator 1116 coupled to a sample and hold device 118 that samples also at duration T ₁ , to provide an integrate and dump function, producing y _2m as another input to the code discriminator 1112. The typical integration duration is one symbol period of the E5 signal, which is four milliseconds. However, the algorithm herein does not depend on the integration duration, or on the knowledge of the secondary code that might be present with the primary spreading code, and hence the algorithm is equally valid for other integration durations.

The code discriminator 1112 outputs the code phase error to a code loop filter (E5) 1126, the output of which is coupled to a code numerically controlled oscillator (NCO) 1128. The code phase error is converted to code frequency updates. The code loop filter 1126 also provides a code delay τ _c (t) as output (not shown in Fig. 11). The output of the code NCO 1128 is provided to the AItBOC reference signal generator 1130 and an E5a and E5b code generator 1154. The code NCO 1128 controls the timing offset of the AItBOC reference signal generator 1130 and the E5a and E5b code generator 1154. Referring again to the mixer 1120, the output ^Sεs ^ ^{~ τ} > of the AItBOC reference signal generator is input to the mixer 1120, which produces the output yo(t) provided to the integrator 1122. The integrator 1122 is coupled to the sample and hold device 1124 that samples at duration T ₂ to perform an integrate-and-dump function producing output y _QI provided to a carrier discriminator 1132. The carrier lock loop discriminator 132 obtains the phase error between the incoming signal and the locally generated carrier. The output of the carrier discriminator 1132 is provided to a carrier loop filter 1134. The carrier phase error is filtered by the carrier lock loop filter 134 and converted to an appropriate frequency generated by the carrier NCO 1136 for the complex carrier module 1104. The carrier loop filter (E5) 1134 also provides as output the composite carrier phase φ _c (t) for E5 (not shown in Fig. 11). Again, the updated frequency is mixed by the carrier mixer 1102 with the input signal. The code phase output is obtained form the code loop filter 1126 and the composite carrier phase output is obtained from the carrier loop filter 1134. The output y _m is used to generate b _c (t), the signal strength estimate of the E5a signal. A method of generating b _c (t) is to compute the magnitude of the complex value y _ol after each sample and hold instant. As the noise in such single sample measurements is generally high, b _c (t) can be obtained by averaging the magnitudes of Jp ₀ , for a few hundreds of milliseconds. Typical averaging durations are 500 ms to 1000 ms. Since the E5a and E5b components are centered on a frequency equivalent to the difference of sub-carrier frequency (from the center of the E5 signal), the translator 1140 frequency translates the E5a and E5b signals to the baseband, using the subcarriers corresponding to E5a and E5b. The E5a/b band translator 1140 provides outputs to two code mixers 1142, 1148. The E5a and E5b code generator 1154 generates the local replica of the E5a signal (E5al and/or E5aQ) and the local replica of the E5b signal s ^* (t - τ) s ^* (t - τ) (E5bl and/or E5bQ). The outputs ^{E5a v} ' and ^{E5b v} > of the generator 1154 are provided to the code mixers 1142 (E5a) and 1148 (E5b), respectively.

The output yo _a (t) of the code mixer 1142 is lowpass filtered using an integrator 1144 coupled to a sample and hold device 1146 that samples at duration T ₂ performing an integrate-and-dump function to produce output y _Oa . The output y _Oa is provided to carrier discriminator (E5a) 1162, which in turn is coupled to a carrier loop filter (E5a) 1164. The carrier lock loop discriminator (E5a) 1162 obtains the phase error between the incoming signal and the locally generated carrier. This error is filtered by the carrier lock loop filter 1164. The carrier loop filter (E5a) 1164 also provides as output the composite carrier phase φ _ca (0 for E5a (not shown in Fig. 11). The output of the carrier loop filter (E5a) 1164 is coupled to E5a carrier NCO 1166, which is coupled to the E5a/b band translator 1140. In this manner, an updated frequency is provided to translator to translate the E5a signal to the baseband. The output y _Oa is used to generate b _ca (t), the signal strength estimate of the E5a signal. A method of generating b _ca (t) is to compute the magnitude of the complex value y _Oa after each sample and hold instant. As the noise in such single sample measurements is generally high, b _ca (t) can be obtained by averaging the magnitudes of y _Oa f° ^{r a} f ^ew hundreds of milliseconds. Typical averaging durations are 500 ms to 1000 ms.

The output yo _b (t) of the code mixer 1148 is lowpass filtered using an integrator 1150 coupled to a sample and hold device 1152 that also samples at duration T ₂ performing an integrate-and-dump function to produce output y _ob . The output y _ob is provided to carrier discriminator (E5b) 1156, which in turn is coupled to a carrier loop filter (E5b) 1158. The carrier lock loop discriminator (E5b) 1156 obtains the phase error between the incoming signal and the locally generated carrier. This error is filtered by the carrier lock loop filter (E5b) 1158. The carrier loop filter (E5b) 1158 also provides as output the composite carrier phase φ _cb (t) for E5b (not shown in Fig. 11). The output of the carrier loop filter (E5b) 1158 is coupled to E5b carrier NCO 1160, which is coupled to the E5a/b band translator 1140. In this manner, an updated frequency is provided to translator to translate the E5b signal to the baseband. The output y _Ob is used to generate b _ct ,(t), the signal strength estimate of the E5b signal. A method of generating b _cb (t) is to compute the magnitude of the complex value y _ob after each sample and hold instant. As the noise in such single sample measurements is generally high, b _cb (t) can be obtained by averaging the magnitudes of y _ob for a few hundreds of milliseconds. Typical averaging durations are 500 ms to 1000 ms. The code delay estimate from the output of the code numerically controlled oscillator (NCO) 1128 used for the E5 tracking is given as input to the E5a and E5b code generators 1154. There are no additional code generators required for the E5a and E5b signals, because the timing for both the 8-PSK AItBOC reference signal and the E5a/b code generator 1154 is the same. In fact, the outputs of the E5a/b code generator 1154 shown explicitly in this diagram are already available in the AItBOC reference signal generator 1130 and hence additional hardware is not required except for the code mixers 1142, 1148 and loop modules.

Combiner Block Fig. 12 shows a combiner block 1200, which receives as inputs: the composite carrier phases φ _c (t), φ _ca (t), and φ _cb (t), the code delay τ{t) , and the signal strength

1C fC estimates b _c (t), b _ca (t), and b _Cb (t). Also, the constants ' and ² are input to combiner block 1200. The error in the code delay estimate 7 ^" without multipath is & and in the presence of multipath, the error is ^τ _c . Figs. 7 and 8 show that the combination of carrier phases according to Equation (19) represents the code multipath error. Therefore the combiner block 1200 subtracts the scaled differences of the carrier phases from the code delay estimate T to obtain the multipath mitigated code delay estimate t _cm . In addition, the signal strength estimates b _c , b _ca and b _cb , are used in accordance with equation (21) to cross verify the formulation of the τ _cm . The criteria for the passing the verification is that at least one multipath indicator flag should have been set to 1 when evaluating Equation (21). The block 1200 combines according to the phase difference formulation described hereinbefore with reference to Equation (19) and Figs. 7 and 8. An application of the SCPC method using the multipath mitigated code delay estimate is shown in the system 1400 of Fig 14. The antenna 1410 receives the signal r _/f (t), which is then fed into the receiver 1420. The receiver 1420 houses the Radio Frequency down conversion and the base band signal processing operations. In this example, the receiver 1420 is implemented using the architecture 1100 of Fig. 11 and the combiner block 1200 of Fig. 12. The Position Velocity and Time solution module 1430 is responsible for the computation of user position, user velocity, and the time information using the code delay estimates 1422 of the satellites tracked by the receiver 1420. The signal 1422 carries the multipath mitigated code delay estimates τ _cm (t) for all the satellites that are tracked in the receiver 1420. The signal 1424 is the control and other parameters (including the carrier phase estimates) between the receiver 1420 and the PVT solution module 1430. In this manner, the receiver 1420 is adapted to implement processing in the receiver of a signal generated using a complex modulation technique involving a spreading code and a subcarrier for a ranging system. The presence of multipath can be identified dependent upon the determined signal-to-noise ratios and the determined errors in the carrier phases. The scaled difference of carrier phases can be used to mitigate the identified code phase multipath error in the code delay estimate of the receiver 1420. The code delay estimate of the receiver 1420 is subsequently used by the PVT solution module 1430 to obtain the multipath mitigated PVT solution.

Computer Implementation and Computer Program Product

The methods in accordance with the embodiments of the invention may be implemented using a general purpose computer system. The methods may be implemented as software, such as one or more application programs executable within the computer system. In particular, the steps of the method are effected by instructions in the software that are carried out within the computer system. The instructions may be formed as one or more computer program code modules, each for performing one or more particular tasks. The software may also be divided into two separate parts, in which a first part and the corresponding code modules performs the method and a second part and the corresponding code modules manage a user interface between the first part and the user. The software may be stored in a computer readable medium, including the storage devices described hereinafter, for example. The software is loaded into the computer system from the computer readable medium, and then executed by a processing unit of the computer system. A computer readable medium having such software or computer program recorded on the computer readable medium is a computer program product. The use of the computer program product in the computer system preferably effects an advantageous apparatus. The system may be implemented as an embedded computer system, e.g. where a small computing unit is part of a bigger hardware system.

The computer system comprises a computer module, input devices such as a keyboard, a mouse pointer device, and output devices including a display device. An external Modulator-Demodulator (Modem) transceiver device may be used by the computer module for communicating to and from a communications network. The network may be a wide-area network (WAN), such as the Internet or a private WAN, for example. The computer may be connected to the network using a high capacity (e.g., cable) connection, and the modem may be a broadband modem. A wireless modem may also be used for wireless connection to the network.

The computer module typically includes at least one processor unit and a memory unit for storing data and computer program code, for example formed from semiconductor random access memory (RAM) and read only memory (ROM). The computer module also includes a number of input/output (I/O) interfaces including an audio-video interface that couples to the video display and loudspeakers, an I/O interface for the keyboard and mouse and an interface for the external modem. The computer module may also have a local network interface that permits coupling of the computer system to a local computer network, known as a Local Area Network (LAN). The local network may also couple to the wide-area network via a connection, which would typically include a so-called "firewall" device or similar functionality. The interface may be formed by an Ethernet™ circuit card, a wireless Bluetooth™ or an IEEE 802.11 wireless arrangement.

Storage devices are provided and typically include a hard disk drive (HDD). Other devices such as a floppy disk drive and a magnetic tape drive (not illustrated) may also be used. An optical disk drive is typically provided to act as a non-volatile source of data. Portable memory devices, such optical disks (e.g., CD-ROM, DVD), USB- RAM, and floppy disks for example may then be used as appropriate sources of data to the system.

Examples of computers on which the described arrangements can be practised include IBM-PC's and compatibles, Sun Sparcstations, Apple Mac™ or alike computer systems evolved therefrom.

Typically, the application programs discussed above are resident on the hard disk drive and read and controlled in execution by the processor. Intermediate storage of such programs and any data fetched from the networks may be accomplished using the semiconductor memory, possibly in concert with the hard disk drive. In some instances, the application programs may be supplied to the user encoded on one or more CD-ROM and read via the corresponding drive, or alternatively may be read by the user from the networks. Still further, the software can also be loaded into the computer system from other tangible computer readable media. Computer readable media refers to any tangible storage medium that participates in providing instructions and/or data to the computer system for execution and/or processing. Examples of such media include floppy disks, magnetic tape, CD-ROM, a hard disk drive, a ROM or integrated circuit, a magneto-optical disk, or a computer readable card such as a PCMCIA card and the like, whether or not such devices are internal or external of the computer module. Examples of computer readable transmission media that may also participate in the provision of instructions and/or data include radio or infra-red transmission channels as well as a network connection to another computer or networked device, and the Internet or Intranets including e-mail transmissions and information recorded on Websites and the like.

The second part of the application programs and the corresponding computer program code modules mentioned above may be executed to implement one or more graphical user interfaces (GUIs) to be rendered or otherwise represented upon the display. Through manipulation of the keyboard and the mouse, a user of the computer system and the application may manipulate the interface to provide controlling commands and/or input to the applications associated with the GUI(s). The methods may also be implemented, at least in part, in dedicated hardware such as one or more integrated circuits performing the functions or sub functions to be described. Such dedicated hardware may include dedicated processors, digital signal processors, or one or more microprocessors and associated memories.

Methods, apparatuses, ranging systems, receivers and computer program products for processing in a receiver a signal generated using a complex modulation technique involving a spreading code and a subcarrier for a ranging system have been disclosed with reference to embodiments of the invention. The foregoing describes only some embodiments of the present invention, and modifications and/or changes can be made thereto without departing from the scope and spirit of the invention, the embodiments being illustrative and not restrictive.