BACKGROUND ART As well known, the diode rectifier has been most widely used as a utility interface converter because of its simplicity and its merit in price. However, the diode rectifier has the serious demerit that the power flow can not be reversed. The diode rectifier also has the demerits that it generates large reactive power and harmonics, and the dc-side current and the dc-side voltage can not be controlled.

There are many applications which need bidirectional power flow capabilities.

An important example is the voltage source inverter (VSI) motor drive system with regenerative braking to recover the kinetic energy of the motor and load. Another important example is the high voltage direct current (HVDC) system for power transmission.

The pulse width modulation (PWM) current source converter (CSC), for example, disclosed in E. P. Wiechmann, P. D. Ziogas, and V. R. Stefanovic,"A novel bilateral power conversion scheme for variable frequency static power supplies,"IEEE Trans. Ind. Applicat., vol. IA-21, pp. 1226-1233, Sept./Oct. 1985, T. Salzmann and A.

Weschta,"Progress in voltage source inverters (VSIs) and current source inverters (CSIs) <BR> <BR> <BR> with modem semiconductor devices,"in Conf Rec. IEEE-IAS Annu. Meeting, 1987, pp.

577-583, and B. K. Bose,"Power electronics-an emerging technology,"in Proc.

IECON'88, 1988, pp. 501-508 and the PWM voltage source converter (VSC), for example, disclosed in B. K. Bose,"Power electronics-an emerging technology,"in Proc.

IECON'88. 1988, pp. 501-508, T. Okuyama and H. Nagase,"Apparatus for controlling ac motor,"U. S. Patent No. 4, 328, 454, 1982, B. T. Ooi, J. C. Salmon, J. W. Dixon, and A. B.

Kulkarni,"A three-phase controlled-current PWM converter with leading power factor," <BR> <BR> <BR> IEEE Trans. Ind. Applicat., vol. IA-23, pp. 78-84, Jan./Feb. 1987, M. Nishimoto, J. W.

Dixon, A. B. Kulkarni, and B. T. Ooi,"An integrated controlled-current PWM rectifier <BR> <BR> <BR> chopper link for sliding mode position control,"IEEE Trans. Ind. Applicat., vol. IA-23,

pp. 894-900, Sept./Oct. 1987, S. B. Dewan and R. Wu,"A microprocessor-based dual <BR> <BR> <BR> <BR> PWM converter fed four quadrant ac drive system,"in Proc. Iras87, 1987, pp. 755-759, B.

T. Ooi and X. Wang,"Voltage angle lock loop control of the boost type PWM converter for HVDC application,"IEEE Trans. Power Electron., vol. 5, pp. 229-235, Apr. 1990, and B. T. Ooi,"Pulse width modulation power transmission system,"U. S. Patent No.

4, 941, 079, 1990 are converters with bidirectional power flow capabilities. The PWM CSC provides a bipolar dc-side voltage capability, so that it is used in current source inverter (CSI) systems in which the power flow is reversed by changing the dc-side voltage polarity, and the PWM VSC provides a bidirectional dc-side current capability, so that it is used in VSI systems in which the power flow is reversed by changing the dc-side current direction. These PWM converters have the additional merits that the reactive power and harmonics problems can be minimized, and the dc-side current and the dc-side voltage can be controlled. However, they are expensive and lack power because self- commutated switching devices, such as insulated gate bipolar transistors (IGBT's) and gate-turn-off thyristors (GTO's), should be employed.

In high power applications, the conventional line-commutated thyristor phase- controlled CSC, for example, disclosed in B. K. Bose,"Power electronics-an emerging technology,"in Proc. IECON'88, 1988, pp. 501-508, N. Mohan, T. M. Undeland, and W. <BR> <BR> <BR> <BR> <P>P. Robbins, Power Electronics. converters, applications, and design, 2nd ed., New York : Wiley, 1995, pp. 138-153, 418-428, 460-468, 494-500, and W. Leonhard,"Adjustable- speed ac drives,"Proc. of the IEEE, vol. 76, pp. 455-471, Apr. 1988. is used. Line- commutated thyristors provide high power ratings and the converter has a bipolar dc-side voltage capability, so that it is used in high power CSI systems. However, the converter generates large reactive power and harmonics and is not applicable to VSI systems. In VSI systems, the dual converter (two back-to-back connected thyristor phase-controlled CSC's) can be used, but with double the cost and with the equally large reactive power and harmonics, as shown in S. B. Dewan and R. Wu,"A microprocessor-based dual <BR> <BR> <BR> PWM converter fed four quadrant ac drive system,"in Proc. Iras87, 1987, pp. 755-759, and N. Mohan, T. M. Undeland, and W. P. Robbins, Power Electronics : converters, <BR> <BR> <BR> <BR> applications, and design, 2nd ed., New York : Wiley, 1995, pp. 138-153, 418-428, 460- 468, 494-500. There is still another type of converter including a diode rectifier and a thyristor converter, as shown in W. Leonhard,"Adjustable-speed ac drives,"Proc. of the IEEE, vol. 76, pp. 455-471, Apr. 1988, and T. Suzuki and B. Tech,"DC power-supply system with inverting substations for traction systems using regenerative brakes,"IEE

PROC., vol. 129. pt. B. pp. 18-26, Jan. 1982, but it requires a large step-up transformer and also generates large reactive power and harmonics, as shown in W. Leonhard, "Adjustable-speed ac drives,"Proc. of the IEEE, vol. 76, pp. 455-471, Apr. 1988, T.

Suzuki and B. Tech,"DC power-supply system with inverting substations for traction systems using regenerative brakes,"IEE PROC., vol. 129. pt. B. pp. 18-26, Jan. 1982, and S. Kawada and H. Ishida,"Operation control apparatus for ac motors,"U. S. Patent No.

4, 353, 023, 1982.

DISCLOSURE OF THE INVENTION It is therefore a general object of the present invention to provide a new utility interface converter employing line-commutated thyristors with a bidirectional power flow capability.

It is a more particular object of the present invention to provide a thyristor phase- controlled VSC with a bidirectional dc-side current capability.

It is another object of the present invention to provide a thyristor phase- controlled VSC in which the reactive power is smaller than the conventional thyristor phase-controlled CSC.

It is another object of the present invention to provide a thyristor phase- controlled VSC in which the ac-side harmonics is smaller than the conventional thyristor phase-controlled CSC.

It is another object of the present invention to provide a thyristor phase- controlled VSC in which the dc-side harmonics is smaller than the conventional thyristor phase-controlled CSC.

It is an additional object of the present invention to provide a thyristor phase- controlled VSC in which the average value of the dc-side current can be controlled from a negative minimum to a positive maximum value in a continuous manner.

It is another additional object of the present invention to provide a thyristor phase-controlled VSC in which the dc-side voltage can be regulated at its desired value.

According to one aspect of the present invention, a power circuit comprising a three-phase bridge, a three-phase inductor, and a capacitor is provided. The power circuit of the present invention has the same structure as that of the PWM VSC. However, the switching devices are not self-commutated devices, but line-commutated thyristors.

According to another aspect of the present invention, a control system comprising a gate firing controller and a firing angle controller is provided. Differing

from the PWM VSC, the converter of the present invention is operated not with the PWM but with the six-pulse phase-control.

BRIEF DESCRIPTION OF THE DRAWINGS The present invention relates to the above objects and features individually as well as collectively. These and other objects, features and advantages of the present invention will become apparent to those skilled in the art from the following detailed description of the preferred embodiments of the invention in conjunction with the accompanying drawings, in which : FIG. 1 is a power circuit diagram of the present invention ; FIG. 2 is blocks of a gate firing controller and a firing angle controller of the present invention ; FIG. 3 is waveforms for describing a function of the gate firing controller and an operation of the converter of the present invention in zero mode ; FIG. 4 is waveforms for describing a function of the gate firing controller and an operation of the converter of the present invention in rectification mode ; FIG. 5 is waveforms for describing a function of the gate firing controller and an operation of the converter of the present invention in inversion mode ; FIG. 6 is a block of a current feedback controller of the present invention ; FIG. 7 is a block of a voltage feedback controller of the present invention ; FIGS. 8A-8C show phasor diagrams for FIGS. 3-5, respectively ; FIG. 9 shows a region of possible operation of V,, on a supposition of purely sinusoidal i,, ; FIG. I OA shows an equivalent circuit of FIG. 1 for waveform analysis, and FIGS.

10B and 10C are circuits for applying the superposition principle ; FIG. 11 shows waveforms in the zero mode of operation ; FIG. 12 shows waveforms in the rectification mode of operation ; FIG. 13 shows waveforms in the inversion mode of operation ; FIG. 14 shows a range of possible operation of the firing angle 8 for a given value of Vd ; FIG. 15 shows a region of possible operation of V,,, ; FIG. 16 shows a dc-side current control characteristic for k = 0. 8 ; FIGS. 17A and 17B show loci of when the dc-side voltage is regulated ; FIGS. 18A and 18B show loci of the voltage VaS"-Va"across the inductor for k = 0. 8 and for k = 0. 85, respectively ;

FIGS. 19A and 19B show loci of the ac-side current zip for k = 0. 8 and for k = 0. 85, respectively ; FIG. 20 shows a reactive power characteristic ; FIG. 21 shows an ac-side harmonics characteristic ; FIG. 22 shows a dc-side harmonics characteristic ; FIG. 23 shows an experimental system ; FIG. 24 shows an experimental result for FIG. 3 ; FIG. 25 shows an experimental result for FIG. 4 ; and FIG. 26 shows an experimental result for FIG. 5.

MODES OF CARRYING OUT THE INVENTION Reference will now be made in detail to the present invention, examples of which are illustrated in the accompanying drawings. Wherever possible, the same reference numbers and symbols will be used throughout the drawings the refer to the same or like parts.

The present invention has two aspects, one is a power circuit and the other is a control system.

FIG. 1 shows the power circuit of the present invention. The power circuit of the present invention is connected between a three-phase ac network 11 and a dc network 12. The power circuit of the present invention comprises a three-phase bridge 13, a three-phase inductor 14, and a capacitor 15.

The three-phase bridge 13 has three ac terminals a, b, and c, a positive terminal d, a negative terminal e, and comprises three half-bridges. Each of the three half-bridges <BR> <BR> <BR> <BR> comprises a first thyristor Ql, Q3, or Q5, a second thyristor Q4, Q6, or Q2, a first diode Dl,<BR> <BR> <BR> <BR> <BR> <BR> D3, or D5, and a second diode D4, D6, or D2. The anode and the cathode of the first<BR> <BR> <BR> <BR> <BR> <BR> thyristor Ql, Q3, or Q, are respectively connected to the positive terminal d and each of the three ac terminals a, b, or c. The anode and the cathode of the second thyristor Q4, Q6, or Q2 are respectively connected to each of the three ac terminals a, b, or c, and the <BR> <BR> <BR> <BR> negative terminal e. The anode and the cathode of the first diode D,, D3, or D5 are respectively connected to each of the three ac terminals a, b, or c, and the positive terminal d. And, the anode and the cathode of the second diode D4, D6, or D2 are respectively connected to the negative terminal e and each of the three ac terminals a, b, or c.

The three-phase inductor 14 is connected between the three ac terminals a, b, and c of the three-phase bridge 13 and the terminals as, bs, and cs of the three-phase ac

network 11. In FIG. 1, the three-phase ac network 11 is expressed as a three-phase voltage source without a source impedance for convenience. However, the three-phase ac network 11 may have a source impedance in practice, then the three-phase inductor 14 is modeled to include the source impedance.

The capacitor 15 is connected between the positive terminal d and the negative terminal e. The positive terminal d and the negative terminal e are connected to the dc network 12. In FIG. 1, the capacitor 15 is divided into two parts, however this division is not necessary in practice. In FIG. 1, the capacitor 15 is expressed with two parts so as to show an assumed neutral point o, which is helpful for explaining the operation of the converter of the present invention.

FIG. 2 shows the control system of the present invention as blocks of a gate firing controller 21 and a firing angle controller 22.

The gate firing controller 21 is responsive to a firing angle 8, and applies the gate current pulses iGl, iG2, iG3, iG4, iG5, and iG6, to the thyristors Q,, Q2, Q3, Q4, Q5, and Q6 in the three-phase bridge 13.

FIGS. 3-5 show waveforms for describing a function of the gate firing controller 21. FIGS. 3-5 show only waveforms for phase a because waveforms for phase b and <BR> <BR> <BR> <BR> phase c are identical with that of phase a except that they are phase shifted by 2/3 n and 4/3 n, respectively. The gate firing controller 21 applies a first gate current pulse il to the first thyristor Q, in the half-bridge under a first condition that the current ia flowing into the ac terminal a is negative. And the gate firing controller 21 applies a second gate current pulse iG4 to the second thyristor Q4 in the half-bridge under a second condition that the current ia flowing into the ac terminal a is positive. The first gate current pulse !'c) is delayed or advanced by the firing angle 8 with respect to the positive zero crossing point <BR> <BR> <BR> <BR> of the phase voltage waveform v,,,,, at the terminal as of the three-phase ac network 11.

The second gate current pulse iG4 is delayed or advanced by the firing angle 8 with respect to the negative zero crossing point of the phase voltage waveform v,.,, at the terminal as of the three-phase ac network 11. Functions of the gate firing controller 21 for phase b and phase c are identical with that of phase a except that they are phase shifted by 2/3 Z and 4/3 n, respectively.

The firing angle controller 22 gives the firing angle 8 to the gate firing controller 21 so that the direction and the amount of the power flow of the converter can be controlled.

FIG. 6 shows a block of a current feedback controller 61 of the present invention.

The current feedback controller 61 is an embodiment of the firing angle controller 22.

The current feedback controller 61 is responsive to an externally applied current <BR> <BR> <BR> command signal Id*, and to a measured signal id'of the current id flowing out of the positive terminal d. The current feedback controller 61 generates the firing angle 8 as a function of the difference of the externally applied current command signal Id from the measured signal idnl.

FIG. 7 shows a block of a voltage feedback controller 71 of the present invention.

The voltage feedback controller 71 is another embodiment of the firing angle controller 22. The voltage feedback controller 71 is responsive to an externally applied voltage <BR> <BR> <BR> command signal F*, and to a measured signal vu'ouf the voltage #d across the capacitor 15.<BR> <BR> <BR> <BR> <BR> <BR> <P>The voltage feedback controller 71 generates the firing angle 8 as a function of the difference of the externally applied voltage command signal Vd from the measured signal #dm.

Principle of Operation FIGS. 3-5 show waveforms for explaining an operation of the converter of the present invention in zero mode, rectification mode, and inversion mode, respectively. In FIGS. 3-5, it is assumed that the ac-side source voltages are purely sinusoidal, the dc-side voltage vd is a purely dc voltage Vd with a large value of the dc-side capacitance, and the converter is operating under a balanced, steady-state condition.

In FIG. 3, prior to cot = 0, the phase-a current ia <0 and it is flowing through diode D4. At cot = 0, a gate current pulse iGI is applied to thyristor Q1. Then, Q, is turned on and D4 is turned off by the dc-side voltage Vd, which results in a switching of v,,,,.

Dring ira < 0, Q, continues to conduct, but, when it becomes ia > 0, Q, ceases to conduct and D, begins to conduct, which is the line-commutation process of Ql. However, v does not change. At (ot = 7r, a gate current pulse iG4 is applied to Q4. Then, Q4 is turned on and D, is turned off, which also results in a switching of vao. During i, > 0, Q4 continues to conduct, but, when it becomes/' < 0, Q4 ceases to conduct and D4 begins to conduct, which is the line-commutation process of Q4. However, i,,,, does not change.

Then, one period of the operation ends, and vao forms a square-wave which is in phase <BR> <BR> <BR> <BR> with the phase-a source voltage vaste The operations of phase b and phase c are identical<BR> <BR> <BR> <BR> <BR> <BR> with that of phase a except that they are phase shifted by 2/3 z and 4/3 #, respectively.<BR> <BR> <BR> <BR> <BR> <BR> <P>Then, vas, formes a well-known 6-pulse wave, which, together with Vasn makes the phase-a current i,. Because #asn and #an are in phase, i, lags #asn by #/2 and, hence no active

power flow is accomplished. Accordingly, the dc-side current id has a zero mean value.

This is the zero mode of operation.

If the gate current pulses are delayed by a firing angle 6, as shown in FIG. 4, vao lags #asn by #. Then, ia shifts to the left, which results in a positive active power flow.

Accordingly, the dc-side current id has a positive mean value. This is the rectification mode of operation. Conversely, if the gate current pulses are advanced by a firing angle 8, as shown in FIG. 5, v leads #asn by #. Then, ia shifts to the right, which results in a negative active power flow. Accordingly, the dc-side current id has a negative mean value. This is the inversion mode of operation.

FIGS. 8A-8C show the phasor diagrams corresponding to FIGS. 3-5, <BR> <BR> <BR> <BR> respectively, where Vas", Van, and Ia are the phasors of #asn, the fundamental component of va, and the fundamental component of i,, respectively.

Region of Possible Operation In the operation of the converter of the present invention, there is a condition under which commutation does not fail. In FIGS. 3-5, ia should be negative at cst <BR> <BR> <BR> <BR> = 8 when the gate current pulse iGI is applied to thyristor Ql, in order that commutation<BR> <BR> <BR> <BR> <BR> <BR> may not fail. Likewise, ia should be positive at cot = n + 6 when the gate current pulse iG4 is applied to thyristor Q4, also in order that commutation may not fail.

Now, if it is supposed that/' is purely sinusoidal for simplicity, the above condition equals that i, should lag v,,,,. Then, the condition can be represented by a phasor diagram, as shown in FIG. 9, where the shaded area within the circle is the region <BR> <BR> <BR> <BR> of possible operation of Vall. However, since the real shape of i, is not purely sinusoidal, a waveform analysis is necessary to obtain a real region of possible operation.

FIG. 10A shows an equivalent circuit of FIG. 1 for the waveform analysis.

FIGS. 10B and 10C are circuits for applying the superposition principle to the circuit of FIG. 10A. By the superposition principle, ia is given as the difference between il and i2.

FIGS. 11-13 show waveforms in the zero mode, the rectification mode, and the inversion mode, respectively. The waveform of i, is invariable because i, is the current by a fixed source. However, the waveform of i2 grows in magnitude as the dc-side voltage Vd increases, and also the waveform of i2 shifts with respect to the phase-angle according to <BR> <BR> <BR> <BR> the firing angle 8. Then, the above condition that/ should be positive at O) t = 7E + 8 when the gate current pulse ic4 is applied to thyristor Q4 equals that the peak of i2 <BR> <BR> <BR> <BR> should be less than !), because reaches its peak at c3t = s + 8. FIG. 14 shows a range of possible operation of the firing angle 8 for a given value of the dc-side voltage Vd.

Consider a source voltage #asn wioth its rms value Vs : vasn =V2sm(().

Then, il is given by and the peak of i2 can be calculated at (ot = # + 8 as Then, the above condition that the peak of i, should be less than i, is expressed as which gives Now, consider the rms value V, of the fundamental component of v,. Because v, is a six-pulse wave, V, always relates to the dc-side voltage Vd as #2 V1=Vd # (6) Then, (5) is written in terms of V, as <BR> <BR> <BR> #2<BR> <BR> <BR> cos(#)# V1 9Vs (7) and, if a conversion factor k is introduced as <BR> <BR> <BR> <BR> <BR> <BR> k=##<BR> S then, (7) is written in terms of k as <BR> <BR> <BR> <BR> 2<BR> #<BR> cos(#)# k 9 (9) The maximum value of the conversion factor k is obtained from (9) at 8 = 0 as 9 kmax= #0.912<BR> <BR> <BR> <BR> #2 (10) If the waveform of v,"is purely sinusoidal, the maximum value of k is 1. But, in reality, the harmonics in vp prevent k from reaching 1. It is worth noting at this point

that FIGS. 3-5 and others so far are for the case that k = 0. 8. From (7), the range of possible operation of 8 for a given value of V, is given as The inequality of (7) or (11) can be represented by a phasor diagram. Recall that V,,, is written in polar coordinates as Van = V1e-j# = kVse-j# (12) For a given value of V,, the range of possible operation of 8 is calculated from (11), and then, the points of possible operation of can be drawn in a complex plane.

By varying V, (or k) from 0 to its maximum, the region of possible operation of Vn is obtained as shown in FIG. 15 by the shaded area within the circle. The dashed circle, which is the one in FIG. 9 on the supposition of purely sinusoidal ia, is also shown for comparison. In effect, (7) is a circular function which can be written in rectangular coordinates as where x = V cos (8), y = V1 sin (8).

Characteristics 1. Bidirectional and Controllable DC-Side Current As can be seen from FIGS. 3-5, the average value of the dc-side current can be controlled from a negative minimum to a positive maximum value in a continuous manner by controlling o. The average value Id of the dc-side current id can be calculated from the waveforms in FIGS. 3-5 and FIGS. 11-13 as a function of 8 : Equation (14) is the dc-side current control characteristic. FIG. 16 shows the <BR> <BR> control characteristic for k = 0. 8. In FIG. 16, bmaX iS the value obtained from (9), which is

about 28. 68°°. As shown in FIG. 16, the control characteristic is fairly linear when k is high.

2. Regulated DC-Side Voltage As explained in the above, the converter of the present invention provides a bidirectional and controllable dc-side current capability. As a consequence, the dc-side voltage can be regulated at its desired value by controlling the dc-side current with the firing angle 8 in response to changes in load.

If the dc-side voltage is regulated, referring to (6) and (12), the magnitude of V,, becomes constant. Then, the locus of forms an arc of which the examples are shown in FIGS. 17A and 17B with thick lines. FIGS. 17A and 17B are for A-= 0. 8 and k= 0. 85, respectively.

3. Rating of AC-Side Inductors FIGS. 18A and 18B show the loci of Vps"-Vp", which is the voltage across the ac-side inductor. As shown in FIGS. 18A and 18B, the voltage across the inductor has its maximum value of 0. 486 Vs when k = 0. 8 and 0. 385 Vs when k = 0. 85. Thus, if the source voltage and power are selected as per-unit bases, the voltage across the inductor dictates the required per-unit rating of the ac-side inductors, which is 0. 486 (pu) when k = 0. 8, and 0. 385 (pu) when k = 0. 85.

4. Reactive Power FIGS. 19A and 19B show the loci of the ac-side current 1,. As shown in FIGS.

19A and 19B, the converter of the present invention always generates lagging reactive power, and the reactive power decrease as k is increased. Note that the maximum active power also decrease as k is increased. FIG. 20 shows the reactive power characteristic of the converter of the present invention, which is obtained from the following equations : As shown in FIG. 20, the reactive power in the converter of the present invention is smaller than that in the conventional thyristor phase-controlled CSC for a larger part of the operating condition.

5. AC-Side Harmonics As seen from FIGS. 11-13, although the shape of the ac-side current ia varies largely with 8, its harmonic contents are retained independent of 8 provided that k is constant. FIG. 21 shows the ac-side harmonics characteristic in terms of the ac-side ripple power QnPple, which is defined as As shown in FIG. 21, the ac-side harmonics in the converter of the present invention are noticeably smaller than those in the conventional thyristor phase-controlled CSC. Q"PP'e <BR> <BR> <BR> in the converter of the present invention does not exceed 10% of pmaX when k = 0. 8 and<BR> <BR> <BR> <BR> 13% of P"when k = 0. 85.

6. DC-Side Harmonics The amount of the dc-side harmonics is the determining factor of the dc-side capacitance. FIG. 22 shows the dc-side harmonics characteristic in terms of the dc-side ripple power Qd pple, which is defined as Because the dc-side harmonics in the conventional thyristor phase-controlled CSC are not currents, but voltages, QdriPP"in the conventional thyristor phase-controlled CSC shown with dashed lines is obtained from the following equations : As shown in FIG. 22, the dc-side harmonics in the converter of the present invention are also noticeably smaller than that in the conventional thyristor phase- controlled CSC.

7. Recommended Range of k FIGS. 17-22 have shown the characteristics for the conversion factor k only from 0. 8 to 0. 85. In general, as k is decreased, the ac-side harmonics decrease, but the

rating of the ac-side inductor, the reactive power, and the dc-side harmonics increase.

On the other hand, if k goes beyond about 0. 88, the rating of the ac-side inductor and the dc-side harmonics decrease, but the reactive power and the ac-side harmonics increase largely. Consequently, around 0. 8-0. 85 is recommended as a range of k for good performance.

Implementation and Experimental Verification The gate firing controller 21 and the firing angle controller 22 can be implemented in various ways using analog circuitry, digital circuitry, and microprocessors.

And it is believed that the drawings and the descriptions presented so far are sufficient for those skilled in the art to implement the controllers without further detailed description.

The current feedback controller 61 and the voltage feedback controller 71 can be implemented in various ways. The well-known proportional feedback controller and the well-known proportional-integral feedback controller are examples. And it is believed that the controllers can be implemented by those skilled in the art without further detailed description.

FIG. 23 shows an experimental system to verify the operation of the converter of the present invention. The experimental system is started up with the help of the bipolar junction transistor (BJT) converter. After the startup, the converter of the present <BR> <BR> <BR> <BR> invention is activated and the BJT converter is disconnected. Then, the firing angle 6 is controlled manually to regulate the dc-side voltage in response to changes in the load (a variable and reversible current source). The experimental results for FIGS. 3-5 are shown in FIGS. 24-26, respectively. The results well confirm the operation of the converter of the present invention.

INDUSTRIAL APPLICABILITY As described above, the converter of the present invention provides a bidirectional dc-side current capability, a reduced reactive power characteristic, a reduced ac-side harmonics characteristic, and a reduced dc-side harmonics characteristic. The converter of the present invention also has characteristics that the average value of the dc- side current can be controlled from a negative minimum to a positive maximum value in a continuous manner, and the dc-side voltage can be regulated at its desired value.

Therefore, the converter of the present invention can be used for applications wherever one or more of these characteristics are useful. The VSI motor drive system with regenerative braking will be an important application. The HVDC transmission system

of voltage source type will be another important application, which has an advantage over the conventional HVDC transmission system of current source type, especially in multi- terminal configurations.

While this invention has been described in connection with what is presently considered to be the most practical and preferred embodiment, it is to be understood that the invention is not limited to the disclosed embodiment, but, on the contrary, it is intended to cover various modifications and equivalent arrangements included within the spirit and scope of the appended claims.