FIELD OF THE INVENTION [0002] The present invention relates to variable gain amplification systems and methods. More specifically, the present invention relates to variable gain amplification systems, including variable gain amplification systems as integral components of mass spectrometers, and methods for using variable gain amplifiers in ion detectors, including time of flight mass spectrometers.

BACKGROUND OF THE INVENTION [0003] Time of flight mass spectrometry (TOFMS) is an analytical process that determines the mass-to-charge

ratio (m/z) of an ion by measuring the time it takes a given ion to travel a fixed distance after being accelerated to a constant final velocity. This time is commonly referred to as the ion's time of flight ("TOF").

There are two fundamental types of time of flight mass spectrometers: those that accelerate ions to a constant final momentum and those that accelerate ions to a constant final energy. Because of various fundamental performance parameters, constant energy TOFMS systems are preferred. A previously known constant kinetic energy TOFMS is shown in FIG. 1A. Ions are created in region 13 typically referred to as the ion source. Two ions with masses M1 and M2 have been created as shown in FIG. 1A. A uniform electrostatic field created by the potential difference between repeller lens 10 and ground aperture 11 accelerates ions Mi and M2 through a distance S. After acceleration, ions pass through ground aperture 11 and enter ion drift region 14 where they travel a distance x at a constant final velocity prior to striking ion detector 12.

[0004] The TOFMS of FIG. 1A is a parallel extraction device. In a parallel extraction mass spectrometer such as shown in FIG. 1A, analog-to-digital conversion may be accomplished by a signal digitizing device such as, for example, time-array recording device 17 which may be a transient recorder or a digital oscilloscope. Digitized signals output by time array recording device 17 are ultimately processed in a software program at stage 18 typically running on a digital computer.

[0005] The time of flight of the ions can be measured to calculate their mass-to-charge ratios. The relationship between ion mass-to-charge ratio and total time of flight is given in the following equation (5).

Referring to FIG. 1A, within the ion optic assembly, accelerating electrical field (E) is taken to be the potential difference (V) between the two lens elements (10 and 11) as applied over acceleration distance s, (E = V/s). Equation (1) defines the final velocity (v) for ion Mi with charge z. The final velocity of ion M2 is determined in a similar manner.

Equation (1) [0006] Inverting equation (1) and integrating with respect to distance s yields equation (2), which describes the time spent by ion Mi in the acceleration region (te) : Equation (2) The total time of flight for ion M1 (tt) is then derived by adding ts to the time spent during flight along distance x (ion drift region 14). Time tg equals the product of the length of free flight distance x with 1/v, as shown in Equation (3): Equation (3) Rearranging equation (3) in terms of M1/z yields equation (4):

Equation (4) For all TOFMS systems, E, s, and x are intentionally held constant during analysis, thus equation (4) can be reduced to equation (5), where K is a constant: M, » t, Equation (5) [0007] Determining the value for k is referred to as calibrating the TOFMS system and is achieved by measuring the time of flight for various ions of known m/z and determining a least squares fit for the value of k within equation (5). A computer software program routinely supports data analysis and calibration. After calibration, determining the m/z of an unknown ion is a simple matter of measuring its total flight time and plugging this determined time into equation (5).

Equation (5) shows that the total time of flight of any ion is directly proportional to its mass to charge ratio.

Thus, in FIG. 1A, it can be seen that the m/z of M1 is less than the m/z of M2. Furthermore, it follows that the total time of flight of any ion is inversely proportional to the square root of its charge (z) and directly proportional to the square root of its mass (m).

[0008] Equations (1)- (5) simplify the TOFMS process by assuming that all ions are created at the same time, within the same location, and have no initial velocity prior to acceleration. Routinely, this is not the case and in many instances, variations in formation time,

original location, and initial velocity (also referred to as initial energy) are often demonstrated for various ions of a given m/z population. Such variations ultimately limit the mass resolving power of the instrument. Mass resolving power is typically defined as the ability to determine subtle differences in m/z. For a TOFMS system, mass resolving power R is mathematically defined by equation (6), where m is the determined mass of an ion, T is the ion's TOF, and dm and dt are the respective full mass or full temporal width of a measured signal at its half magnitude: R_ m _ T dm 2dt Equation (6) [0009] Ultimately, factors that limit mass resolving power are dictated by the ionization means, geometry of the ionization source, geometry and stability of the TOFMS, as well as the nature of the sample itself.

Various strategies have been adapted to improve mass resolving power in time-of-flight mass spectrometry. The most successful approaches include use of time lag focusing (TLF), ion mirrors or reflectrons, and the combination of these two.

[0010] Another previously known TOFMS is shown in FIG.

1B. The TOFMS of FIG. 1B is an orthogonal extraction device that uses a reflectron. In the device of FIG. 1B, ions are generated from ion source 20 and directed to repeller lens 22 via RF ion guide 21. A uniform electrostatic field created between repeller lens 22, extractor lenses 29, and ground apertures 28 accelerates ions. After acceleration, ions pass through ground apertures 28 and enter an ion drift region along path 35

where they travel through reflectron 27. Reflectron 27 functions to narrow ion energy spread, and then it redirects the ions to detector 26.

[0011] The output signal of ion detector 26 is then converted to a digital signal for further processing.

Ideally, in an orthogonal extraction mass spectrometer, a single ion per mass-to-charge ratio is transmitted in each ion detection cycle. Therefore, analog-to-digital conversion may be accomplished, for example, by using a time-interval recording device, such as a time-to-digital converter (TDC). Detector 26 outputs a signal to high speed time-to-digital converter (TDC) 24 when an ion impacts its detecting surface. TDC 24 converts analog signals from detector 26 to digital information suitable for software processing at stage 25. TDC 24 records a single impulse when the detector 26 output signal exceeds a predetermined threshold. HV pulser 23 indicates to TDC 24 the start of an ion detection cycle when ions are beginning to be accelerated by repeller lens 22.

[0012] Previously known systems have employed means for providing gain in the output signal of detector 26 prior to digitization. Such gain has been provided by primary ion to secondary product or primary ion to secondary electron conversion prior to striking an electromissive detector surface. Primary ions are converted to secondary products through the mechanisms of surface induced dissociation, generating ion and neutral fragments, and/or fast ion bombardment of solid surfaces, creating sputtered products. Primary ions can also be converted to secondary electrons by directing them to strike a metal of low work potential, ultimately releasing low energy electrons. These secondary products are then directed to strike an electromissive device,

creating an amplification cascade provided by the generation of secondary, tertiary, quaternary, etc. electrons. Typical electromissive devices used in such applications include the discrete dynode, electron multiplier, and microchannel plate assemblies.

[0013] The actual gain provided by the detection of ions in TOFMS is mass dependent. Ion detection probability has been shown to be related to final ion velocity. FIG. 1C depicts the ion to electron conversion probability for ions of various mass-to-charge ratios (m/z) at two different kinetic energy levels: 50 KeV (line 30) and 25 KeV (line 31). As the mass-to-charge ratio of ions increases, the probability of secondary electron emission decreases as shown in FIG. 1C. As a direct consequence of this, overall gain and attendant sensitivity decreases with increasing ion molecular weight. Placing an electron amplifier between the detector 26 output and signal processing electronics often improves additional signal gain. Such amplifiers have typically been chosen so that their bandwidth and gain characteristics closely match some established ideal mass range being scanned at any given time. However, TOFMSs employing such amplifiers often suffer from low resolution and high frequency noise.

[0014] Time of flight mass spectrometers typically have several sources of signal noise including sampling noise, Johnson noise, flicker noise, and high frequency noise created by the detection apparatus. The major detection signal frequency components of ions with lower mass-to-charge ratios are typically much higher than ions with higher mass-to-charge ratios. If a broad bandwidth amplifier is used to accurately depict the signals of low molecular weight ions to preserve analyzer resolving

power for such ions without adding undue peak broadening, unneeded amplification of high frequency noise is created during the detection of high molecular weight ions.

[0015] This ultimately reduces the system's sensitivity to high molecular weight ions by both diminishing the signal-to-noise ratio, as well as by possibly saturating the amplifier and/or the analog-to- digital converter headroom, the latter reducing the system's vertical dynamic range. The signal-to-noise ratio may be improved by using a variable or fixed bandwidth digital filter. However, a digital filter does not restore the diminished vertical dynamic range. If a low bandwidth amplifier is utilized, high molecular weight range detection sensitivity may be augmented, but at the expense of broadening low molecular weight ion signals effectively minimizing low molecular weight ion resolving power.

[0016] It would therefore be desirable to provide an amplification system for a mass spectrometer that provides adequate resolving power and signal-to-noise ratio for low and high molecular weight ions without introducing additional high frequency noise.

[0017] It would also be desirable to provide an amplification system for a mass spectrometer that increases the sensitivity of higher molecular weight ions by amplifying only those frequency domains that contain ion signal information.

SUMMARY OF THE INVENTION [0018] It is therefore an object of the present invention to provide an amplification system for a mass spectrometer that provides adequate resolving power and

signal-to-noise ratio for low and high molecular weight ions without introducing additional high frequency noise.

[0019] It is therefore an object of the present invention to provide an amplification system for a mass spectrometer that increases the sensitivity of higher molecular weight ions by amplifying only those frequency domains that contain ion signal information.

[0020] Variable gain amplification systems and methods of the present invention provide increased sensitivity for detecting ions with a broad range of mass-to-charge ratios in a mass spectrometer. Variable gain amplification systems and methods of the present invention increase the gain provided to the output of an ion detector in a mass spectrometer for ions with higher mass-to-charge ratios. By increasing the gain for ions with higher mass-to-charge ratios, the system provides improved sensitivity for high molecular weight ions.

[0021] One embodiment of the present invention includes a variable gain amplification system with a constant gain-bandwidth product that has an increased signal-to-noise ratio and increased resolution. This embodiment includes a voltage mode amplifier. Another embodiment of the present invention includes a variable gain, constant bandwidth amplification system. This embodiment includes a current mode amplifier.

BRIEF DESCRIPTION OF THE DRAWINGS [0022] The above-mentioned objects and features of the present invention can be more clearly understood from the following detailed description considered in conjunction with the following drawings, in which the same reference numerals denote the same structural elements throughout, and in which:

[0023] FIG. 1A is a prior art parallel extraction time of flight mass spectrometer; [0024] FIG. 1B is a prior art orthogonal extraction time of flight mass spectrometer; [0025] FIG. 1C is a graph of the ion-to-electron conversion probability for ions with different mass-to- charge ratios at 25 and 50 KeV of total kinetic energy; [0026] FIG. 2 is a block diagram of a mass spectrometer with a signal amplification system in accordance with the principles of the present invention; [0027] FIG. 3 is a block diagram of a variable gain amplification system for a mass spectrometer in accordance with the principles of the present invention; [0028] FIG. 4 is a schematic diagram of a variable gain amplification circuit for a mass spectrometer in accordance with the principles of the present invention; [0029] FIG. 5 is a graph of timing signals for the variable gain amplification circuit of FIG. 4; [0030] FIG. 6 is a simplified schematic of the variable impedance circuit shown in FIG. 4; [0031] FIG. 7A is a graph illustrating bandwidth curves in a variable gain amplification system of the present invention; [0032] FIG. 7B is a graph illustrating gain curves in a variable gain amplification system of the present invention; and [0033] FIG. 8 is a graph illustrating mass spectra for a prior art mass spectrometer and a mass spectrometer with a variable gain amplification system of the present invention.

DETAILED DESCRIPTION OF THE INVENTION [0034] The variable gain amplification system of the present invention may be used in various mass spectrometers, but is best suited for use in a time of flight mass spectrometer (TOFMS) and various TOF tandem hybrid systems such as quadrapole-TOFMS, an ion trap- TOFMS, an electrostatic analyzer-TOFMS, and a TOF-TOF MS.

A block diagram of a time of flight mass spectrometer with an amplification system is shown in FIG. 2. The mass spectrometer of FIG. 2 may be configured as a parallel extraction device or an orthogonal extraction device.

[0035] A sample containing matter that is to be analyzed by the mass spectrometer is introduced through sample inlet system 70. The sample may be introduced as a solid, liquid, or gas. The sample is transferred into ion optics 72. Ionization source 60 causes a portion of the sample to become an ionized gas in ion optics 72.

Ionization source 60 may comprise a laser desorption ionization device, a plasma desorption ionization device, a fast atom bombardment ionization device, an electron ionization device, a chemical ionization device, or an electrospray ionization device. A laser desorption device may be used to performed laser desorption/ionization, surface-enhanced laser desorption/ionization, and/or matrix-assisted laser desorption/ionization (MALI).

[0036] Ion optics 72 accelerates ions toward mass analyzer 74. Ion optics 72 may, for example, comprise electrostatic lenses such as a repeller lens and ground aperture as discussed above. Mass analyzer 74 directs the ions to ion detector 76. In a TOFMS, mass analyzer 74 comprises the free flight region of the ions after

they are accelerated. TOFMS analyzers may comprise a linear system, in which ion free-flight occurs with rectilinear motion, or a reflected system, in which ions are turned about in an ion mirror or reflectron by an array of electrostatic sectors. Ion detector 76 may comprise, for example, a microchannel plate detector, multi-stage electron multiplier, or a hybrid combination of these. Ion detector 76 detects ions that impact its detecting surface and passes an output signal indicative of detected ions to signal amplifier 78.

[0037] Signal amplifier 78 may comprise a variable gain amplifier in accordance with the principles of the present invention. Variable gain amplifiers of the present invention amplify the output signal of detector 76 and provide appropriate frequency response and amplification gain to enhance ion detection sensitivity.

Signal amplifier 78 outputs a signal to data acquisition device 80, which converts the analog output to a digital signal suitable for software processing using, for example, an analog-to-digital converter. Analog-to- digital conversion may be accomplished, for example, using a time-interval recording device, such as a time- to-digital converter, in an orthogonal extraction mass spectrometer, or using a time array recording device such as a transient recorder or a digital oscilloscope, in a parallel extraction mass spectrometer. Data acquisition device 80 then transfers that digital signal to computer 82 which processes the data to determine the mass-to- charge ratio of the detected ions.

[0038] The overall detection efficiency for ions in a typical time of flight mass spectrometer generally decreases with increasing molecular weight.

Consequently, a given population of low molecular weight

ions produce stronger detection signals when compared to an identical number of higher molecular weight ions. For ions with 25 keV of total kinetic energy, such diminished detection efficiency is first seen around 5-10 kilodaltons (kDa) as shown in FIG. 1C, line 31.

Furthermore, ion populations with lower mass-to-charge ratios produce detection signals that have comparatively higher frequency components than ions with larger mass- to-charge ratios as shown in the following table (1) that describes typical ion flight time, target resolution, and major frequency components (as determined by required peak width to obtain target resolution), for ions with 25 keV of total energy as analyzed in a one-meter, linear, time lag focusing mass spectrometer: Ion Mass-to-Target Ion Flight Major Charge Ratio Resolution Time (uSec) Component (m/z) Frequency (MHZ) 500 4, 000 10. 2 740 1,000 4, 000 14. 4 500 2,000 3, 000 20. 4 250 3,000 2, 000 24. 9 120 5,000 1, 000 32. 2 70 15,000 1, 000 55. 8 19 40,000 400 91. 1 2 150,000 50 176. 3 0.290 250,000 40 227. 6 0.130 500,000 35 321. 9 0.063 Table (1)

[0039] Consequently, it is desirable to provide an amplification scheme that matches both gain and bandwidth requirements when amplifying signals of ions of various mass-to-charge ratios (m/z) in TOFMS. For low m/z ions, ion-to-electron conversion efficiency is high, thus diminishing the need for additional amplification.

Furthermore, if mass resolving power for low molecular weight ions is to be conserved, any attendant amplification must be achieved without diminishing required frequency response. For high m/z ions, ion-to- electron conversion efficiency decreases, thus creating a need for further signal amplification. Additionally, the frequency response required to accurately reproduce high molecular weight ions is such that total bandwidth may be constrained to eliminate unwanted high frequency noise, providing signal amplification without undue amplification of extraneous high frequency signals, improving analytical sensitivity for high m/z ions while preserving analog-to-digital converter headroom.

[0040] As previously explained for a given TOF geometry and acceleration potential, low molecular weight ions have shorter times of flight than larger molecular weight ions. Therefore, low molecular weight ions begin to impact the ion detector before larger molecular weight ions. Ion detection signal gain should increase for higher molecular weight ions to compensate for the fact that higher molecular weight ions possess comparatively diminished detection efficiency with respect to low m/z ions. The variable gain amplification system of the present invention increases the gain G provided to the output signal relative to the input signal as a function of time. Thus, as the molecular weight of ions striking the ion detector increases, the amplification system of

the present invention increases the gain factor G. The gain G may be proportional to the square of the ion detection time. Because of limitations of various components employed by the amplifier, there is a given ion detection time for which the amount of amplifier gain reaches a maximum.

[0041] Signal amplifier 78 of FIG. 2 may comprise signal amplification circuit 100 shown in FIG. 3.

Amplification circuit 100 is an embodiment of the variable gain amplification system of the present invention. Amplification circuit 100 receives an input signal voltage VIN from detector 76 and provides an output signal voltage VOUT to data acquisition device 80. Input impedance matching network 101 sets the input impedance of circuit 100. Input voltage limiting circuit 102 limits the input voltage range of VIN. Amplifier 103 amplifies signal VIN to provide signal VOUT. In a variable gain, constant bandwidth embodiment of the present invention, amplifier 103 in FIG. 3 is a current mode amplifier. In a constant gain-bandwidth product embodiment of the present invention, amplifier 103 in FIG. 3 is a voltage mode amplifier. Output impedance matching network 104 sets the output impedance of circuit 100.

[0042] Ramp initialization circuit 114 generates a voltage square waveform. The square waveform output signal of circuit 114 is integrated into a voltage triangular waveform V1 by an integrator circuit within linear ramp generator 115. The triangular waveform output signal V, of linear ramp generator circuit 115 is integrated by an integrator circuit inside positive parabolic voltage generator 112 to produce a rising parabolic voltage waveform output signal V2. Negative

parabolic voltage generator 113 comprises an inverting amplifier that inverts the output signal V2 of positive parabolic voltage generator 112 and outputs a falling parabolic voltage waveform signal V3.

[0043] Positive parabolic voltage generator circuit 112 and negative parabolic voltage generator circuit 113 provide rising and falling parabolic waveform signals V2 and V3 to variable impedance circuit 107 via voltage-to- current converters 110 and 111, respectively. Feedback network circuit 105 and variable impedance circuit 107 control the gain of amplifier 103 in response to rising and falling parabolic signals V2 and V3 of generator circuits 112 and 113. The impedance of variable impedance circuit 107 varies in response to the parabolic signals V2 and V3 of positive and negative parabolic generators 112 and 113. As the impedance of circuit 107 varies, the gain provided by amplifier 103 to VOUT varies.

[0044] Ions are sent to the ion detector of the mass spectrometer in ion detection cycles. A dead time may exist between ion detection cycles when no ions impact the ion detector and signals inside circuit 100 are reset to their initial values. Once one cycle is complete, the next set of ions can be sent to the ion detector.

[0045] Timing and synchronization circuit 117 sends a signal VRP to ramp initialization circuit 114 causing circuit 114 to step up its output voltage at the beginning of an ion detection cycle. Positive and negative parabolic signals V2 and V3 are about zero volts at the beginning of an ion detection cycle. After the start of an ion detection cycle, signal V2 ramps up in voltage, and signal V3 ramps down in voltage. At the end of an ion detection cycle, Vs and V3 are respectively reset to zero volts by circuits 114 and 116.

[0046] Ions with lower molecular weights have a shorter time of flight than ions with higher molecular weights. Therefore, lower molecular weight ions impact the ion detector before the higher molecular weight ions in a time of flight mass spectrometer (TOFMS). Circuit 100 is designed so that the gain of amplifier 103 increases later in the ion detection cycle when higher molecular weight ions impact the ion detector. This provides greater sensitivity for higher molecular weight ions.

[0047] At the start of an ion detection cycle when low molecular weight ions impact the ion detector, signals Vs and V3 are near zero volts, and the impedance of variable impedance circuit 107 is large causing the gain of amplifier 103 to be small. As ions of increasing molecular weight begin to strike the ion detector, signal V2 increases, signal V3 decreases, and the impedance of variable impedance circuit 107 decreases causing the gain of amplifier 103 to increase. Thus, as higher molecular weight ions impact the ion detector, the gain of amplifier 103 increases to provide greater sensitivity for these ions.

[0048] Maximum gain limiter circuit 108 provides an upper limit to the gain provided by amplifier 103 to VoUT.

At the end of an ion detection cycle, timing and synchronization circuit 117 sends a signal VRP to ramp initialization circuit 114 that causes its output voltage to step down and the output signals V2 and V3 of positive and negative voltage parabolic generators 112 and 113 to be reset to zero. Timing and synchronization circuit 117 sends a signal VCL to pre-ramp clamp circuit 106 causing it to reset Vouer to zero volts or some other predetermined voltage prior to the beginning of the next ion detection

cycle. Reset and stabilization circuit 116 is a loop circuit that is open during an ion detection cycle as ions impact the ion detector. Circuit 116 is closed loop after each ion generation when output signals V2 and V3 of generators 112 and 113 are resetting to zero in response to signal VRS from timing circuit 117.

[0049] A detailed schematic diagram of a variable gain amplification system of the present invention is shown in FIG. 4. Amplification system 200 contains circuit elements 101-116 that correspond to the block diagram of FIG. 3 and additional circuit components. Input impedance matching network 101 comprises a resistor coupled between input signal VIN and ground. Output impedance matching network 104 also comprises a resistor coupled between the output of amplifier 103 and VouT.

Input voltage limiting circuit 102 comprises diodes 201- 204. Diodes 201-204 clamp input voltage VIN to within a voltage range that is determined by the voltage drop across these four diodes. Diodes 203-204 are Schottky diodes and may have a forward biased diode voltage drop that is less than the forward biased voltage drop of diodes 201-202. The maximum voltage of VIN is the diode voltage drop of diode 204, and the minimum voltage of VIN is the sum of the diode voltage drop of diodes 201-203.

For example, if the diode voltage drop of diodes 201-202 is 0.7 volts and the diode voltage drop of diodes 203-204 is 0.3 volts, then the maximum voltage of VIN is 0.3 volts and the minimum voltage of VIN is-1.7 volts. Amplifier 103 operates between supply voltages Vsl+ and Vsl- (e. g., +5 and-5 volts, respectively). Amplifier 103 in FIG. 4 may be a voltage mode amplifier.

[0050] Voltage signals VCL, VRS, and VRP are digital signals that control the timing and operation of circuit

200. Signals VCL, VRS, and VRP in FIG. 4 control the opening and closing of switches 271,263, and 250, respectively. Example waveforms for signals VCL, VRS, and VRP are shown in FIG. 5. At time tl in FIG. 5, VCL goes LOW causing switch 271 to open decoupling resistor 270 from resistor 272. Pre-ramp clamp circuit 106 comprises a loop circuit around the feedback network of amplifier 103 that is open when VCL is LOW. VRP also goes LOW at time tl causing switch 250 to couple resistor 251 to ground. VRS goes HIGH at time tl causing switch 263 to close. Reset and stabilization circuit 116 comprises a loop circuit around linear ramp generator 115 and parabolic voltage generators 112 and 113 that is closed when VRS is HIGH.

[0051] In circuit 200, linear ramp generator circuit 115 is an integrator circuit that comprises input resistor 251, amplifier 230, and capacitor 231, where capacitor 231 integrates the current flowing through resistor 251. Positive parabolic voltage generator 112 is also an integrator circuit that comprises input resistor 232, amplifier 234, and capacitor 233, where capacitor 233 integrates the current flowing through resistor 232. Negative parabolic generator 113 includes input resistor 237, inverting amplifier 240, and feedback resistor 241. Capacitors 236 and 242 compensate for the finite bandwidth of amplifier 240.

[0052] The output signals of amplifiers 230,234, and 240 are voltage signals V1, V2, and V3, respectively.

Before time tl, V, and V3 are at a negative voltage, and V2 is at a positive voltage. Signals Vl, V2, and V3 all should be substantially at ground before an ion detection cycle begins. Switch 250 couples resistor 251 to ground, and switch 263 closes at time tl so that signals Vl, V2,

and V3 can be brought substantially to ground. The inverting input of amplifier 230 is pulled from a positive voltage down to ground through resistor 251 after time tl. Between times tl and t2 (e. g., a time lapse of about 350-400 usec.), V, is pulled up to zero volts by amplifier 230. The voltage at the inverting input of amplifier 234 increases as V1 rises, causing voltage V2 at the output of amplifier 234 to decrease to zero. V3 is the inversion of V2 and therefore rises to ground in the same manner that V2 falls to ground. Diode 229 prevents V2 from falling more than a diode voltage drop below ground, and diode 239 prevents V3 from increasing a diode voltage drop above ground.

[0053] Resistors 265 and 260 comprises the DC feedback path around the two integrators 115 and 112, and inverter 113. However, such a configuration by itself is unstable. To stabilize this loop, a pole-zero combination comprising resistor 261 and capacitor 262 may be introduced. This circuit, while somewhat oscillatory, is nevertheless stable, and settles to zero volts well within the time that VRS is high. Example component values include 20.5 kQ for resistor 260,1.78 kQ for resistor 261, and 2.2 nF for capacitor 262. Other appropriate component values may be used. Diode 266 maintains the voltage at the node between resistor 265 and switch 263 between a diode drop above ground and its breakdown voltage below ground.

[0054] By time t2, voltage signals Vl, V2, and V3 are all substantially at zero volts. At time t2, VRS transitions from HIGH to LOW, opening switch 263. The time lapse between times t2 and tO (e. g., 20 psec.) is sufficient to allow enough lead time for voltage transients caused by opening switch 263 to settle out.

At time tO, an ion detection cycle begins, and ionization source 60 begins to generate ions. The time between tO and t3 provides an appropriate delay until a time at which amplifier bandwidth does not result in significant distortion of poorly resolved ion populations. Because the optimum delay between tO and t3 may be dependent on the characteristics of a particular embodiment, this delay may be placed under the control of the operator of the mass spectrometer. The time between tO and t3 may be, for example, about 60 ps. Of course, longer or shorter time intervals may be used.

[0055] At time t3, VRP transitions from LOW to HIGH (FIG. 5), causing switch 250 to couple resistor 251 to resistor 252. The current flowing toward the inverting input of amplifier 230 is now Vsl+ (e. g., 5 volts) divided by the values of resistors 251 and 252 (e. g., 40.2 kQ and 2 kQ, respectively). This current flowing toward the inverting input of amplifier 230 continues to flow away from this input through capacitor 231, causing V1 to decrease linearly until amplifier 230 saturates.

Integrator circuit 112 receives the negative going linear signal V1 at the inverting input of amplifier 234 through resistor 232 and integrates it to provide an increasing positive parabolic voltage output signal V2. Circuit 113 receives the positive going parabolic signal V2 at the inverting input of amplifier 240 through resistor 237 and inverts it to provide a decreasing negative parabolic voltage output signal V3.

[0056] Referring to FIG. 4, VIN is coupled to the non- inverting input of amplifier 103. A feedback loop circuit is coupled between the inverting input and the output of amplifier 103. The feedback loop comprises feedback network 105, variable impedance circuit 107, and

maximum gain limiter circuit 108. When amplifier 103 operates in its linear region, amplifier 103 drives its output voltage VG so that the voltages at its inverting and non-inverting inputs are approximately equal. The output voltage Vs of amplifier 103 is determined by VIN and the gain G of amplifier 103, as shown in the following equation: yc-V iN G Equation (7) [0057] The gain G of amplifier 103 is determined by impedance of feedback network 105, variable impedance circuit 107, maximum gain limiter circuit 108, and associated circuitry. Variable impedance circuit 107 has four Schottky diodes 211-214 that have an impedance that varies as the current through diodes 211-214 varies.

When the impedance of diodes 211-214 decreases, the gain G of amplifier 103 increases. Positive parabolic voltage generator 112 outputs an increasing parabolic voltage waveform V2, and negative parabolic voltage generator 113 outputs a decreasing parabolic voltage waveform V3, as previously discussed. As V2 increases and V3 decreases, the currents through diodes 211-214 increase, and the impedances of diodes 211-214 decrease, as shown by the following equations. The relationship between the current through a diode and the voltage across a diode is shown by equation (8): Equation (8) where i is the current through the diode, Io is the reverse leakage current, e is the base of Naperian or

natural logarithm, q is the elementary charge of an electron, V is the voltage applied to the diode, K is the Boltzmann constant, n is a value between 1 and 2 (dependent on the doping level of the crystal), and T is the temperature in degrees Kelvin. At room temperature, KT/q equals 25.6 mV. The impedance of a diode is defined by the following equation: e Z d V KT, 7 Equation (9) [0058] For currents greater than about 30 pA, eqV/KTn > > 1. Therefore, the impedance of a diode may be expressed in terms of the current i through the diode using equations (8) and (9) as shown in the following equation: Z = 25.6mV#n i Equation (10) [0059] Equation (10) shows that the impedance across a diode decreases as the current through the diode increases. Thus, the impedances of diodes 211-214 decrease as the currents through these diodes increase.

The gain G provided by amplifier 103 to signal VG is determined by the voltage drop across resistor 208 in FIG. 4, the voltage drop across diodes 211-214, and the voltage drop across resistor 108.

[0060] For a given current through each of diodes 211- 214, their impedances are constant, and each of diodes 211-214 acts as a resistor. The four resistors corresponding to diodes 211-214 can be reduced to their Thevenin equivalent between node 282 and the node coupled to the inverting input of amplifier 103. The Thevenin equivalent resistance of diodes 211-214 equals the

resistance of each one of diodes 211-214 separately, assuming all four diodes have about the same impedance.

For example, if the resistance of each one of diodes 211- 214 is 700 Q separately, then the Thevenin equivalent resistance of the four diodes 211-214 is 700 Q. The Thevenin equivalent resistance of diodes 211-214 can be thought of as a variable resistor, because the resistance of each of diodes 211-214 changes as the current through the diodes changes.

[0061] A portion of circuit 200 is shown in FIG. 6.

Variable resistor 290 in FIG. 6 represents the Thevenin equivalent variable resistance of diodes 211-214 for purposes of illustration. The resistance of variable resistor 290 equals the resistance of any one of diodes 211-214 for a given current through each of diodes 211- 214. Resistors 208,290, and 108 form a voltage divider between VIN and Vc as shown in FIG. 6. Ohm's law may be applied to calculate the voltage drop across variable resistor 290 as well as the voltage drop across resistors 208 and 108. Voltage VIN is a divided down voltage from VG. Therefore, voltage Vs at the output of amplifier 103 can be calculated using the resistive ratio of resistors 208,290, and 108 as shown in the following equation: <BR> <BR> <BR> <BR> Rzos + R9o + Rios<BR> <BR> <BR> vu-yin R290 + R108 Equation (11) where R208 is the resistance of resistor 208, R290 is the resistance of variable resistor 290 for a given current through each of diodes 211-214, and Rice is the resistance of resistor 108. The gain G of amplifier 103 varies as the resistance of resistor 290 and the impedance of diodes 211-214 varies. Substituting equation (11) into

equation (7), the gain G of amplifier 103 may be expressed as: G = R208 + R290 + R108 R290+ R108 Equation (12) [0062] The resistance of variable resistor 290 equals the resistance of diodes 211-214. The initial resistance of each of diodes 211-214 can be calculated before the start of an ion detection cycle when V1, V2, and V3 all equal zero volts by measuring the gain of amplifier 103.

For example, if the gain of amplifier 103 is 2 before t3, resistor 108 equals 1.0 Q, and resistor 208 equals 698 Q, then the resistance of each of diodes 211-214 is about 697 Q, according to equation (12). The current through diodes 211-214 equals about 55 uA according to equation (10) if n is assumed to be 1.5. A minimum gain in amplifier 103 of 2 is required to maintain a minimum gain of 1 from Vin to Vout, since the impedance matching resistor 108 causes an attenuation of the signal from Vg to Vout to be one-half. In systems where the gain from Vin to Vout can be less than 1, it is evident to those skilled in the art that the minimum gain setting resistors 220 and 221 can be increased indefinitely to lower the initial gain from Vin to Vg to a limit of 1, and therefore the gain from Vin to Vout to one-half, when resistors 220 and 221 are raised to infinity. Such an approach will increase the bandwidth and also increase the dynamic range at the expense of initial gain.

[0063] As V2 increases, the voltage at node 291 increases. As V3 decreases, the voltage at node 292 decreases. As the voltage differential from node 291 to node 292 across diodes 211-214 increases, the current

through diodes 211-214 increases, lowering their impedances according to equation (10).

[0064] Resistors 220 and 110 can be reduced to their Thevenin equivalent. For example, if resistor 220 equals 70 kQ and resistor 110 equals 1 kQ, then the Thevenin equivalent resistance is 986 Q. Resistors 221 and 111 can also be reduced to their Thevenin equivalent. For example, if resistor 221 equals 70 kQ and resistor 111 equals 1 kn, then the Thevenin equivalent resistance is also 986 Q. The voltages at nodes 291 and 292 may be calculated using these equivalent resistor values. For example, if Vs2+ equals 15 volts and VS2-equals-15 volts, then the voltage at node 291 would be 1.2 volts if diode 213 were not attached to this node, and the voltage at node 292 would be-1.2 volts if diode 214 were not attached to this node when Vs reaches +1 volt and V3 reaches-1 volt.

[0065] Assuming that Io in equation (8) equals 3.3 uA, the voltage across diodes 211-214 can be calculated at approximately 192 mV using an iterative approach. The voltage across the 986 Q equivalent resistor then becomes 1.008 volts, and the current through the equivalent resistor is 1.02 mA. The current through each of diodes 211-214 is half of this current (i. e., 510 uA) using Kirchhoff's current law and assuming diodes 211-214 have the same impedance. Diodes 211-214 have the same impedance when VIN equals zero. The impedance of diodes 211-214 (and resistor 290) can be calculated to be about 75 Q using equation (10) for n = 1.5. Therefore, the gain G of amplifier is about 10 at this time, as can be seen from equation (12).

[0066] Signal V2 increases to a maximum value, and signal V3 decreases to a minimum value as determined by

resistor 224, resistor 226, and diode 228 (FIG. 4). For example, if the resistor 224 is 4. 87 kQ, resistor 226 is 3.32 kQ, Vs2+ is 15 volts, VS2-is-15 volts, then resistor 224, resistor 226, and diode 228 limit the maximum value of V2 to +11.2 volts, and the minimum value of V3 to-11.2 volts. These component values are chosen as an illustration, and other component values may be used instead. Using these example values and the Thevenin equivalent of resistors 220/110 and 211/111 (e. g., 986 Q), the voltages at nodes 291 and 292 are +/-11.25 volts, respectively, when V2 reaches its maximum value and V3 is at its minimum value. The current through the Thevenin equivalent of resistors 220/110 is 11.4 mA, and the current through diodes 211-214 is 5.7 mA.

[0067] Using equation (10), the impedance of diodes 211-214 is theoretically about 6.7 Q (for n = 1.5).

However, testing may reveal that the gain G of amplifier 103 is 55 at this point. Therefore, the impedance of diodes 211-214 is actually about 13 Q (assuming resistor 208 is 698 Q and resistor 108 is 1 Q). Subtracting out the resistance of resistor 108 which is small (e. g., 1 Q), reveals that there is some extra resistance in diodes 211-214 that is ignored by equations (10) and (12). This extra resistance is due to the resistance of the diode crystal, which is about 5.3 Q using these exemplary component values. This resistance becomes significant for higher gain values.

[0068] It should be noted that diodes are readily available that have much less resistance than 5.3 Q.

However, the equivalent circuit for a diode includes a parallel capacitor. Diodes that have a smaller resistive component also have a large capacitive component, which makes them unsuitable for applications in the frequency

ranges associated with mass spectrometry. The resistance of the diode crystal and resistor 108 together provide an upper limit to the gain G of amplifier 103, as can be seen from equation (12). The gain G of amplification circuit 200 is proportional to the square of the ion detection time.

[0069] When an input signal appears at the non- inverting input of amplifier 103 (i. e., VIN varies from 0 volts), amplifier 103 drives its output voltage VG until the voltage at the inverting input of amplifier 103 equals VIN. For example, if VIN is-1. 0 mV, the current through diodes 211 and 214 increases slightly, lowering the impedance of these diodes. Because the current through resistors 220 and 110 is constant (for given values of V2 and V3), an equal amount of current decreases through diodes 213 and 212, causing their impedances to increase. However, as long as VIN is small, the Thevenin equivalent impedance represented by variable resistor 290 remains essentially constant. Amplifier 103 drives VG according to the relationship shown in equation (11) until the voltage at its inverting input equals VIN.

Voltage Vous vis a divided down voltage from Vs that is determined by the resistor ratio of resistor 104 (FIG. 4) and the resistance of the load coupled to VouT. Resistor 104 is selected to match the impedance of the cable connecting VOUT to the load.

[0070] If VIN is large enough, the impedance of resistor 290 increases (for given values of V2 and V3), which is advantageous because it reduces the gain of large peaks on VIN, therefore keeping an analog-to-digital converter that is part of data acquisition device 80 within its input voltage range. If VIN exceeds the voltage drop across diodes 211-214 when VIN equals zero,

diode 213 is reverse-biased, and the gain G of amplifier 103 is reduced (e. g., to 1.7). This generally only occurs on signals early in the parabolic voltage ramps, when the gain is relatively low, and VIN is relatively large.

[0071] Because of the variations in parameters in the electronic components from which circuits are built, reflected in their tolerances, several components can usefully be added to improve the functional operation of the entire circuit. Some examples are as follows: optional resistors may be placed in parallel with resistors 220 or 221 to match them precisely, resistors may be placed in parallel with resistors 110 or 111 to compensate for dynamic differences in impedance of the four diodes in bridge 107, capacitors may be placed in parallel with capacitors 236 or 242 to match the two parabolas due to minor differences of the input capacitances of amplifiers 234 or 240, or resistors may be placed from a power supply to the two inputs of amplifier 103 to compensate for input bias current or input offset voltage.

[0072] One embodiment of the present invention includes a constant bandwidth, current mode amplifier.

If amplifier 103 in FIG. 3 is a current mode amplifier, then its bandwidth is constant as its gain varies. The gain G of amplifier 103 increases as ions with larger molecular weights strike the ion detector. Amplifier 103 increases the amplification of ions with larger molecular weights, providing an amplified signal for these ions with respect to previously known techniques discussed above.

[0073] Furthermore, the constant bandwidth, variable gain amplifier would demonstrate great utility in TOFMS

applications for which frequency response needs to be conserved when increasing gain amplification is required.

Specific examples of the latter are amplification needs for detectors in a reflectron, time lag focusing (TLF) instruments, and orthogonal extraction instruments with molecular weight ranges limited to less than 20 kDa.

[0074] Another desirable embodiment of the present invention includes a voltage mode amplifier 103 that has a variable bandwidth that is inversely proportional to the gain G of amplifier 103. Amplifier 103 in FIGS. 3 and 4 may be a voltage mode amplifier in which the - product of its gain times its bandwidth is a constant (e. g., 1200 MHz). Thus, the bandwidth of a voltage mode amplifier decreases as the impedance of diodes 211-214 decrease (and the gain G of amplifier 103 increases). At the beginning of an ion detection cycle, the gain G of amplifier 103 is small and the bandwidth of amplifier 103 is high so that the high frequency components of smaller molecular weight ions can be detected when these ions strike the ion detector. As parabolic signal V2 ramps up and parabolic signal V3 ramps down, the gain G of amplifier 103 increases (as the current through diodes 211-214 increases) and the bandwidth of amplifier 103 decreases, improving the detection signal-to-noise ratio for larger molecular weight ions, while preserving sensitivity and mass resolving power for the detection of low molecular weight ions.

[0075] The increased gain allows for further amplification of the low frequency detection signals that are characteristic of large molecular weight ions, which otherwise possess poor detection efficiency. The reduced bandwidth filters out high frequency noise while detecting larger molecular weight ions. The reduced

bandwidth along with the increased gain provides an even greater signal-to-noise ratio for detection signals of ions within a broad range of molecular weights. In addition, the vertical dynamic range of amplification circuit 200 is not compromised when detecting low molecular weight ions, because amplification of unnecessary high frequency components is avoided.

[0076] A graph illustrating variable bandwidth curves in an embodiment of the present invention that includes a voltage mode amplifier is shown in FIG. 7A. The bandwidth of the amplification system with a voltage mode amplifier decreases over time as ions with larger molecular weights strike the ion detector. During time of flight mass spectrometry, the amplitude of input signals VIN for a given ion population may vary over different ion detection cycles, due to irregularities in the ionization process, changes in micro-environmental conditions and positional differences of the sample for laser based ionization schemes. Therefore, the bandwidth of the amplification system is preferably independent of input signal amplitude to increase the accuracy of the output signal. Bandwidth curve 300 in FIG. 7A is representative of an input voltage signal VIN of 20 mV; bandwidth curve 301 is representative of an input voltage signal VIN of 50 mV; and bandwidth curve 302 is representative of an input voltage signal VIN of 100 mV.

Thus, the curves of FIG. 7A demonstrate that the bandwidth of the variable gain amplification system of the present invention is substantially the same for input voltage signals with varying amplitudes.

[0077] A graph illustrating variable gain curves in accordance with the principles of the present invention is shown in FIG. 7B. The gain of the amplification

system of the present invention increases over time as larger molecular weight ions impact the ion detector.

The gain (expressed in decibels in FIG. 7B) increases in proportion to the square of the ion detection time.

Curve 400 of FIG. 7B is representative of an input voltage signal of 20 mV; curve 401 is representative of an input voltage signal of 50 mV; and curve 402 is representative of an input voltage signal of 100 mV. At input signals of 100 mV, slight saturation of the amplifier may occur limiting the maximum gain (e. g., at about 22 dB).

[0078] It is often the case that two or more ion detection signals are in close proximity to or overlap each other in terms of frequency. Such frequency overlap may be created by poorly resolved isotopic and isobaric ionic species. The resultant signal could be composed of lower than usual frequency components as the two overlapping signals constructively or destructively combine during their respective periods. When this occurs, this group of lower frequency signals may be excessively amplified by the variable gain amplification system of the present invention. To avoid this situation, the onset of the amplifier gain ramp at time t3 is delayed a period of time after time t0 until a time at which amplifier bandwidth does not result in significant distortion of poorly resolved ion populations. This delay period (from t0 to t3) is dependent upon ion total kinetic energy and ion free flight distance as well as the fundamental bandwidth limitation of the detection apparatus, and may be varied by the operator of the mass spectrometer to optimize different embodiments.

[0079] Referring again to FIG. 5, the time lapse between times t3 and t4 is contemporaneous with the duration of time that ions are impacting ion detector 76.

If desired, VCL may remain LOW for a period of time after the last ion in an ion detection cycle strikes ion detector 76. The time lapse between t3 and t4 may be, for example, 1500 usec. During the period of time between t1 and t4, switch 271 in FIG. 4 is open. At time t4, VCL goes HIGH, and switch 271 closes causing pre-ramp clamp loop circuit 106 to close around feedback network 105 and variable impedance circuit 107. Amplifier 280 in clamp circuit 106 drives VG to a desired starting voltage value (e. g., near zero volts) before the beginning of the next ion detection cycle. Pre-ramp clamp circuit 106 removes any undesirable offset voltages in the feedback loop around amplifier 103 between times tl and t4.

[0080] When VCL transitions from LOW to HIGH at time t4 and switch 271 closes, current begins to flow through resistors 270 and 272 until the voltage at the non- inverting input of amplifier 280 equals Vs. The DC voltage at the inverting input of amplifier 280 is determined by a voltage divider comprising resistors 275 and 278. For example, if resistor 275 is 49.9 Q, resistor 278 is 69.8 kQ, and Vsl-is-5 volts, then the voltage at the inverting input of amplifier 280 is - 3. 57 mV. These component value are illustrative, and other component values may be used instead.

[0081] Amplifier 280 drives its output voltage V4 until the voltages at its inverting and non-inverting inputs are equal to each other. Amplifier 280 controls a current through resistors 222 and 108 by controlling its output voltage V4. The current through resistors 222 and 108 determines the voltage at node 282. The voltages at

VIN and node 282 determine the voltage at VG. Amplifier 280 adjusts the current through resistors 222 and 108 until Vs equals the voltage at the inverting input of amplifier 280. For example, if the voltage at the inverting input of amplifier 280 equals-3.57 mV, VIN is 0 volts, resistor 208 is 698 Q, resistor 290 is 13 Q, resistor 222 is 3.01 kQ, and resistor 108 is 1 Q, then the voltage at node 282 is driven to +66.5 pV by amplifier 280, VG becomes-3.57 mV, and V4 becomes approximately 3.24 volts.

[0082] After VG has risen to the voltage level at the inverting input of amplifier 280, the next ion detection cycle may begin. The signals shown in FIG. 5 are periodic signals that repeat for each ion detection cycle. Sometime after t4, VCL, VRS, and VRP change state again in the same direction as they did at time tl, and a new ion detection cycle begins.

[0083] The initial and final bandwidth requirements of both the variable gain constant bandwidth and variable gain fixed gain-bandwidth product amplifiers are determined by the design goals of the TOFMS as well as its fundamental geometry. Parallel extraction devices typically employ time array recording instruments to digitize analog data. These types of devices operate in an analog mode and produce a digital replica of input analog signals.

[0084] Input signal profiles should be preserved for faithful digital reproduction. For linear parallel extraction devices employing resolution enhancing technology such as time lag focusing, initial amplifier bandwidth should be broad to preserve the narrow peak widths seen during the detection of low molecular weight ions. For parallel extraction devices employing time lag

focusing and electrostatic ion mirrors, low molecular weight ions peak widths may be even narrower than corresponding peak widths of linear parallel extraction devices employing resolution enhancing technology.

Amplifier bandwidth should be even broader to accurately reproduce low molecular weight ion detection signals without spurious signal broadening. For these two previous examples, it is desirable to provide maximum amplifier bandwidth between 300 and 2000 MHz. During the detection of high molecular weight ions in a time lag focusing, linear, parallel extraction device, inherent temporal width of these ion signals are such that often bandwidth extending from 100 kHz to 20 MHz is sufficient to preserve inherent peak shape while concomitantly eliminating unwanted high frequency noise.

[0085] Parallel extraction, time lag focusing, electrostatic mirror geometries are inefficient in detecting ions of high molecular weight. For this reason, such devices are often toggled between linear and reflectron operational modes. In some cases, each operational mode creates a distinct ion transit path often requiring the use of separate detectors for linear and reflectron operation. Under such conditions, each detector could be fitted with an individual variable gain amplifier, each with bandwidth characteristics well suited for their intended ion detection populations.

Specifically, the linear ion detector variable gain amplifier would operate in a fashion identical to that of a variable gain amplifier in a fixed linear system, while the reflectron ion detector variable gain amplifier would have a higher initial bandwidth and would have a different gain ramp function, because this ion detector is rarely used to detect ions of m/z greater than 10,000.

[0086] Orthogonal extraction devices, on the other hand, often make use of time interval recording devices that function to create digital signals in a pulse counting mode. In a pulse counting mode, a given impulse is recorded each time an analog signal exceeds a given threshold. Within limits, the rate at which the threshold is achieved can be somewhat diminished compared to the temporal width of the original input signal while still accurately creating digital signals without temporal broadening. Under these circumstances, the bandwidth requirements for the detection of low molecular weight ions are less than that for a parallel extraction system employing time-array recording devices. Typical bandwidth used for orthogonal extraction platforms during the detection of low molecular weight ions is on the order of 100-700 MHz. During the detection of high molecular weight ions, bandwidth requirements of the orthogonal geometry closely mirror that of parallel extraction platforms.

[0087] Referring now to FIG. 8, example mass spectra for a prior art mass spectrometer and a mass spectrometer with the variable gain amplification system of the present invention are shown. The trace labeled"Fixed Gain"is a signal indicating a mass spectrum created by a TOFMS with a fixed gain, fixed bandwidth amplifier. The trace labeled"Variable Gain"is a signal indicating a mass spectrum for a TOFMS with a fixed gain-bandwidth product (voltage mode) amplifier of the present invention. The x-axis is in units of Daltons (Da). The mass scale (x-axis) for the"Variable Gain"trace has been offset by +5% with respect to the"Fixed Gain"trace for clarity purposes. Each trace has been vertically

normalized with respect to each other for the protein signal generated at 16,950 Da (Equine Cardiac Mb).

[0088] The effective difference in signal to noise between these two plots is demonstrated. The detection signal created by the variable gain amplifier demonstrates a more vertical amplitude for the various proteins detected at 16,950 Da, 29,024 Da, (Bovine Red Blood Cell Carbonic Anhydrase II), and 66,430 Da (Bovine Serum Albumen). Additionally, the bandwidth restriction applied by the fixed gain-bandwidth product amplifier of the present invention creates a noticeable reduction in high frequency noise without introducing undue signal broadening and associated diminution of mass resolving power. This example demonstrates the superior performance attainable by the use of a fixed gain- bandwidth product amplifier for time of flight mass analysis when compared to a fixed gain, fixed bandwidth amplifier of the prior art.

[0089] Persons skilled in the art further will recognize that the present invention may be implemented using structures and process steps other than those shown and discussed above. All such modifications are within the scope of the present invention, which is limited only by the claims which follow.