Field of the Invention

This invention relates to an ultrawideband isolating transformer, which uses microwave junctions to interconnect its windings and external ports. The transformer, particularly though not exclusively, is an isolating transmission line transformer (TLT) for use within a data communications circuit or system, and microwave devices and equipment. The invention also relates to a method of constructing such ultrawideband isolating transformer.

Background of the Invention

Data communications and measurement equipment is often required to couple broadband signals to and from transmission lines with some D.C. and low frequency isolation, e.g. to reject common mode signals such as mains hum in ‘earth loops’. A D.C. isolating transformer is commonly employed for this purpose.

It is generally accepted, however, that the parasitic reactance of such known transformers will limit the upper usable frequency (fU) that may be communicated over the transmission line by introducing loss and mismatch. Further, the lower frequency limit (fL) will be limited by a shunt reactance to make it difficult to increase the ratio fU/fL beyond a certain limit, typically 100,000. There is therefore placed a limitation on the achievable overall bandwidth.

Currently, only conventional isolating transformers are used in local and wide-area networks (LANs and WANs) and, in their current form, by virtue of the above characteristics, these limit bandwidth and are therefore not conducive to optimising the potential benefits of high speed networks, fibre optic backbones and networks, for example.

Another form of transformer is a Transmission line Transformer (TLT) in which the physical properties of the conductors used for the transformer windings are considered and disposed in such a way as to also form part of a transmission line.

GB2556359 discloses such an isolating Transmission line Transformer (TLT) with good properties.

We improve on these TLTs and present designs of isolating transformers as a composition of three components, two microwave junctions, and the windings. Scattering characteristics for these components that enable the device to transport travelling pulses with durations much less than the transmission time between ports. Such a device is extremely wideband, with high frequency differential mode characteristics like those of a segment of differential transmission line of half the length used in the windings and low frequency characteristics of a large shunt inductance. These devices can have a much greater ratio of high-frequency cut-off to low-frequency cut-off for the pass band than any others known to the author at this time.

Summary of the Invention

In a broad sense, there is provided an ultrawideband Isolating Transmission Line Transformer (ITLT) for use in data communications, the ITLT being arranged with first and second ports connected to respective first and second windings, the ports being D.C. isolated from one another.

According to a first aspect of the invention, there is provided an isolating transformer for use in a data communications system, the transformer comprising: a first port formed of two separate first terminals, with respective first terminal conductors, located at or close to a first edge, comprising outer and inner port sections; a second port formed of two separate second terminals, with respective second terminal conductors, located at or close to a second edge, comprising outer and inner port sections; first and second conductive winding paths connected in series to the first and second ports respectively, said paths being electrically isolated from one another; a first microwave junction formed at the first port comprising a first predefined uniform gap, along a predefined length section within the first port, between each of the first terminal conductors and the second conductive path; a second predefined uniform gap between end sections of the first terminal conductors; and a third predefined uniform gap at a section where the first terminal conductors transition from the predefined length section into a first terminal end section.

According to a second aspect of the invention, there is provided a method of providing DC isolation in a data communications system, the method comprising connecting the isolating transformer above with one port connected to a computer, computer modem, or data communications equipment, and the other port connected to a transmission line or the like, and in which the data communications system is configured to transmit and/or receive data to and/or from the further transmissions line. According to a third aspect of the invention, there is provided a method of manufacture of an isolating transformer, the method comprising: providing a first port formed of two separate first terminals, with respective first terminal conductors, located at or close to a first edge, comprising outer and inner port sections; a second port formed of two separate second terminals, with respective second terminal conductors, located at or close to a second edge, comprising outer and inner port sections; first and second conductive winding paths connected in series to the first and second ports respectively, said paths being electrically isolated from one another; a first microwave junction formed at the first port comprising a first predefined uniform gap, along a predefined length section within the first port, between each of the first terminal conductors and the second conductive path; a second predefined uniform gap between end sections of the first terminal conductors; and a third predefined uniform gap at a section where the first terminal conductors transition from the predefined length section into a first terminal end section.

Preferred aspects are defined in the dependent claims.

Brief Description of the Drawings

The invention will now be described, by way of non-limiting example, with reference to the accompanying drawings, in which:

Figures 1a and 1b show schematic illustrations of an isolating transformer, as currently known in the prior art, with no centre-tap and which is a two-port device with two terminals at each port, i.e. a four terminal device;

Figures 2 shows a schematic illustration of the isolating transformer with improved junctions 1 and 2, in perspective view with reference planes, in accordance with the new invention;

Figures 3a show a block schematic depicting the isolating transformer, in accordance with the new invention, with no centre taps as a single-ended four-port device. The dotted line represents the boundary of the transformer;

Figure 3b shows a block schematic of the same transformer as described in Figure 3a, but shown in its equivalent mixed-mode representation;

Figure 4a shows a schematic depicting the internal connectivity of an isolating transformer such as are depicted in the block schematics Figure 3a and 3b; it is a decomposed into windings and two multimode two-port microwave junctions; the solid lines represent conductors internal to the shaded blocks, and the dotted lines show the location of reference planes;

Figure 4b shows a schematic depicting the internal connectivity of an isolating transformer with two centre taps; it differs from Figure 4a only by the addition of these centre taps;

Figure 5a shows a schematic depicting the internal connectivity of an isolating transformer with no centre taps; illustrating alternative disposition of reference planes to that shown in Figure 4a.

Figure 5b shows a schematic depicting the internal connectivity of isolating transformer with two centre taps; it differs from Figure 5a only by the addition of these two centre taps, and the additional reference plane required as a result; illustrating alternative disposition of reference planes to that shown in Figure 4b;

Figure 6a shows an illustration of an embodiment of a Junction, with terminals on the same plane and abut a winding, in accordance with the invention; the rectangles indicate waveguide boundaries on the reference planes for the purposes of simulation and analysis;

Figure 6b shows an illustration of a further embodiment of a Junction, with terminals on the different planes and sandwiching the winding between them, in accordance with the invention;

Figure 7 shows an illustration of a Junction in accordance with this invention that has a centre tap;

Figure 8 shows an illustration of the same embodiment as depicted in Figure 6a, but with an alternative disposition of reference planes (as illustrated by the alternatively placed waveguide port boundaries);

Figure 9 shows an illustration of an alternative placement of waveguides for the embodiment shown in Figure 7;

Figure 10a depicts graphical results of a transformer with 10-turns according to this invention; the curves show the differential scattering parameters in the frequency domain. The x-axis represents the frequency in Hz, and the y-axis the scattering parameter values in decibels (dB). The insertion loss (IL) between ports is the differential scattering parameters Sdd21 and Sdd12 (one for each direction); the differential return loss (RL) is the scattering parameters Sdd11 and Sdd22, respectively for port 1 and port 2;

Figure 10b depicts graphical results of a transformer similar to that one of Figure 10a but the gap between the conductors in the Junction were prevented from opening;

Figure 11a depicts graphical results for a 34 -turn device otherwise similar to the 10-turn whose results are depicted in 10b;

Figure 11b depicts graphical results of a transformer nominally identical to that presented in Figure 11a, but with degraded Junctions;

Figure 12a depicts graphical results derived from the same differential scattering parameters as Figure 10a, but depicted as the step response in the time-domain;

Figure 12b depicts graphical results derived from the same scattering-parameters as Figure 10b;

Figure 13a depicts graphical results derived from the same s-parameters as Figure 11a, and depict the step response in the time-domain; this response is that of the 34-turn device with the better Junction realisation;

Figure 13b depicts graphical results is derived from the same s-parameters as Figure 11b, and depict the step response in the time-domain; this response is that of the 34-turn device with the poorer Junction realisation;

Figure 14 is a single loop model used for comparison;

Figure 15a and 15b illustrate a multi-turn transformer embodiment in accordance with the invention, in 8-turn and 52-turn forms, respectively; the topology is that of a helix wrapped around a toroid;

Figure 16 illustrates a solenoidal transformer with 9 turns;

Figure 17 shows the simulation result for a 2-turn helically wound toroidal transformer compared with a circular single turn transformer with the same length of winding; Figure 18 illustrates the S21 (IL) for a 24-turn helically wound toroidal transformer compared with a circular single turn transformer with the same length of winding;

Figure 19 shows the resonant frequency of a toroidal device is consistently higher than that of a single turn device;

Figure 20 shows the bandwidth of a toroidal helix improves with the number of turns in the helix, though with diminishing returns above 20 turns or so;

Figure 21 depicts a normalised measure of bandwidth of the transformers versus the number of turns formed in the windings; this is the number of cycles of a sinusoidal signal distributed around the winding at the fundamental resonance (i.e. at the lowest frequency of resonance) on the y-axis (vertical-axis) for a given number of turns of winding on the x-axis (horizontal axis); and

Figure 22 illustrates the bandwidth of a solenoidal transformer for a given length of winding.

Description of the Invention

This invention relates to an ultrawideband isolating transformer, which uses microwave junctions to interconnect its windings and external ports. The transformer, particularly though not exclusively, is an isolating transmission line transformer (TLT) for use within a data communications circuit or system, and microwave devices and equipment. The invention also relates to a method of constructing such ultrawideband isolating transformer.

Broadband signals travel as waves through the two ports of an isolating transformer such as is shown in Figure 1a and Figure 1b. These type of isolating transformers are described in detail in the applicant’s prior patent publication GB2556359B.

Relationships amongst these waves can be measured with a vector network analyser (VNA) from low to very high frequencies with excellent accuracy. This requires the definition of a reference plane (such as indicated by the dashed lines in Figure 1a and 1b) through a short section of uniform transmission line that guides the travelling waves at each port. The VNA presents the scattering parameters (s-parameters) that present the relationships amongst these waves traveling though these reference planes that completely characterises the small signal properties of the device. The internal structure of a transmission line transformer, by design, also guide travelling waves, and reference planes can be defined through its internal structure, dividing the device into microwave interconnected components that can be individually characterised and analysed to describe their relationships to the microwave characteristics of the device as a whole. Figure 1b, shows such a decomposition of a transmission line transformer. This decomposition is used to analyse characteristics required of a mixed-mode two-port that consists of two mixed-mode three ports interconnecting its windings with its external ports.

We present designs of isolating transformers as a composition of three components, two microwave junctions, and the windings. Scattering characteristics for these components that enable the device to transport travelling pulses with durations much less than the transmission time between ports. Such a device is extremely wideband, with high frequency differential mode characteristics like those of a segment of differential transmission line of half the length used in the windings and low frequency characteristics of a large shunt inductance. These devices can have a much greater ratio of high-frequency cut-off to low- frequency cut-off for the pass band than any others known to the author at this time.

Figure 1a and 1b show an isolating transformer from the author’s prior patent publication that can be divided into three sections, as described above for analysis.

In more detail, Figures 1a and 1b, show a transformer divided by vertical reference planes through the device along the innermost dotted lines (labelled Port A and Port B) into three components (Junction 1, Windings, and Junction 2). The junctions serve to interconnect the Windings with each of the two external ports (Port 1 and Port 2), which are situated on a vertical plane through the outermost dotted lines. We do not change the properties of the winding from that in the prior art (which consist of differential transmission lines of substantially constant characteristic impedance, Zo/2, and a delay of Td) and present the same external characteristics to the device at Port 1 and Port 2, a substantially constant characteristic differential impedance of Zo (twice that of the winding).

The reference planes in Figure 1b each cut the device in a plane normal to a section of uniform transmission line so that we can define, analyse, and in principle, measure waves travelling through those planes. These waves can be measured using a vector network analyser by embedding the component in a test fixture that can be calibrated.

Ports 1 and 2 each have two conductors (in addition to any surrounding screening conductor), and thus support two principal modes for the electromagnetic fields of the waves travelling in each direction. These two modes can be, for example, an even mode, and an odd mode, with the total field of a travelling wave an admixture of these two components. The odd mode of a two-conductor (bifilar or twin) transmission line is often referred to as the differential mode, and is in general the mode intended to be excited and used to transport travelling waves. The even mode is often called the common mode, and in contrast it is often undesirable to excite the common mode (which when unscreened can lead to radiation and susceptibility to interference). Indeed, isolating transformers are often employed to suppress common mode signals. Ports A and B of Figure 1b each have four conductors (over and above any for a surrounding screen or ground reference), and so support four modes.

In more details and in reference to the drawings we describe below:

Figures 1a and 1b show schematic illustrations of an isolating transformer, as currently known in the prior art, in a top view and perspective view, respectively. The illustrations show the isolating transformer with reference planes. The dashed lines within the structure indicate the location of reference planes through the structure that serve to divide it into three components. Two components are designated as microwave junctions, each interconnecting an external port with one end of the winding.

The figures shows a schematic illustration of an isolating transformer, as currently known in the prior art, with no centre-tap is a two-port device with two terminals at each port, i.e. it is a four terminal device. The dashed lines indicate the location of a reference plane through the short section of transmission line at each port.

In more detail Figures 1a and 1b, show a transformer 10, which comprises a binocular (or bead) core 11 with two parallel bores 14, 15 through which twisted conductors 13, 16 pass to provide a transmission line. The core can actually be toroidal, binocular or a pot.

A first port (Port 1) is provided to one side of the core 11 , and comprises a first conductor 13 which runs from one port terminal, through the first bore 14, whereafter it exits and returns back through the second bore 15 and terminates at the other port terminal. A second port (Port 2) is provided on the mechanically opposite side to the core 11, and comprises a second conductor 16 which runs from one port terminal, through the second bore, whereafter it exits and returns back through the first bore 14 and terminates at the other port terminal. The conductors 13, 16 therefore execute a single turn or winding, as with the previous embodiment, which was found to exhibit particularly advantageous results for a binocular core. Conductors 13 and 16 are twisted together within the core 11 as shown, but are insulated from one another by surrounding insulating material and have a substantially constant gap.

The reference planes in Figure 2 each cut the device in a plane normal to a section of uniform transmission line so that we can define, analyse, and in principle, measure waves travelling through those planes. These waves can be measured using a vector network analyser by embedding the component in a test fixture that can be calibrated.

Ports 1 and 2 each have two conductors (in addition to any surrounding screening conductor), and thus support two principal modes for the electromagnetic fields of the waves travelling in each direction. These two modes can be, for example, an even mode, and an odd mode, with the total field of a travelling wave an admixture of these two components. The odd mode of a two-conductor (bifilar or twin) transmission line is often referred to as the differential mode, and is in general the mode intended to be excited and used to transport travelling waves. The even mode is often called the common mode, and in contrast it is often undesirable to excite the common mode (which when unscreened can lead to radiation and susceptibility to interference). Indeed, isolating transformers are often employed to suppress common mode signals. Ports A and B of Figure 2 each have four conductors (over and above any for a surrounding screen or ground reference), and so support four modes.

Figure 2 shows a schematic illustration of the isolating transformer 20 with improved junctions 1 and 2, in perspective view with reference planes, in accordance with the new invention. The junctions are modified so that when the device is excited by wholly balanced signals at its external ports (1 and 2), waves reflected back from Port A or B to the windings are exactly cancelled by an opposing wave transmitted from the other half of the windings.

Figure 2 shows an improved isolating transformer over the one shown in Figure 2. The improvement is in junctions 1 and 2, and can be applied to planar or different topology transformers.

In more detail Figures 2, shows an improved transformer 20, which comprises a binocular (or bead) core 21 with two parallel bores 24, 25 through which twisted conductors 23, 26 pass to provide a transmission line. The core can actually be toroidal, binocular, pot or any type of transformer core. The transformer can be planar or non-planar, with or without twisted conductor, and with or without center taps.

One of many examples is shown below, purely for illustration purposes. A first port (Port 1) is provided to one side of the core 21 , and comprises a first conductor 23 which runs from one port terminal, through the first bore 24, whereafter it exits and returns back through the second bore 25 and terminates at the other port terminal. A second port (Port 2) is provided on the mechanically opposite side to the core 21, and comprises a second conductor 26 which runs from one port terminal, through the second bore, whereafter it exits and returns back through the first bore 24 and terminates at the other port terminal. The conductors 23, 26 therefore execute a single turn or winding, as with the previous embodiment, which is found to exhibit particularly advantageous results. Conductors 23 and 26 are twisted together within the core 21 as shown, but are insulated from one another by surrounding insulating material and have a substantially constant gap. As previously mentioned, the conductors do not need to be twisted and the core can be of any type. A first microwave junction 28 formed at the first port, comprising predefined and uniform gaps or uniform spacings d1 and d1 *, along a predefined winding length section, sections between Port 1 and Port A, between each of the first terminal conductors and the second conductive path; predefined and uniform gap d2 between a length of port end sections of the first terminal conductors; and predefined gap d3 at a section wherein the first terminal conductors transition from the winding section into a first terminal port end section.

A second microwave junction 29 formed at the second port comprising predefined and uniform gaps d4 and d4*, along a predefined winding length section, sections between Port 2 and Port B, between each of the second terminal conductors and the first conductive path; predefined and uniform gap d5 between a length of port end sections of the second terminal conductors; and predefined gap d6 at a section wherein the second terminal conductors transition from the winding section into a second terminal port end section.

Referring now to Figure 3a, it is shown a block schematic depicting the isolating transformer with no centre taps as a single-ended four-port device. The dotted line represents the boundary of the transformer. Its four ports, labelled p1-4 just inside this boundary, can each simultaneously guide single-ended signals (i.e. a single mode of travelling wave) in both the inward (a) and outward (b) directions. The schematic shows the transformer is composed of three elements in cascade: single-ended six-port, Junction 1, to the left, a single-ended eight-port, the Windings, in the middle, and another single-ended six port element to the right, Junction 2. Figure 3a is a nodal description that is the counter part of the modal description of Figure 3b. There are well defined relationships between these descriptions. The single-ended description is more directly suited to the measurement of these devices using a VNA (which commonly has single-ended physical ports that are mathematically related to virtual mixed-mode ports). The single-ended description is also a useful alternative to analyse the relationships amongst the travelling waves in the device.

Figure 4a shows a schematic depicting the internal connectivity of an isolating transformer such as are depicted in the block schematics Figure 3a and 3b. It is a decomposed into windings and two multimode two-port microwave junctions. The solid lines represent conductors internal to the shaded blocks, and the dotted lines show the location of reference planes.

Figure 4b shows a schematic depicting the internal connectivity of an isolating transformer with two centre taps. It differs from Figure 4a only by the addition of these centre taps.

Figure 5a shows a schematic depicting the internal connectivity of an isolating transformer with no centre taps. It serves to illustrate alternative disposition of reference planes to that shown in Figure 4a.

Figure 5b shows a schematic depicting the internal connectivity of isolating transformer with two centre taps. It differs from Figure 5a only by the addition of these two centre taps, and the additional reference plane required as a result. It serves to illustrate alternative disposition of reference planes to that shown in Figure 4b.

Figure 6a shows an illustration of an embodiment of a Junction, with terminals on the same plane and abutting a winding, in accordance with the invention. The rectangles indicate waveguide boundaries on the reference planes for the purposes of simulation and analysis, showing a 2-port microwave junction in 3D of the representation of Figure 4a.

Figure 6b shows an illustration of a further embodiment of a Junction, with terminals on the different planes abutting and sandwiching the winding between them, in accordance with the invention, again showing a 2-port microwave junction further embodiment in 3D of the representation of Figure 4a. This junction is a further improvement on the one shown in Figure 6a. There is better balance between reflected and transmitted waves circulating around the loop formed by the Windings and Junctions.

Figure 7 shows an illustration of a Junction in accordance with this invention and that has a centre tap, showing a 2-port microwave junction with centre tap in 3D of the representation of Figure 4b. Figure 8 shows an illustration of the same embodiment as depicted in Figure 6a, but with an alternative disposition of reference planes (as illustrated by the alternatively placed shaded waveguide port boundaries), showing a 3-port microwave junction in 3D of the representation of Figure 5a.

Figure 9 shows an illustration of an alternative placement of waveguides for the embodiment shown in Figure 7, showing a 3-port microwave junction with centre tap in 3D of the representation of Figure 5b.

Thus the invention describes an isolating transformer for use in a data communications system, the transformer comprising: a first port formed of two separate first terminals, with respective first terminal conductors, located at or close to a first edge; a second port formed of two separate second terminals, with respective second terminal conductors, located at or close to a second edge; first and second conductive paths connected in series to the first and second ports respectively, said paths being electrically isolated from one another; a first microwave junction formed at the first port comprising a first predefined gap, in a predefined winding length section, between each of the first terminal conductors and the second conductive path; a second predefined gap between end sections of the first terminal conductors; and a third predefined gap at a section where the first terminal conductors transition from the winding section into a first terminal end section.

The transformer may further comprise a second microwave junction formed at the second port comprising a fourth predefined gap, in a predefined winding length section, between each of the second terminal conductors and the first conductive path; a fifth predefined gap between the end sections of the second terminal conductors; and a sixth predefined gap at a section where the second terminal conductors transition from the winding section into a second terminal section.

The transmission line transformer is arranged to have a characteristic impedance which is substantially equal to that presented at the first and second ports, and with ferrite or non ferrite material core. The transformer provides an operating bandwidth in excess of 2 GHz, and operable at data speeds of 2G or greater. A transformer system, comprising a mounting member which carries a plurality of isolating transformers described above can be arranged.

A method of providing DC isolation in a data communications system, the method comprising connecting the isolating transformer according to any preceding claim with one port connected to a computer, computer modem, or data communications equipment, and the other port connected to a transmission line or the like, and in which the data communications system is configured to transmit and/or receive data to and/or from the further transmissions line.

Figure 10a depicts graphical results of a transformer with 10-turns according to this invention. The curves show the differential scattering parameters in the frequency domain. The x-axis represents the frequency in Hz, and the y-axis the scattering parameter values in decibels (dB). The insertion loss (IL) between ports is the differential scattering parameters Sdd21 and Sdd12 (one for each direction). The differential return loss (RL) is the scattering parameters Sdd11 and Sdd22, respectively for port 1 and port 2. This device had 10 turns in windings around a small high permeability toroidal core (as illustrated in Figure 11). However, the realisation of the Junction is less than ideal, and as a result there is significant ripple in the pass band that limits the 3dB-bandwidth of this device to around 2.5 GHz. The dashed line towards the top of the graph represents the I L to be expected from equal length of equivalent transmission line directly connecting the ports instead of the transformer. The dotted curve depicts the limit of RL specified by the IEEE in 802.3 for the Media Dependent Interface (MDI). RL should be above this line.

Figure 10b depicts graphical results of a transformer according to this invention that differed from the device whose results given in Figure 20a substantially only in the way that the gap between the conductors in the Junction were prevented from opening. This Junction is a better realisation of the invention, and as a result the scattering parameters show greater bandwidth (4.8 GHz), and greater margin over the IEEE MDI RL limit.

Figure 11a depicts graphical results for a 34 -turn device otherwise similar to the 10-turn whose results are depicted in 10b. A notable thing here is that the IL is very close to the IL to be expected from an equal length of equivalent transmission line directly interconnecting the ports. It was found that this was characteristic of devices as the number of turns is increased. Over the band of interest the differential scattering parameters of a transformer of this invention can be all but indistinguishable from an equivalent length of transmission line.

Figure 11b depicts graphical results of a transformer nominally identical to that presented in Figure 11a, but with degraded Junctions. A larger gap was allowed to form between the conductors at the corners in the Junctions, much as those of Figure 10a. However it was found that the scattering parameter of a device like this, with a greater number of turns, do not degrade quite so dramatically as when there are fewer turns. This is attributed to an increase in dampening factor introduced by the longer length of windings. Figure 12a depicts graphical results derived from the same differential scattering parameters as Figure 10a, but depicted as the step response in the time-domain. The effect of the poor Junctions is evident in the reflection step responses of the two ports (tdd 11 and tdd22) as spikes. The location in time of these spikes clearly indicates the Junctions as the origin of the reflections that make for a poor RL. Similarly the transmission step responses (tdd21 and tdd12) show reverberation, which is the cause of the ripple in the IL of Figure 10a. The delay between the main response and echoes is consistent with a reverberation between the two Junctions.

Figure 12b depicts graphical results derived from the same scattering-parameters as Figure 10b. Like figure 12a, they are the step responses in the time domain. The reduction in the height of reflection compared to Figure 12a is evidence of the improvement in the Junction. The higher rise in the transmission response and the reduced reverberation are also artefacts of an improved realisation of Junction.

Figure 13a depicts graphical results is derived from the same s-parameters as Figure 11a, and depict the step response in the time-domain. This response is that of the 34-turn device with the better Junction realisation.

Figure 13b depicts graphical results is derived from the same s-parameters as Figure 11b, and depict the step response in the time-domain. This response is that of the 34-turn device with the poorer Junction realisation. Although the height of the spike in the reflection step response at time zero (corresponding to the near end), is greater than that of Figure 13a, it is notable that there is little returned from the distant end (at time 1.3 ns). This contrasts with Figure 12a, for the device with fewer turns, where the distant end reflection is clearly evident. This would be due to increased dampening (due to increased length of transmission line loss in the windings).

MULTI-TURN ULTRA WIDEBAND

The following section of the invention describes a multi-turn transformer that can have wideband performance comparable with a single turn device, as described above, of the same winding length.

Freeing the design of the transformer from the constraint of a single turn enables simultaneously: greater bandwidth, better Open Circuit Inductance (OCL), smaller size, higher saturation limits ( Isat and VxS/turn), lower core loss, lower radiation loss, better EMC performance, reduced crosstalk, less variation with temperature, increased temperature range, and permits use of two centre taps (CTs), say one with a ground or Bob Smith termination and the other for transceiver ( PHY) bias. In more detail, Isat is the value of current that will saturate the core. It is the limit amount of current that the transformer can handle without distortion; and OCL stands for Open Circuit Inductance and is the inductance of the windings when not otherwise loaded.

Further, the freedom to reduce the size enables Ethernet transformers to be made using new and existing miniature chip constriction techniques, such as might be derived from multilayer and conductor wound chip inductor designs. These techniques can be highly automated, and very cheap. As a result smaller and better performing Ethernet transformers can be developed for all markets at lower cost.

Two forms of structure were studied: a toroidal helix, and a cylindrical helix (i.e. solenoid). For a toroid the bandwidth per unit length of winding actually increases as turns are added, albeit with a limiting factor of a little over 2. For a solenoid (which is topologically compatible with binocular, pot, bobbin, and E- cores), the bandwidth per unit length of winding reduces slowly as turns are added. However a figure of merit ( FoM) of Isat x OCL x Bandwidth per unit winding length still increases significantly as turns are added.

The immediate implications for the development of a split binocular core is that OCL and Isat can be improved as needed without comprising bandwidth.

Below are generally recognised rules in relation to transformer design

• Form windings from uniform transmission line (in the sense that the characteristic impedance and phase velocity are uniform along its length);

• Use transmission line of characteristic impedance half that desired for the ports;

• Arrange the ports so that they are on opposing positions around the windings such that the principal time domain reflection at one port for respectively open- and short- circuit connections at the other follow with the same delay with respect to the signal applied;

• Uniformly distribute the loops of the windings;

• Keep the distance between each pair of turns substantially greater than the distance between the two windings;

• Couple to the ports using transmission line of characteristic impedance equal to the characteristic impedance of the ports; and

• A core can be used to extend performance at low frequency. To date, the applicant has simulated with distances between the loops that are substantially larger than the distance between the windings, and it remains at this time only a supposition that this is necessary.

The applicant had earlier concluded that inter-winding capacitance introduces an unavoidable roll-off in the insertion loss (IL) at a lower frequency than the upper limit of a single-turn transformer of the same winding length. An experiment was performed in which a two-turn transformer was constructed on a large toroidal core around which coaxial windings were formed into a “figure of 8”, and a very poor bandwidth resulted (approximately 1 GHz for a winding length of 40 mm).

An experimental transformer developed by the applicant has the loops of the windings cross each other in close proximity in the centre, but are maximally distant where the opposing ports are located. It was only the sheath of the coaxial transmission line used for the windings that prevented the loops electrically shorting together as they crossed.

Such a structure neither has a uniform spacing between the loops of the windings, nor a minimum spacing that is substantially larger than the distance between the windings themselves.

It is thought that, if the turns are uniformly spaced a greater distance from one another, the structure may perform well.

The experiments leading to the present invention question whether these surmises are correct. The applicant uses simulation as it allows precise control of the geometry, permitting constructional difficulties or inaccuracies to be excluded. The simulator has 3D full-wave numerical codes for both frequency domain and time domain models.

Maximally simple solid models were used to get early results. The only materials employed were the non-physical lossless Perfect Electrical Conductor (PEC) and free space - but this does not change the substance of the conclusion we can reach respecting relative bandwidth of single-turn and multi-turn transformers, but rather promotes good fidelity for the simulation, and speeds simulation time.

Figure 15a and 15b illustrate a multi-turn transformer embodiment in accordance with the invention, in 8-turn and 52-turn forms, respectively. The topology is that of a helix wrapped around a toroid. The toroidal structure was chosen for the multi-turn design as it admits a purely analytical description of the geometry, which greatly simplifies its control by parameters, and has perfect symmetry - which is important for a device that is to have electrical symmetry.

A 50 Ohm transmission line of two circular conductors was used, which was wrapped around a toroidal space. The conductors are disposed as inner and outer around the toroid, rather than side-by-side, as this has a simpler analytical description of the geometry for a uniform impedance.

The model is controlled with numerical parameters is shown in figure 25. Parameter N controls the number of turns in the model. The model currently supports only an even number of turns. The location of the second port is different for an odd number of turns, and this was not parametrised.

For comparison a single turn transformer with windings formed into a circular loop was modelled. This is shown in Figure 14.

Figure 16 illustrates a solenoidal transformer with nine turns. A solenoidal construction (Figure 16) is not so geometrically pure, as it needs a means to connect the open ends of the solenoid, which necessarily destroys its symmetry. This topology is consistent with the use of a binocular core - though in that case, one would likely arrange for the loops to be more rectangular in shape, perhaps with one dimension substantially longer that another.

The simulations results for a multi-turn toroidal device responses are of the same comb-filter form as a single turn loop.

Figure 17 shows the simulation result for a 2-turn helically wound toroidal transformer compared with a circular single turn transformer with the same length of winding. The two devices have the same bandwidth.

Figure 18 illustrates the S21 (IL) for a 24-turn (N=24) helically wound toroidal transformer compared with a circular single turn transformer with the same length of winding. It is evident that the 24-turn device much more the bandwidth.

Figure 19 shows the resonant frequency of a toroidal device is consistently higher than that of a single turn device. Figure 20 shows the bandwidth of a toroidal helix improves with the number of turns in the helix, though with diminishing returns above 20 turns or so.

Figure 21 depicts a normalised measure of bandwidth of the transformer versus the number of turns formed in the windings. This is the number of cycles of a sinusoidal signal distributed around the winding at the fundamental resonance (i.e. at the lowest frequency of resonance) on the y-axis (vertical-axis) for a given number of turns of winding on the x-axis (horizontal axis). One can see in the figure that for a winding made with a fixed length of conductor the bandwidth for a toroidal winding increases with the number of turns formed, whereas for a cylindrical winding it falls (reduces) with increasing number of turns formed. As can be seen in Figure 21, the bandwidth of a solenoidal transformer can be less than that of a single-turn loop (or a toroidal) one, but it can also be more than that of a single-turn loop. It depends on some specifics of the geometry wherein one result that has more bandwidth is a long solenoid with small diameter.

Figure 22 illustrates the bandwidth of a solenoidal transformer for a given length of winding. Experiments concluded that the multi-turn transformer works in just the same way as a single turn device. A toroidal design typically has superior bandwidth performance than a solenoidal one, but one gets the benefit of multiple turns in terms of improvements in Isat and OCL for a given bandwidth.

It was found that the relative bandwidths of the classes of transformer forming part of the experiments (single-turn circular loop, toroidal and solenoidal vary from unity only by a small factor, whereas the benefits for OCL for a given winding length vary as linearly with N (the number of turns), and the OCL-/saf product for a given winding length has a maximum of N=4. It is concluded that it should be possible to get the L x Isat and bandwidth needed for NGAUTO PoDL in an attractively small package with a multi-turn split-bead design. Other applications that don’t demand such a high Isat can have greater OCL and be smaller and or deliver greater bandwidth.

It will be appreciated that the above-described embodiments are purely illustrative and are not limiting on the scope of the invention. Other variations and modifications will be apparent to persons skilled in the art upon reading the present application.